DIY multi flash sync

Parallel port logic analyzer

DIY wrist strap holder for D2H

Turvasytytin ja elektoninen sytytin ovat sellaisia, jotka
huomaavat rikkoutuneen loisteputken ja lopettavan sytyttämien.
Näitä on harmittavan vaikea löytää koska tavaratalot pitävät
yleensä myynnissä vain tavallisia sytyttimiä. Turvasytyttimiä
saa hyvällä lykyllä sähköalan erikoisliikkeistä. Hinnat ovat
yleensä aika kovia mutta esim. seuraavassa verkkokaupassa
varsin kohtuulliset 1,84 euroa.

 http://www.e27.fi/kauppa/index.php?act=viewProd&productId=174

Turvasytyttimen tunnistaa siinä olevasta punaisesta napista.

 http://www.mattikaki.fi/sahkoturvallisuus/turvasytytin.jpg

Jatkot elektroniikkaan.

TESTING ESR OF ELECTROLYTIC CAPACITORS

I recently found an easy and cheap way to test ESR (Equivalent Series Resistance) of electrolytic capacitors, in circuit, that might save some people a lot of time. It requires only an oscillosope and a simple signal generator.

I had an oscilloscope that I was trying to repair (Intensity control had little effect. Horiz sweep was only halfway across screen at higher freqs. One power supply rail was too low and others were too high.) and I had already checked every electrolytic capacitor in several/many different ways (all in-circuit), and even compared each of the readings to those from an identical unit: Powered off: Looked at signature from component tester (single-curve tracer) across each cap, and from each end of cap to ground, did resistance check with DMM, did capacitance check with DMM, checked resistance from each end of cap to ground. Powered on: Put scope across each cap, and scope from each end of cap to ground, used DMM and measured DC and AC voltages across each cap and from each end of cap to ground. I did find some bad caps (and some other bad components) and replaced them. But the problems were still there!

I had been wanting to order an ESR meter, but hadn't done it yet, and needed to get this scope repaired immediately. I went to Sam Goldwasser's excellent repair-FAQ site, at http://www.repairfaq.org/ and found a GREAT method for testing ESR of capacitors in-circuit that requires only a signal generator and an oscilloscope (and some cables), and had found and fixed the problem within about ten minutes!! Here's what I did (This technique is basically directly from Sam's repair faq site):

I used a signal generator and an oscilloscope to set up what I now call an "ESR Scope": At the output of the generator, I connected a BNC "tee" adapter. I ran one 50 Ohm BNC cable from the tee to a good (Tek 2465A) scope (with a 50-ohm BNC terminator on the scope input). On the other side of the tee, I connected another BNC cable that had alligator clips on its other end (It might have been a 75 ohm cable; shouldn't matter too much?), which I clipped onto the banana plugs of a set of cheap DMM-type probes.

(Terminator note: I used a Tektronix 50 Ohm "pass-through" terminator, on the scope end of the BNC cable. But, you should also be able to use, instead, another BNC "tee" on the scope input, with an "endcap" terminator on one side and the cable coming in on the other side of the tee. A standard 10BaseT Ethernet 50 Ohm coax terminator (and 50 Ohm Ethernet BNC coax cables) should work fine. And they're available at Radio Shack, and probably Staples, et al.)

I set up the signal generator to produce square waves at about 100 kHz, with about 100 mv peak-to-peak amplitude as seen on the attached scope, and no DC offset (A simple 555 timer circuit would do the job, too!). Then, I turned the scope's v/div to 5 mv/div, with time/div at 1 microsec, with AC coupling of the input.

Shorting the probes together gave me a display on the scope that was about one division high. It was basically a square wave, with large narrow peaks at each leading edge. But I only looked at the horizontal part's p-p amplitude.

That's the whole setup! No resistors. No nothing. Just cables (and a terminator). I did also try it with a decade resistor box in series with the probes, just to see what it would look like. I could clearly see each one-ohm increase, on the scope display, with the probes shorted together as well as with the probes across a good electrolytic capacitor.

When I applied the probes across a GOOD capacitor in-circuit, there was little, if any, change in the scope display, compared to when the probes were shorted (since, depending on the frequency, a capacitor should look more-or-less like a short circuit, to AC). But, when I tried it across a BAD capacitor, usually the display would be almost-totally off the screen. And, there were some caps that looked marginal, making the display go from about one div p-p up to about 3 to 5 divs (which probably corresponded with somewhere between 5 ohms and 20 ohms of ESR, if I recall correctly.)

Anyway, within just a few minutes I had found one more bad electrolytic filter cap in the power supply, two smaller bad electrolytics in the P.S., a bad one on the horizontal sweep switch's board, four bad ones near the middle of the main board, and a couple more that I can't remember right now.

I made a note of each one. When I was all done checking, the first thing I did was replace the filter cap in the power supply, and then power it on and check the power supply rails' voltages. BINGO!!!!! YESSS!!! They were all normal again! Not only that, but the horizontal sweep problem and the Intensity control problem were both GONE!! Yippee!

That filter cap had checked out as perfectly OK, using every one of the other methods that I described above (all were "in-circuit", though), and compared OK to the other identical scope's same cap, in all of those cases. But with this "ESR Scope" method, it was totally obvious, immediately. And the same cap on the other scope tested good, with this method (So, the earlier comparisons WERE bad cap vs. good cap, but showed nothing!). [I also noted that after the bad cap was removed, it tested bad in the same way that it had while it was in-circuit, with a basically identical scope display. And all of the other ones that I replaced also tested bad, when OUT of the circuit, even with the other methods.]

This " ESR Scope " method isn't a perfect panacea, of course: There were some cases where, without an identical unit to compare to, the displays would have been difficult for me to interpet, and possibly misleading. (However, it *always* worked with every *electrolytic* that I tried it on, IIRC, from 10 uF 10v to at least 1000 uF 100v, with no need for an identical unit to compare to.) But, then again, I haven't played around with it enough, yet, either. I assume that adjusting the frequency for different capacitances might be helpful, especially if non-electrolytics were to be tested. I also seem to remember that a DC offset in the signal is usually used, when testing ESR. I'll try that, later. And maybe increasing the amplitude of the square wave would be useful, sometimes, too. But, usually, I think I'll want it to be low-amplitude, probably less than +/- 0.4v, so the signal doesn't turn on any semiconductor junctions.

High Voltage B+ Regulator

An amplifier is a device that modulates the power supply to the loudspeakers.  In this regard, the power supply is of the utmost importance.  There are numerous resources on the internet for high voltage regulators.  Most use a zener diode control the gate of a MOSFET or BJT:

This type of regulator is simple and is controlled by a single zener diode.  The 220uF capacitor across the zener filters the zener’s output, and also provides some turn-on delay through the 8.2kW resistor. 

DC Motor Control Circuit

   Notes:
 Here, S1 and S2 are normally open , push to close, press button switches.  The diodes can be
 red or green and are there only to indicate direction. You may need to alter the TIP31 transistors
 depending on the motor being used. Remember, running under load draws more current.  This
 circuit was built to operate a small motor used for opening and closing a pair of curtains. As an
 advantage over automatic closing and opening systems, you have control of how much, or how
 little light to let into a room.  The four diodes surriunding the motor, are back EMF diodes. They
 are chosen to suit the motor. For a 12V motor drawing 1amp under load, I use 1N4001 diodes.

12V halogen dimmer

I use a 12V 20W halogen lamp (MR16) and a 4.2Ah SLA battery for my bike light system. The battery has only limited life at this power rating, so I designed this cheap light dimmer to reduce the battery drain and allow for longer rides at night.

Based on a simple 555 timer circuit and Mosfet switch Q1, it works by pulse-width modulating the 12V supply to the lamp. The 555 (IC1) is wired as a free-running oscillator, with two different mark/space ratios selectable via a 2-pole, 5-position rotary switch (S1).

The third switch position bypasses the electronic circuitry and connects the lamp directly to battery negative. This gives three power levels of about 7W, 13W and 20W.

A logic-level IRL530N Mosfet with a drain-source "on" resistance of only 0.1Ω ensures low losses and eliminates the need for a heatsink. An STP30NE06L Mosfet would also be suitable.

A Practical PWM Circuit

LM324 pin connections (top view)

This uses the LM324, a 14-pin DIL IC containing four individual op-amps and running off a single-rail power supply. The sawtooth is generated with two of them (U1A and U1B), configured as a Schmitt Trigger and Miller Integrator, and a third (U1C) is used as a comparator to compare the sawtooth with the reference voltage and switch the power transistor.

Rather than have the fourth op-amp sat there doing nothing, it's used as a voltage follower to buffer the reference potential divider. The high input and low output impedance of this draws very little current from the PD, so high value thermistors can be used in the thermal version of this controller.

Doing it discretely.

The next method is one devised by Winfield Hill, co-author of the classic Art of Electronics textbook, and posted by him in this sci.electronics.design thread (post#20).

As simple as it gets, and it works well.

When powered up, Q1 base is connected to 12V through the load and R4; Q2 gate is connected to about 10V (the potential at the R1-R2 junction). The significant gate capacitance of the MOSFET coupled with the high R1 value delays Q2 turn-on, Q1 switches on first and pulls Q2 gate low keeping it off.

With Q1 on, any charge in C1 leaks away through R2; if the momentary-on button SW1 is pressed, Q1 base voltage is pulled low, turning Q1 off and allowing Q2 to turn on, switching the load to full power instead of the few microamps it's been limited to by the path through R4 and Q1.

With Q1 off, capacitor C1 charges up to 12V through R1 & R2, so when the button is pressed again, it applies its voltage to Q1 base through R3 turning Q1 back on and so turning Q2 and the load off.

Some components shown are optional – the LED and its resistor R5 just indicate when the switch is on, diode D1 is to prevent damage from back-EMF pulses with inductive loads such as brushed motors or relays. Capacitor C2 reduces any HF noise on the supply.

Transistor Q1 can be any low-power NPN type, for ease of fitting with the leads in the right order, ie, 'base' lead in the middle. A BC337 or BC547 fit as shown below, a 2N3904 will need spinning 180° so check the pin-out for whatever type you use.

The n-channel power MOSFET Q2 also has plenty of alternatives such as the IRF630 or BUZ71; look for gate capacitance C_ISS around 500pF or more and a low on-resistance.

The mains voltage monitor

It's very important to choose a resistor that can support the desired voltage or connect two or more usual resistors in series. The power of that resistor is quite low and a 0.25 W is usually enough.

Nikon hot shoe pinout

This is the pinout when looking at the bottom of the flash:

+-----+
|o    |
|2 3  |
|  O  |
|4   5|
|o   o|
+-----+

The actual functions are most likely more complicated than this (and will certainly include a digital protocol between camera and flash to transfer the lens/shutter settings), but this was enough for my flash and this project:

• 1: Ground (side contact of the shoe)
• 2: No idea (maybe AF assist light? This flash has one.)
• 3: Short to ground to trigger flash (the classic center-pin-trigger)
• 4: Short to ground to quench flash
• 5: ca. +5V from flash when charged (not used in this project)

Optoisolated Adapter for Older Flash Units to Low Voltage Cameras

(From: Brian L. Zimmerman (blz@home.com).)

Many electronic flash units can have a very high voltage between the trigger contacts that are shorted to trigger the flash. For example, the trigger voltage of the "Digi-Slave" DSF-1s flash unit being sold in 2001 for use with digital cameras (See for example, Slave Flash Products) measures 218 V fully charged. This would be dangerous to use on many modern electronic cameras such as Canon digital cameras which specify no more than 6 V.

This design adapts such a high voltage flash for use on a low voltage camera by using an optocoupler to electronically isolate the camera's contacts from the high voltage. It can then be triggered by the camera using the 6 volt supply from the flash unit's batteries, but also works at a lower voltage. The use of a triac optocoupler has the added advantages of using fewer parts than other optocoupler designs and can use the power from the flash's trigger circuit to fire the SCR switch instead of a separate power source.

The adapter circuit can be inserted into the lines inside the flash unit going between the flash trigger circuit and the flash unit's contacts as shown below. Since there are only 4 small parts in this circuit, there is a good chance that you can build it right into the available space inside the flash unit's case.

(WARNING: High voltage precautions apply here - be sure to safely discharge all capacitors!)

The following schematic is available as a PDF in Optoisolated Adapters for Older Flash Units to Low Voltage Camera or in ASCII, below.

System connections

              o--------------------- | ----------------------- (+)
   Flash hot-shoe contacts           |                Flash trigger circuit
              o--------------------- | ----------------------- (-)
                                    \|/
                               Cut and insert
                           adapter circuit below

Flash Adapter Schematic 1

                  +---------------------+      R2
 (-) o-----------(1)---+         +-----(6)----/\/\----+----o Flash Trigger (+)
                  |    |  OPTO1  |      |     5.6K    |A
 To Hot-Shoe      |  __|__   ____|____  |           __|__
 (and camera)     |  _\_/_-> _\_/_/_\_ (5)NC        _\_/_   SCR1
                  |    |         |      |           / |     400V,6A
 (+) o---+   +---(2)---+         |      |          |G | K   (RS#276-1020)
         |   | NC(3)             +-----(4)---------+  |
         /   |    +---------------------+             |
     R1  \   |          Optotriac                +----+
    330  /   |           MOC3010                 |    |
         \   |         (RS#276-134)              /    |
         |   |                               R3  \  +_|_ C1   See note 5.
         o   o                             100K  /   --- 1uF
        (+) (-)                                  \  - |
   To 6V Battery (Flash unit battery pack)       |    |
                                                 +----+----o Flash Trigger (-)

Parts list

1. SCR1: SCR, 400V, 6A, (RS#276-1020, $1.29, TIC106D, etc.).
2. OPTO1: Triac optocoupler (MOC3010, RS#276-134, $1.99)
3. R1: 330 ohm resistor.
4. R2: 5.6k ohm resistor.
5. *R3: 200k ohm ressistor.
6. *C1: 1 uF, 300 V capacitor (electrolytic or other).

* R3 and C1 may not be needed depending on holding current of SCR1.

RS# indicates Radio Shack part numbers. Total cost (October, 2001) is $3.75.

Operation

The camera shutter shorts the hot-shoe contacts, causing current to flow from the 6V battery through the IR-emitting diode at pins 1 & 2 of the OptoTriac. The current is transferred via light pulse which switches on the triac at pins 4 and 6 of the OptoTriac which is powered by the voltage from the flash trigger circuit. Current flows into the gate of the SCR, switching it on and causing discharge of the flash trigger through the anode and cathode of the SCR. The 330 ohm resistor (R1) limits the current through the hot-shoe to about 18 mA, and the 5.6k ohm resistor (R2) limits the current through the triac to about 40 mA. You may need to use a different value for R2 for your particular flash and SCR, since this is based on a 218 V trigger voltage. The triac can handle a larger current (1.2 A peak), but SCR's typically only use a small gate current for triggering.

Note

The circuit with component values shown seems to work reliably (at least so far) for this particular combination of Canon digital camera and slave flash. Others may be quite different. Some info can be found at Kevin Bjorke's: Non-Canon Strobe Page with a List of Trigger Voltages. Just knowing the trigger voltage isn't really enough information as it doesn't imply anything about the available current. Adding an input buffer using a transistor or CMOS gate would eliminate this as a concern.

SCRs and triacs should be driven hard when they are controlling high current sources to make sure they turn on quickly and minimize time where they are passing significant current with a significant voltage drop. The optotriac's output is current limited so this isn't much of an issue. However, the SCR discharges the trigger capacitor through the trigger transformer and this could amount to several A switched in a few microseconds. A gate current 10 to 20 times the minimum spec in the datasheet is recommended so long as this doesn't exceed the maximum rating in the datasheet. In any case, the worst that will happen is that the SCR will fail or become unreliable after running with marginal gate drive - no great loss considering its cost.

With minor modifications, this circuit could also be used with SCR triggering. All that would be required is to add a small capacitor in series with the camera/hot shoe so that the SCR would turn off. (Similar to what's shown in the output of the circuit, above). A high value resistor should be put in parallel with the capacitor to discharge it between shots. Using 0.1 to 1 uF with 100K ohms across it will probably work.

Zener Protected Adapter for Older Flash Units to Low Voltage Camera

(From: Jean-Paul Brodier (jeanpaul.brodier@free.fr).)

(Page en français : Adaptateur de flash à haute tension (ancien) pour appareils photo (récents) qui ne supportent qu'une faible tension aux bornes du contact de synchronisation.)

The adapter above would not work with my camera and probably not with a number of other ones either. It is certainly able to trigger the flash once, but you'll have to cut the auxiliary power before the next shot. I explain: the camera triggers the flash by means of a thyristor. As long as there is some current flowing through the thyristor, the circuit stays closed. So the auxiliary power source keeps powering the LED of the optocoupler. I tested and confirmed that behaviour on my FZ20. Unless your camera works with a transistor and blocks it after a given time, you'll not trigger two flashes without somehow disconnecting the power source.

The adapter I devised does not have these drawbacks and it works without any auxiliary source.

Here it is:

Flash Adapter Schematic 2

The following schematic is available as a PDF in Zener Protected Adapters for Older Flash Units to Low Voltage Camera or in ASCII, below.

                                    R2      R1
     (+) o-----------+-------+-----/\/\----/\/\-----+----o Flash Trigger (+)
                     |       |     4.7M    4.7M     |
                     |      _|_ C1              ____|____
                     |      --- 22nF            _\_/_/_\_ Triac TO92
 To Hot-Shoe   ZD1  _|_,     |                   /G | A1  400V,1A
   (camera)   5.6V '/_\      +------------------'   |
                     |     __|__ D1                 |
                     |     _\_/_ 1N4148             |
                     |       |                      |
     (-) o-----------+-------+----------------------+----o Flash Trigger (-)

One suggested triac is the TICP206D (RS 638-481). Even the old-timer TIC201D (TO220) was successfully tested!

The camera never sees a voltage higher than about 5 volts, which should be acceptable even for the Canon G1. :-) The voltage from the flash unit may vary from 6 V to 300 V! The capacitor loading resistance of 10 Mohms is split into two 4.7M resistors. Usually, small resistors withstand but a voltage limited to 200 V or so. The resistor loads the capacitor through D1, up to a bit less than 5.6 V as the current is very low due to the huge resistance R1+R2. When the internal thyristor of the camera closes, the upper pin of the capacitor is connected to ground. The lower pin thus applies a negative voltage on the gate of the triac, the diode being then reverse biased and blocked. The triac fires the flash. A thyristor would not do the job, it has to be a triac to be fired with a negative pulse (relative to A1 terminal).

The time delay to reload the capacitor between two successive flashes (computed, not tested) is 0.154 second. Any flash unit needs a bit longer to recharge and be ready for the next shot.

Portable systems often use LEDs of different colors-white for the display backlight, green for keypad illumination, red for power, etc., and in varying quantities of each. Typically the LEDs are driven by at least two power supplies, one for "standard" LEDs (red and green) and one for white LEDs (white LEDs require a higher forward voltage.) The keypad and other indicator LEDs each have current-limiting resistors. The circuit in Figure 1 drives the LED strings via transistors Q1-Q4, which operate as current mirrors. This technique offers the following advantages: eliminates the current-limiting resistors drives groups of dissimilar LEDs requires just one power supply voltage allows each string to operate at a different current and allows the brightness of all the LEDs to be adjusted with one control point (U1's ADJ pin). Figure 1. In this LED-drive circuit, a switching converter (U1) and associated components let you mix LED quantities and types. Transistors Q2-Q4 mirror the current in diode-connected transistor Q1. Note that the Q1 current-set string (LEDs D3-D5) should have an equal or larger voltage than that of subsequent LED strings. (If it doesn't, the current-mirrored strings may not have enough voltage overhead to function properly.) You can easily meet that requirement in the first string, by placing either LEDs with larger forward voltage drops (such as the approximate 2.8V to 3.7V range of white LEDs), or a greater number of similar LEDs. Then, the subsequent strings with lower voltage burdens can be easily accommodated. The matched-transistor current mirrors maintain a constant and equal current in all LEDs, regardless of quantity and type. That configuration allows the use of a single power supply and a single point for adjusting LED brightness. Any power difference between the reference string and a mirrored string is dissipated in the current-mirror transistor for that string: Pmax (transistor) = (VOUT - 300mV - VLEDs) × ILEDMAX. The current-sense resistor value is R2 = 300mV/ILEDMAX, where ILEDMAX is the sum of currents from all the strings. (For a comprehensive circuit and parts list, refer to Maxim's MAX1698 EVKit data sheet.)

Here is what you need:

1 10k trimmer. A cheap one is fine, the setting is not too exact.
1 1k resistor
1 10k resistor
1 1.8k resistor
1 1 ohm 2W resistor (one of the big square ones)
2 2N3906 or pretty much any TO-92 PNP transistor
1 2N4401 NPN (or any TO-92 NPN transistor)
1 IRF1405 N channel MOSFET (or equivalent, see below)
1 heat sink for the MOSFET (TO-220 variety)
1 mounting screw and insulator for mosfet and heat sink
1 regulated 24V@1A power supply

Here is the circuit:

 24V o-----o------------------o--------------------o +
           |                  |
           |                 .-.
           |                 | | 1k
           |                 | |                 Batt Here
           |                 '-'
           |                  |
           |              .---o---.
           |              |       |           .----o -
          .-.           |<         >|   ___   |
      10k | |<----------|    PNPs   |--|___|--o
          | |           |\         /|   1.8k  |
          '-'             |       |           |
           |              |       |           |
           |              |       |        ||-+  N-Channel
           |              |       |        ||<-  Mosfet
           |        .---- o-------)--------||-+  Use Heat Sink
           |        |     |       |           |
           |        |     |       |           |
           |        |     |       |           |
           |       .-.     \|     |           |
           |       | | NPN  |-----)-----------o
           |   10k | |     <|     |           |
           |       '-'    |       |           |
           |        |     |       |          .-.
           |        |     |       |          | | 1.0 Ohm
           |        |     |       |          | |  2W
           |        |     |       |          '-'
           |        |     |       |           |
           |        |     |       |           |
 GND o-----o--------o-----o-------o-----------'

   Adjust by putting a 10k resistor between the outputs, and
   setting the - output to 6V using the trimmer.

(created by AACircuit v1.28.6 beta 04/19/05 www.tech-chat.de)

How it works:

The mosfet and NPN transistor form a fixed current source. They limit the
current through the battery to a value near 700mA. The two PNP transistors
compare the output voltage with the voltage on the wiper of the trimmer.
When the output voltage on the - terminal gets down to the voltage on the
wiper, the current source is turned off. As the battery keeps sucking
current, this differential pair will keep the voltage constant.

You adjust it by putting a 10k resistor across the output (instead of the
battery). Then, use the trimmer to set the - of the output (using your
multimeter) so that it is 18V below the + output.

You could probably get away with a 20V Power supply with these parts. With
an IRF1405, you might be able to skip the heatsink. However, you probably
won't be able to get one of those at radio shack. Get a couple of TO-220
case N channel mosfets and a heat sink (along with some heat sink paste) and
a mounting kit. Handle the mosfets carefully, because you can blow out the
gate pretty easily with a static shock.

Regarding NPN vs PNP (in case you don't know)

For NPN, the collector is the more positive one, and current flows from
collector to emitter.

For PNP, it is the other way around, and the emitter is the more positive
one, and current flows from emitter to collector.

You need an LM317!

The LM317 is an adjustable regulator that can output 1.25V to 37Volts! It does this with two resistors and a feed back loop. You simple pick two resistors based on the datasheet equation:

Where R1 is 240-Ohm. You may ignore the I_AJD term for the most part.

And hook them to the 'ADJ' pin as follows:

The capacitors C1 and C2 help and are good engineering practice. But if you don't got them, don't fret. They are only AC surge suppressors.

Our LM317 configured for 5V operation

With R1 as 240-Ohm, R2 as 720-Ohm, V_OUT you get 5V out.

If you assume a dc current gain, H_FE of 50 for the 2N3055 power transistor, then the effective value of the smoothing capacitor would be 50x this value; or be the same as using a 5000uF capacitor without the power transistor. The graph below shows the output voltage and current through the load :-

Q We’ll have it back in a minute. Somebody, Asa, wanted to know, are 741s widely used in new designs?

Relatively not so much. They’re still excellent for certain things like driving capacitive loads, but a lot of people are doing low voltage work, and 741s are not really suitable for anything below plus or minus 5 volts. And people don’t use them for that much anymore. They’re not cheap, they’re not excellent for anything, they’re not low noise, they’re not wonderful, but if you have them around, use them.

Simple white noise generator

This two-transistor white noise generator has a surprising feature – about 30dB more noise than the more traditional designs.

Q1 and Q2 can be any small-signal transistors with a beta of up to 400. The reverse-biased emitter-base junction of Q1 provides the noise source, which is fed into the base of Q2. Q2 forms a simple amplifier with a gain of 45dB. The improved output level is due mainly to the inclusion of C1, which provides a low-impedance AC source to the noise source while not disturbing the DC bias of Q1.

The low amount of feedback also makes this circuit very resistant to oscillations and tolerant to circuit layout. Unfortunately, the truism of "no such thing as free lunch" also applies; C1 makes the circuit very sensitive to power supply ripple.

Q Asa has a very interesting subtle question about banging to see if the op amp rings. He wanted to know, do you do that on the output of the amp?

You can do it at the input, you could do it at the output, you can do it on the power supply. It’s not that critical how you do it. You do it to see if it’s ringing. If it isn’t ringing, you don’t have to do it very often.

Okay, my turn. A guy called me up; I think it was honestly more than 20 years ago. He said, “Well, we just plugged in our power supply to our circuit board, and we put the minus and plus 12 volts on the plus and minus 12-volt terminals. We want to launch it tomorrow. Should we launch it? You know, the minus and plus 12 volts was only applied for a few seconds. And I said, Gee, if it’s working at all, it's still horrifying. Throw it away. I would not take that LM108A/MIL38510 and put that in my car radio. It does nothing trustworthy about any op amp with a minus or plus 12 volts on the supplies. Throw it away. And then a year later, some other unfortunate guy came back with the same question, a different guy. I told him the same answer. And I said, By the way, what if you put some anti-reversal diodes on there? And he said, Wouldn’t adding the diodes hurt the reliability? And I said, Compared to the reliability you got after you blew up the LM108, no, it would greatly improve reliability.

We’ll discuss several other little things you could do. What about anti-reversal stuff? Radio — an in expensive radio. You put your battery on it and then you absent-mindedly cross it. Isn't it wonderful that 9-volt batteries have these little prongs and you can put one between it and short it out, and if you put it — and touch them in there, it’ll blow up? Isn’t that a shame? But, if you put a diode in series, it wouldn’t blow up. Except now, you lose a lot of your battery life because if this gets to 6-volts, so this is 6.6, you’ve lost a fraction of the voltage of your battery. It’s less efficient, it’s wasteful, your battery life is less. Then hey, here’s another great idea. You put a diode, a big rectifier (1N4002) across the inputs. Yeah, you put it across the battery, and it croaks the battery and rips out everything in sight — explodes your battery, especially a car battery. Oh, it’s wonderful.

Well, those aren’t such wonderful solutions, although if you don’t care about your voltage, you could set 12.6 volt supplies; you’d get about 12 here. Some people could use this. Some people could use this if you have a current limited supply. If you’re always using a 12-volt supply that you could short and it doesn’t care. This is not a terrible solution. It doesn’t waste any power; it just gets warm when you cross the power supply coming in. But this is not wonderful. And I said, How about this? And I invented this, and I said, “This is such a wonderful invention, I’m going to apply for a patent on it.” And we searched and nobody had invented this thing up until about 1990, but one guy had done it about eight months before I invented it. You think, oh, in the 1980s. Why didn’t anybody think of this? Well, by finally in 1989, some guy did invent it. He invented it a few months ahead of me. But, you put our N-channel FET in here and you put it backwards. So instead of having the current flowing in the drain and out the source, it flows in the source and out the drain, which is okay. This resistor is optional; I just put it in there for a joke. You could put zero ohms, you could put a mega ohm; it doesn’t make any difference. Well, the mega ohm would turn on a little slower. Now, does this have to be a big fancy FET? If you need a lot of load current, you could put in a $2.00 FET and switch dozens of amperes with a tiny IR drop. If you were switching in for a radio, you could just use a 2N7000 that just has a few ohms of impedance; nobody really cares. Now you put it the correct way, the correct power supply, and this thing turns on and the current goes through it. You swap this power and this thing just turns off. You turn this minus and this doesn’t turn on; it just does nothing. This is a reasonably good circuit and that’s on page 164 of this book, or whatever it is. So, this is a fairly good what would you call it? A fairly good application to avoid the damage and the loss of series diodes or the disaster of shunt diodes, and it has very low loss. And it only costs, you know, a dime or two.

Here’s another good solution to a system. You make the PC card edge connector a little bit shorter for your power pins. You plug in your ground, and you start to shove it in this way, at this angle, and this hits before those. You shove it in that way and this hits before those. No matter what you do, ground is always connected before the power supplies. And then you can install your anti-reversal rectifiers here. As I was saying, when you plug it in, this is pretty safe, and you can have an anti-reversal thing on your plus 12 and your plus 5 and your minus 12. And you could even have an anti anti-reversal between your plus 5 and your plus 12 so that this doesn’t go appreciably above the plus 12. That is a very good system solution for many cases.

Q That answers Myron’s question about, do you believe in analog simulators, especially for AC simulation?

If -- if you applied it right and if you knew it was going to be lousy, and if you got an answer saying it wasn’t lousy, you’d say there’s something wrong with spice. You have to be smarter than SPICE. I’m giving a three-hour session Tuesday night and Thursday night on SPICE down at San Jose State. And I’m warning people, and I’m going to show them this example: don’t trust SPICE. You have to be smart enough that when SPICE lies, you have to say something is wrong.

Q Here’s a great question from Kim that I know you know the history of where the bootstrap comes from, the term when we say we "bootstrapped an amplifier."

All right. Let’s say we have an op amp that has a certain input impedance. We want to drive it, drive the input. But it looks like a big lag because here’s a CR, looks like a big lead. This resistance is a heavy load. But, if we drive this and then take the midpoint of this capacitor and drive it again, this is called a bootstrap connection where as this goes up and down and this goes up and down and this goes up and down, there ain't no voltage across here. This is pulling this up by its bootstraps; the amplifier has to drive this. This goes back to when you had cathode followers and the bootstrap resistor comes in here and the bootstrap comes in here. And that’s only about what, 70, 80 years old?

So, that’s what it’s called. It’s like picking your boots up by your own bootstraps, which with boots you can’t do it, but with input impedances, you can.

“Fools you are when you say you like to learn by your experience. I prefer to profit by the mistakes of others and avoid the price of my own.”

We should really point out that: “the best model of a cat is a cat; preferably the same cat.” And that was Norbert Wiener who invented that one, that saying. I tend to agree with it.

Now, Bob Pease says, “How can I help you get your analog task done with less grief?” Well, two general rules are: Always use an appropriate CF feedback cap around your op amp, and always add appropriate bypass caps on your amplifier or on your circuit. Now, let’s go into A in a little more detail. On a general-purpose op amp, you should always add a small feedback cap, 20 or 200 pico farads, unless you can prove it is not needed. On a current feedback amplifier, you should never add CF unless you can prove it’s okay.

How do you prove it? Section AA-Prime. Don’t just look to see if your circuit is oscillating; bang on it and see if it rings. Hey, that’s pretty deadbeat. It didn’t ring. If it rings, you know it may not be oscillating right now, but it might be soon. So, watch out for that; don’t just see if the oscillation dies out. If it doesn’t die out darn fast, you’re close to trouble.

And B - power supply bypassing. Every group of one of linear circuits needs some ceramic and some electrolytic or tantalum bypass caps. Hot, fast amplifiers really need to get a few capacitors right next to the power supply terminals of that amplifier. And if you have a hot amplifier that’s faster than 2 MHz, 4 MHz, a data sheet will remind you, yeah, you got to add your microfarads. And if it’s a really hot op amp like 100 MHz, 200 MHz, it will remind you, you have to have the capacitor about a tenth of an inch maximum away from the power supply pins. If in doubt, bang on the thing.

Let's say we have a circuit like this, and we want to bang on it. So we’ll take the output and just put in a square wave. And, 0.01 and 10K; you bang on the output, and it would ring. I mean you take this output right here and put 10K in a square wave generator and the output would ring a little bit. So, that’s what I mean by banging on it. You don’t have to literally bang the table, although that’s pretty deadbeat. At least it’s not going, ding, ding, ding. And, with a circuit, you want it to die out. Now this circuit, with the R and the C, you could bang on this until it, sort of, it might overshoot just a little bit, but it settles out and does not give much problem. And, when you’re asking it to have low slew error, it would do pretty good at having the ramp error go away.

Q Martin wants us to say more about ferrites, but in what context?

I love ferrites in switch mode regulators. I don’t use them for a damn thing else. And if you do, you may know a lot more about it. But, I can’t possibly answer every question about ferrites – enough. Okay?

Another possibility is some amplifiers, if they’re small, can overheat with any kind of extra heating source, such as, you hit the rails with it, you drive the input, your output hits the rails, and the internal drivers can get too warm. How about a nice little op amp that’s small and it’s got about 100 MHz bandwidth, and you put a capacitive load on it, and the poor little dear breaks into oscillation and hammers as hard as it can to drive that capacitive load, and it gets hot. Please avoid that. Now, we have some excellent new op amps, which are rail-to-rail on the input, rail-to-rail on the output; it is bipolar, it's not CMOS. But it has almost perfect infinite tolerance of CL. You put a capacitive load on it; it slows down. You put more load; it slows down some more. But it doesn’t go berserk.

Another guy said, “Your LM317, the output is oscillating at 90 hertz.” And I said, “The LM317 cannot oscillate at 90 hertz, but are you talking about the tiniest little LM317L in the smallest little package?” He said, “Yes, how did you know that?” And I said, “Well, it’s going into thermal limit.” And thermal limit often goes in and out of operation at about 90 hertz. So, I told him to go back to put a little tiny bit of heat sinking on it to get a little bit of heat out of the poor little thing, and it should be okay.

Oh, let’s see. Let’s talk about the kinds of diodes. With 1N914’s, they’re fast; they can actually take 75 milliamperes DC and a couple hundred milliamperes of transient pulse. But they are leaky, five, 10 nanoampers at room temp. Much worse if you ever let them get warm. So you normally don’t use 1N914s, unless you can tolerate one of the world’s leakiest diodes, although I’ve got to say this is leakier. Something like the 1N484 and 1N457, ordinary low leakage diode, you can’t even find them anymore. They’re low leakage but they’re not very fast, and they’re not very low leakage either. 1N4002, it’s amazing. This is a one-amp rectifier. What's it got for leakage? A few picoamperes? Nobody’s going to guarantee that, but you can easily go out and select something inside 50 picoamperes and you’ll never find a bad 1N4002. It will leak as bad as 50 picoamperes. So, for some applications, hey, you could put in some 1N914s, you could put in some 1N4002s. The 1N914s will turn on fast; these won’t. These will handle the ampere and these won’t. So, there's a whole bunch of funny things you can do with store-bought diodes. And we’ll also discuss in a little while the 1N5817, which is a Schottky.

This thing can actually carry the three amperes, and it can turn on in less than a nanosecond. Unfortunately, there are some cases where one nanosecond is not fast enough, and we’ll be discussing that later.

There’s one other thing you can buy if you really need blast proofing is a Transzorb, and I think they still make them, and they’re much better than zeners. They say, “Well, it’s like a zener, but it’s about 20 times more robust at taking sheer damage and current times voltage than any zener. You can get them in all sorts of voltages. I don’t know how much they cost these days, but there are applications where a Transzorb will protect you, and a lot of these circuits don’t quite protect you.

Q Bob, how about a JFET hooked up as a diode for low leakage? I’ve seen that in circuits.

There are JFETs that have low leakage, but they’re not a hell of a lot better than a 2N3904; those are subpicoampere, also.

Now, let me drag out of my book, which has a lot of junk information and a lot of useful information. It has Appendix E somewhere in here, and has a list of various diodes. And we got a Schottky rectifier. I’m going to go down this list. The Schottky rectifier is very high leakage. It’s got several micro-amperes. There are other Schottky rectifiers that will leak milliamperes, but they will soak up and absorb amperes, like that 1N5819, 5817, 5818, it’s all in the same family.

B: Piece of Germanium. Look how soft it is. It's soft as a grape. Now, F is sort of interesting because F is a big old 1N4001 family which doesn’t leak very much. It’s not high conductance, it doesn’t have a good slope. Here’s a good slope, but it’s a healthy old rectifier. Here’s a good slope, L. What is L? L is a 2N3904, base emitter junction. That is one of the steepest curves; it’s just about theoretical, and is almost as good, but that’s a LM3046, which is a much smaller device.

What make is K? Oh, LM194; that’s an even bigger device. So there’s a very nice clamp if you need a clamp; it’s about $2.00 per clamp. You wouldn’t normally waste your money. Whereas, L is very steep, M is pretty shallow, M is the collector base junction of the 2N3904. So the collector-base junction is really inferior at having a poor slope, and the base emitter junction is very steep; it has a very good slope. So this is one of the things I like, every time I could find my book, I can always find this information. It’s never going to go away. But I can always find it. It’s always useful to publish some things that you can find later.

What if you have large signals? If you have plus or minus one volt, you can protect it with clamp diodes to ground, but for plus or minus 10 volts, or five volts, you don’t want to fool around with that. So you clamp it against the supply. What diode do you need? Like if you’re trying to go real fast, the 1N914 might be one picofarad; it might be appropriate. How much blast proofing do you need? You have to engineer it. There’s no simple answer. You could put low leakage diodes with 10K, 10K, 10K; low leakage, not much delay, or you might put a meg, a meg, a meg. It’s not a big deal. There are many things you can do. So, what ”R” values? Whatever you need, you have to engineer it, 1K, 10K, a meg. Why would we put two 10K resistors in series? If you blast this with ESD sparks, you’re walking across the rug, and you reach over and you touch the input to your little amplifier, then you buy a new amplifier because you killed it. Well, to avoid that, you could put 10K in series with 10K. And if you put a couple thousand volts across 10K these days, 10K tends to break down across its spirals because most resistors these days are spiral-cut with thin-film, thick-film resistors. You used to be able to buy, for a penny, 10K Allen Bradley quarter watt, or half watt. You put a couple of them in there and they do not break down. And they can survive large KV without sending through huge shoot-through currents.

Oh, I never said that, nope. People have asked me, why don’t I just put one capacitor from the plus rail to the minus? There may be cases where that’s a good idea; I can’t name one. If you try it and it’s okay, you have my permission to try it, but I don’t think I’m going to recommend it. I think you’re better off with one bypass to each rail from ground. In most cases, I think it will work much better. There might be cases where it’s not much better, but I don’t recommend that. I think you’ll find it’s not a good solution.

Q Okay. Martin clarified his ferrite question. He wants to know if he can use a ferrite to keep the noise from the switching power supply out of an analog linear circuit for the power.

In some cases, I think he’s right; you could do that. In other cases, it depends upon how low a noise you need. If -- if he only needs an improvement of 5dB, and the ferrite gives him 10dB, he should go away happy.

I just published an article. This engineer should go to the recent ”What’s all this Ripple Rejection Stuff, Part III"? And I will tell you where to find the darn thing. Go to www.national.com/rap and we show several applications where you can use an amplifier to get that noise down possibly a lot better than a ferrite. It depends on the frequency. Every application like that, every case is different. You go to ED Columns, and you go down just about a half inch down in the List of Columns, and it’s right there. Oh, I should write the name of it: "Ripple Rejection Part III." You can read "Ripple Rejection Part I and II," which are not — Part II isn’t that bad. And Part III gives more results, and that might be useful because if you need a lot of ripple rejection improvement, I can’t tell from this distance whether you have a chance with a ferrite. It depends on the frequency; it depends on the ferrite. Go ahead and try it because if it don't work, you know it doesn’t work. And if it does work, you’re done

A runt pulse is a very small pulse, smaller than expected. Like you’re expecting a 10 MHz clock, which would be 50 nanoseconds per cycle. If you then have a 5-nanosecond pulse and that’s a runt pulse, and many systems can’t handle that. So, runt pulses do happen. But this guy who asked, “What is the LM6762 doing giving a runt pulse?” We do not have enough information to answer that today. But if you tell us more about it, we’ll try to answer it later.

Adjustable Duty-Cycle Oscillator

In the study of electronics, the concept of duty-cycle pops up in various places such as digital circuits, one-shots, switching regulators, and D/A converters to mention a few. Lab experiments to examine duty-cycle usually require two parts: a square-wave oscillator driving a monostable multivibrator (one shot). A common circuit involves two 555 chips and a bunch of resistors and capacitors for each chip. Also, the RC time constant associated with the capacitor coupling the two 555s is critical.

In contrast, the circuit shown in Figure 1 can be built with a single IC, two capacitors, three resistors, two trim-pots, and a diode. The component values and time-constants are not critical. And with this simple circuit, here is what you can demonstrate:

• What is hysteresis
• What is a Schmitt-trigger input
• How can hysteresis be used to build a square-wave oscillator
• What is duty cycle
• How do you adjust duty-cycle to different values
• How does duty-cycle relate to DC value
• What is a low-pass filter
• How does filter cut-off relate to square-wave frequency
• How does filter time-constant relate to speed of response to changing duty-cycle

Figure 1

Of course, with such a simple circuit you would expect that there was some kind of trade-off. And you're right: this circuit changes frequency as you change duty-cycle. But even with that limitation, you can still demonstrate all the items listed above. And you can use this circuit as a lead-in to a follow-up experiment with a circuit that maintains constant frequency as you vary duty-cycle.

The key to the circuit is the CD4093 CMOS digital IC. It's a quad two-input NAND gate chip with Schmitt-trigger inputs. In an inverting configuration, driving the inputs high will force the output low, while driving the inputs low will force the output high. The value of the input voltage that causes the output to change is the switching-threshold. The switching-threshold on a Schmitt-trigger input is not fixed; it has one of two different values depending on whether the output is high or low. In the 4093, the input voltage to force the output low is higher than the input voltage that forces the output high (see Figure 2). The result is a hysteresis effect.

Figure 2

We all have experience with the property of hysteresis. It can be seen in an old-fashion oil-can, the kind with a long flexible spout on a semi-spherical can with a wide, flat bottom. You pick the can up with one hand: the spout between your fingers and your thumb on the bottom. As you press on the bottom of the can, nothing happens until there is enough pressure to "pop" the bottom in and a squirt of oil comes out. As you release the pressure, the bottom will "pop" back out at less pressure than it took to "pop" it in.

The way hysteresis leads to oscillation can be seen in an automobile with a loose "front end". As you turn the steering wheel, nothing happens at first. Then, at a certain point, the car will turn. As you turn the wheel back to "straighten out", again nothing happens until you get to a point where the car suddenly swerves the other way. The result is that, as you travel down the road, the car is swerving left and right. You can't get it to go in a straight line. In effect, you're oscillating.

In this circuit, let's assume that C1 is discharged, making the input of the IC low and causing its output to go high. The voltage on the high output is fed back to the input through R1, R2, D1 and R3. The resistors limit the current, so C1 charges up with a certain time-constant. When the Voltage on C1 reaches a certain point, call it V1, it will be high enough to force the output low. At that point C1 will start to discharge through R3 only, since D1 will be reverse-biased. When the Voltage on C1 drops to a certain point, call it V2, it will be low enough to force the output high again. The cycle then repeats, and we have made a square-wave oscillator. Note that it is necessary that V1 be a higher value than V2 so that there is a fixed amount of Voltage that C1 must charge and discharge to produce a cycle. Then by changing the resistance, the time needed to charge and discharge (and thereby the frequency) can be changed.

The square-wave output can be seen by placing the probe of a oscilloscope at output 'A'. We will define the "on-time" of the cycle to be when the output is high, and the "off-time" when the output is low. Duty-cycle is defined to be:

                   On-Time
Duty-Cycle =   -----------------
              On-Time + Off-Time

Since C1 discharges through R3 only, the "off-time" of the cycle is fixed. But C1 charges partly through the R1-R2 path, so adjusting the R2 trim-pot will vary the "on-time" and thereby vary the duty-cycle. Since the square-wave output goes from 0 to +12 Volts and back to 0, it can be thought of as an AC signal riding on top of a DC voltage. The value of the DC can determined from the duty-cycle by the relationship:

DC Voltage = (Duty-Cycle) x (High-Output Voltage)

where, in this case, the high-output voltage is 12 Volts. The square-wave output is fed through an R-C low-pass filter made up of R4, R5, and C2. The purpose of the filter is to "smooth-out" (or "integrate") the square-wave. In effect, it removes the AC signal and leaves only the DC Voltage on C2. the filtered voltage can be seen on output 'B'. Just how "smooth" the DC will appear (i.e. how much AC "ripple" will be seen) depends on the RC time-constant of the filter. The "cut-off" frequency (f0) of the filter is given by:

       1
f0 = -----
      2pRC

Define N to be the ratio of the oscillator frequency (f) to f0 as follows: N=f/f0. Then the bigger N is, the smoother the DC will be, as will be seen by adjusting the R5 trim-pot. One note about using CMOS gates: don't leave any inputs "floating". Be sure to ground all unused inputs on the CD4093 chip.

So have some fun with this circuit. Play around with it and try different values for the resistors and capacitors. Try reversing the direction of D1 and see what happens. And see if you can think of some interesting applications for it. Maybe you can get it published in this newsletter.

Cheap Hardware Evaluation Boards

19. October, 2007 in Hardware by Michael Neumann

TI-MSP430 MICROCONTROLLERS  

The MSP430 series from Texas Instruments is an ultra-low power family of RISC based microcontrollers. They are extremely cheap and have a fantastic support plus easy availability by way of samples from TI.

During our work in the development of the General Purpose Bluetooth Interface we had to code the Bluetooth L2CAP layer in software and run it on a microcontroller. We did a big survey of the available uCs and their specifications. After taking many factors into account and jumping from one uC to another, we finally settled for this the MSP430F149 which is one of the most easy to use and program microcontroller. These were the major advantages that we saw while using this chip:-

• Had all the features that we needed
• Easy availability :: Free samples from TI
• Cheap cost ~$ 6 = Rs. 300/- (for MSP430F149, others are cheaper)
• Free gcc compatible C compiler: MSP-GCC for both Windows and Linux
• Free software: Insight or IAR C-SPY to burn the code into the uC Flash
• Excellent user base and support (http://groups.yahoo.com/group/msp430)
  

The other options that we were considering were the ATMEL ATMega series of uCs. The factors which went against them were cost and availability in India. Also, only the high end uCs were available which were way beyond our budget. Apart from the MSP430F149, there are also a lot of other uCs in the same family which can be used for different applications and needs.

One word of CAUTION is that there no in-built support in the MSP430 series of uCs through which external memory can be attached to the chip. If you use these uCs and reach a point wherein the memory has to be extended, you will need to do all the bit-banging and addressing through the software which can cause unneccesary trouble.

Some Tips:

• We used the dongle (or JTAG) available at http://www.olimex.com as the hardware for programming the uC.
• A proto-board is also available at http://www.olimex.com but there seems to be some problem in it w.r.t. one of the crystals. They have attached some external capacitors with the Low Frequency crystal which cause problems in its operation. We removed them when made our PCB and programs which did not run on the Olimex board, worked peacefully with our board.

History

The MSP430 family is a microcontroller family which is established for approx. 10 years. However, the primary usage was in measurement applications which are battery powered, e.g. intelligent sensors, with or without LCD-display. Peripherals were included with these applications in mind. In the beginnings, no UARTs were supported. And the development tools were not very attractive for those having smaller target quantities in mind. In short: The MSP430 was a good choice for OEM´s.

By end of the nineties the MSP430 family and its development tools have become attractive for a vast range of potential applications and also for low quantities, since the cost of developent tools has been lowered dramatically, especially due to the introduction of JTAG-based programming & debugging in flash-based devices. Alas, competitors in the low budget range already control these low budget markets, especially Microchip (PIC) and later Atmel (AVR). Of course, the oldmobile under the 8-bit controllers, the 8051-family may be considered a part of these low-budget markets.

Changeover Cons

Given this constellation, it is very difficult for the MSP430 family to raise its market potential, despite its significant better properties in all relevant issues. Of course, this is due to present usage of these competitor products and corresponding development tools. Further, the time which was spent in learning a new architecture is considerable (for a quick overview of MSP points of differences, look at Architectural specifics). And last but not least, the software pool of these products is immense. But the fact of common usage of C lowers this as a factor of keeping "old" products. The only problem is the conversion of low level peripheral related modules to new modules. This site will help you to establish such a conversion. See SW-(re)sources.

Changeover Pros

For medium to bigger projects, there is an economical argument to switch over to MSP430, when current targets are 8051, PIC or AVR. The creation of software will be less error-prone and provides a better overview when MSP430 is targeted.

• An example of less susceptibility to software errors: the 16-bit architecture provides single transfer of up to 16 bit memory or peripheral values to registers and vice versa. This allows for variable updating in interrupt routines without the need to temporarily disable interrupts when reading such values in the main program. Assembler programmers traditionally are aware of this problem and mostly do not forget to insert di & ei instructions (or whatever these mnemonics are named). C-programmers however often forget these pitfalls. They firstly must study the macro names of their compiler which takes care for these interrupt enables & disables, and secondly: they must be applied.

• Why a better overview? This is due to the linear memory organization of the MSP430 architecture. In the past, the Harvard architecture was praised as being economical because of providing separate address spaces for code and data. Practically, this limits the comfortability of maintaining (high-level) programs, especially when more than one data space is available (PIC: RAM-banks, 8051: internal & external data memory). This affects maintainability, portability and reusability of programs & modules. This topic is extensively explained here. At this point it should be understood that long term effects are clearly gained when switching over from 8 bit to 16 bits with linear address space. Note that many 16 bit processors still have separate code & data spaces.

For the sake of completeness, it should be noted here that several good architectures exists which support linear memory and good code utizilation. Bigger projects, especially when program code is beyond 60KB are managed better with e.g.
- ARM family of controllers & cores (industry standard, licensed by ARM);
- HC12 family of controllers (Motorola, see also HC12WEB-page).

Development kits

The development kits of TI represent a clear argument to changeover to this controller family. When an entry to any controller right from the start shall be made, the kits will provide best price/performance ratio available on the market. More information here. Hobbyists who consider using these kits should read the tips found here.

Architectural specifics compared to common micros

• The port registers' direction (in resp. output) are commonly specified by data direction registers. With MSP430, a "1" represents an output. Most controllers bypass the port usage automatically, when an associated peripheral (e.g. UART) is activated. The MSP430 requires the programmer to specify this explicitly in a register. By default, all shared port/peripheral pins are selected to port usage.

• The MSP430 runs on an internal RC-based clock of approx. 800 kHz when started. For many applications, this isn´t precise enough. A certain sequence of commands must be carried out in order to select an external crystal clock up to 8 MHz. The sequence of commands is quite crucial, since a mechanism exists which let the MSP430 automatically switch back to the internal RC-clock when the external clock is lacking. An associated error flag must be reset prior to switching over to the external clock. A recommended sequence of C-statements is given in the SW-resources page.

Accurate clocks are also possible without HF-Crystals. A simple 32768Hz crystal can be utilized as a time reference. Software can adjust the RC oscillator in order to achieve a sufficient degree of accuracy for most applications.

• There is no bit addressabe RAM- or peripheral space, contrary to 8051, PIC and AVR. This is a matter of getting used with. Those who want to work with single bits may define bit fields in structures, which are associated with corresponding peripheral addresses.

However, this lack does not necessarily mean that the MSP is not effective with bit manipulation. The assembler language supports bit test instructions which tests bits against a mask (e.g. test if bit7 is set using the mask 0x80).  This way also multiple bits can be tested in one instruction.

• The instruction set has some peculiarities. Any basic knowledge about this can help optimize designs right from the hardware design start of a project. Regarding the hardware, one should try to assign port pins that must be accessible as single bits to bits 0....3 of any port. This will facilitate immediate mode with short constants, saving machine cycles and code space. The same applies to software flag bits.

To understand this - and also to better understand the assembler -, one must consider the addressing scheme of the CPU, which is described in the MSP430 family user's guide (chapter "RISC 16-Bit CPU").
Each instruction has its opcode packed in the first word, together with addressing/operand information bits. There exists 7 addressing modes, but only 2 bits seem to decode this for the source field. Further, 4 destination addressing modes are selectable. However, only 1 bit seems to decode this. Compare with the scheme for 2-operand instructions given here (other schemes have longer opcodes):

(S/D-Reg = source/destination register 0...16; As/d = source/dest. addressing mode; B/W = byte/word)

The crux lies in the register bank decoding for source operand and destination operand. There are 16 registers, of which 12 are general purpose. When one of the first four registers are targeted in source or destination field, another addressing mode is activied. E.g. Register 0 is the PC (Program Counter). For example, targeting R0 with indirect register autoincrement mode will inherently do immediate addressing! It is important to realize this: the user writes assembler with the common notation for immediate addressing (or the C-compiler does so), but actually an indirect register access is used.

Another example: when absolute addressing is selected, the assembler will use indirect indexed register addressing mode with R2. R2 actually contains the STATUS flags, but in such operation, it will read as zero, so the final target address simply equals the index.

Then finally an example related to immediate addressing with short constants. When the assembler sees immediate addressing (for constant setting, adds/subtracts and bit set/reset masks), with constants -1, 0, 1, 2, 4 or 8, it will target R2 or R3. Together with the "As" bit field setting, the appropriate constant is inserted instead of an extra operand word. This implies that R3 cannot be used as a register for addressing purposes. Of course, the register exists and it can be modified and read from (as is also the case with all special purpose registers). Please note that the machine cycles used for these instructions using short constants is not explicitly specified in the documentation.

Up to now all but R1 are used as special cases. R1 actually isn't a special case: R1 is the stack pointer. So indirect or indexed R1 operations facilitate stack relative addressing (which is not listed as one of the 7 addressing modes in the documentation).

Finally a note to those who are interested in technical details: indirect register addressing (with or without autoincrement) is possible only in the source operand. The destination can be targeted indirectly only via the indexed addressing mode, which will always "consume" an extra instruction word: The index, mostly 0. In this case, 0 is not a short constant, because the R2/R3 constant technique is not subject when another register which actually holds a pointer value is given as the destination field.

pyBSL

Software for the bootstrap loader. Works with Flash devices (MSP430F1xx and F4xx): erase and download new software or upload RAM or Flash data from the device back to the PC.

Features:

• loads TI-Text, Intel-hex and ELF files
• download to Flash and/or RAM, erase, verify
• reset and wait for keypress (to run a device directly from the port power)
• load addres into R0/PC and run
• password file can be any datafile, e.g. the one used to program the device in an earlier session
• upload a memory block MSP->PC (output as binary data or hex dump)
• download a program, execute it, resynchronize and uplaod results. (for testing and calibration)
• written in Python, runs on Win32, BSD, Linux (and other POSIX compatible systems) (and Jython)
• use per command line, or in a Python script
• downloadable replacement MSP430-BSLs, which also allows higher baudrates.

MSP430 DIP Programmer

Using the TI ez430

In this brief article I hope to show how to get up and running with those DIP samples you ordered from TI using the ez430. I'll be working with the same microcontroller unit (MCU) as on the ez430 target board, namely the MSP430F2013.  Any MSP430x2xx will work with the same method, and as far as I know the entire MSP430 line uses the same programming connections.  If you use a package other than a dual in-line package (DIP or DIL) or a device other than the MSP430x2xx then you'll need to refer to the datasheet for  the device to find the appropriate pin locations.

ez430

Boards

Programming an MSP430 requires only 4 wires, including the Vcc and Vss connections. The schematic below should help. These are the only connections you need to download a program to the MCU.

I've built two programming boards. The first uses a small breadboard, ZIF socket, and 4 wires from an old IDE cable with the 4 pin socket soldered to one end. The Zero Insertion Force socket just saves on effort when moving the MCU around.

 
The second uses a DIP socket from Mill-Max and some pins I had laying around, as well as some other components. I've wire wrapped most of the connections. It was only necessary to solder the 4 pin socket. Basically it's a proto-board without the oscillator. See here for the proto board schematic.

 
Table 2-2 of TI document slau144c (the MSP430x2xx Users Guide) shows the unused pin terminations. This is important for your project board, but not the programming. The MCU won't run your program unless you pull the RST pin high using the 47k resistor.

 
Another note on pin sockets. When you connect the ez430 pins to your 4 pin socket be sure that the Vcc connection is correct. You can verify this by looking at the schematic in the ez430 Users Manual. Notice that R10 is attached to Vcc on the ez430 side. You should be able to follow the lead on the board back to the nearest connector pin which is pin 1 of the connector.

 
As an aside, you could build the 4 wire programming connection, called Spy-Bi-Wire, into your final application, and you wouldn't have to remove the chip once it's installed. If you decide to do it that way, then you can power the MCU from your application's 3V power source and just connect pins 2 and 3 (see J1) on the USB programmer to your MCU.

 Another programming resource to learn to use the msp430 can be found at this website .

MSPFET - FREE MSP430 flash programming utility

Supports full range of Texas Instruments flash-based MSP430 mixed-signal MCU's.

Hardware support - FET (flash emulation tool) or BSL (bootstrap loader).

Try our new Fusee - multy-functional, easy to use and low cost programming adapter.

I believe 2N4391, 2N4392 & 2N4393 were very popular FET's
in industrial equipment.  They should cross to SK or ECG
and be readily available.

2N7000 is cheap (US0.10 in quantity) and ubiquitous. Specs are not
so impressive.

XXXX3055 seems to be popular for power N-channel, kind of a marketing ploy
giving it a number like that but it seems a fine device. Eg. Motorola.

One solution is to add an enable input to a MUX. When the enable is active, the output is selected from one of the inputs. When the enable is not active, then the output is Z.

A very simple circuit to experiment with AT90S2313, 2x16 LCD display and 4x4 keypad.

Using a Keypad and LCD Display with the MAXQ2000

Embedded systems which require user interaction must interface with devices that accept user input (such as a keypad, bar code reader or smart card acceptor) as well as devices that display information to the user (such as LED or LCD displays). This application note, using the MAXQ2000 microcontroller, covers the use of two such typical devices - a 4x4 switch keypad and an LCD display.

Miksi 8-bittinen prosessori on huono valinta moniajoon?

Ongelma *ei* ole tilanvaihdon järjestäminen, vaan osoitearitmetiikka.
Osoitteen laskenta on kahdeksanbittisissä vehkeissä aina pitkähkö prosessi.
Asian varmistamiseksi voi katsoa C-kielen base->component -tyyppisestä
rakenteesta kääntynyttä koodia.

Moniajettavassa koodissa kukin säie (tehtävä, prosessi, task jne) tarvitsee
identiteettinsä säilyttämiseen vähintään oman pinon. Tämä on hankalaa
8051-tyyppisissä prosessoreissa: pino on sisäisessä muistissa, joka loppuu
hyvin pian kesken, kun sitä joutuu jakamaan kaikkien säikeiden kesken.
Tiedän yhden 8051 -moniajojärjestelmän, joka rahtaa koko pinon ulkoiseen
muistiin ja toisen sisään aina tilanvaihdon yhteydessä (IAR:n O'Tool).

Tämän lisäksi kaikki yhteisessä käytössä olevat aliohjelmat joutuvat
pitämään työtilansa joko pinossa tai säikeen mukaan valitussa
muistitietueessa. Puuha onnistuu melko helposti, jos prosessorilla on
käytettävissä rekisteri, jota voi helposti käyttää kantarekisterinä.
Esimerkkinä käy 8086 -perheen rekisteri bx. Rekisterin avulla voi osoittaa
tietuetta, jossa olevaa komponenttia voi vielä indeksoida si- tai
di-rekisterillä. Rekisterirakenteen kannalta minimiprosessori, jolla moniajo
sujuu mukavasti, on 8086 tai 6809 perillisineen.

Miksi AVR ei oikein käy: siinähän on kolme kantarekisteriksi kelpaavaa (X,
Y, Z)?

AVR on rekisterirakenteeltaan RISC -kone: rekistereitä on status + 32
työrekisteriä. Tilanvaihto menee työlääksi: 33 rekisteriä ulos ja toiset 33
sisään. Keskeytyksen palvelussa voidaan ottaa toinen lähestymistapa:
talletetaan vain ne rekisterit, joita keskeytys käyttää - keskeytyksen
palvelu ei saa olla niin työläs, että kaikki tarvitaan.

MSP430 on turmellut merkittävän osan PDP:n säännöllisestä arkkitehtuurista:
PDP:ssä on 8 rekisteriä, joista yksi on ohjelmalaskuri (r7, pc) ja toinen
pinon osoitin (r6, sp). Kaikki käskyt, joissa on muistiosoitus, käyttävät
reksitereitä samalla tavalla kaikissa muistioperandeissa.

MSP430:ssa on 16 rekisteriä, joissa on RISC -koneesta napattu idea tuottaa
tiettyjen rekisterien tietyillä osoitetavoilla pieniä vakioita (-1, 0, 1, 2
jne). Kun rekistereitä on enemmän kuin PDP:ssä, ei kaikissa paikoissa voi
käyttää enää kaikkia osoitetapoja symmetrisesti (destination-operandilla on
rajoitetut osoitetavat). Lisäksi osa PDP:n osoitetavoista puuttuu kokonaan
(esim. auto-decrement).

Tämä on yksinään riittävä syy, että PDP:n tail-endin kopiointi ja pieni
muuttaminen ei riitä.

Been there, done that.

ulkoisen reset-piirin käyttö on joka tapauksessa "must" noitten
avr:rien kanssa.

vanhan prosessorityypin käytössä on etunsa, lastentauteja lienee
8031-sarjasta turha hakea. Samoin huomasin että vanha rca1802 cosmac
(joka on siis todella vanha, jo 70-luvun puolivälistä, ja hidas cmos-
prosessori) elää uutta renesanssia dallasin valmistamana, nyt tyypin
kuvaukseksi on vain vaihtunut "high reliability processor", ilmailu ja
avaruuskäyttöön näkyvät noita myyvän.

t: Jari Lehtinen

Kaikkeahan voi tehdä. Tiedän kaverin, joka rakensi kahdestatoista
8031:stä moniprosessorijärjestelmän. Kaikki prosessorit pääsivät
samoihin muisteihin käsiksi, kun niillä oli yhden kellojakson
erotus keskenään...

Eihän tuon prosessin vaihtaminen nyt kovin ihmeellinen operaatio voi olla.
Tallennetaan rekisterit ja vaihdetaan pinoa. Asiasta selviytyy sitä
huonommin mitä useampia rekistereitä on tallennettavana. Mutta ihan sama
ongelma on keskeytyksissäkin. Jokainen keskeytys aliohjelma on myös
prosessin vaihtavaa koodia ja niihinhän noita 8-bittisiä juuri enimmäkseen
käytetään.

Mikähän siinä prosessin vaihtamisessa niin hankalaa on,
rekisterit pinoon, pino-osoitin uuden prosessin pinoon ja
rekisterit ml. program counter pinosta ?

Olen itse ollut tekemässä muutamaakin erilaista  8 bitin
vehkeillä toimivaa moniajokäyttistä sekä myös käyttänyt
valmiita käyttiksiä. Moniajava ydin ajastin- ja I/O-palveluineen
on näissä ollut kooltaan 2-16 kB, joten 64 kB osoiteavaruudesta
jää vielä hyvin tilaa varsinaisen sovelluksen käyttöön. Mikro-
kontrolleri, jossa on 8 kB sisäistä ohjelmamuistia, on ihan käypä
alusta tuollaiselle yksinkertaiselle ytimelle. Toki ylimääräiset
palvelut voi riisua pois ja muutenkin on selvää että käyttis konfataan
sovelluksen mukaiseksi, eli muistia ei hukata 50 timeriin jos 5
riittää. Mutta varsinainen moniajo ja taskinvaihto vaikka
kellokeskeytyksen ohjaamana ei montaa sataa tavua vie.

Kyllä noita kasibittisiä vehkeitä on moniajettu ihan yllättävänkin
tutuissa käyttökohteissa ja tullaan varmaan vielä jatkossakin
käyttämään. Eivät kaikki käyttökohteet vaadi muutamassa
mikrosekunnissa tapahtuvaa taskinvaihtoa, kymmenien MIPS:ien
tehoa tai muistinsuojauksia.

A crude form of PLA can be produced by simply heating powdered lactic acid with powdered stannous chloride - commonly used in pottery glazes - in a test tube. Extracting it from the test tube afterwards is left as an excercise for the diligent student.

See papers in footnote for further details.

FET (Flash Emulation Tool) programming adapter

Bidirectional converter between TTL levels from PC's parallel port and MCU's JTAG interface. The MSP430 family supports in-circuit programming of flash memory via the JTAG port, available on all MSP430 devices. All existing IDE's for MSP430 MCU support in-circuit debugging and emulation of application program in target device by means of JTAG interface. Very useful and convenient thing during software development and debugging. Also it can be used for in-field program updates if security fuse is not blown. You can buy it from Texas Instruments distributors or third-party developers or make your own using, for example, see MSP430-JTAG schematic. Does not support fuse blowing.

BSL (BootStrap Loader) programming adapter

The MSP430 bootstrap loader (BSL) enables users to communicate with embedded memory in the MSP430 microcontroller during the prototyping phase, final production, and in service. Both the programmable memory (flash memory) and the data memory (RAM) can be modified as required. The commonly used UART protocol with RS232 interfacing is supported, allowing flexible use of both hardware and software. To use the bootstrap loader, a specific BSL entry sequence has to be applied to specific device pins. An added sequence of commands initiates the desired function. A boot loading session can be exited by continuing operation at a defined user program address, or by the reset condition. Access to the MSP430 memory via the bootstrap loader is protected against misuse by a user-defined password. Does not support fuse blowing. For more information refer to Application Report SLAA096 - Application of Bootstrap Loader in MSP430.

|                             |
|                             |
|                    ___      |
|   o----o-----o----|___|-----o
|        |     |     1K       |
|       /-/   ---             |
|        ^    ---             | Atmel uC
|1N4733A |     | 100nF        |
|       ===   ===             |
|       GND   GND             |
|                             |
|                             |
|                             |
|                             |
|                             '----
|
|
(created by AACircuit v1.28.6 beta 04/19/05 www.tech-chat.de)

A zener-based protection scheme has benefits over diodes to each
supply rail.  The speed of the diode isn't significant -- in fact,
you're adding the 0.1uF cap in parallel with the diode.  And if
there's a positive-going spike, the charge will flow to the +5V rail
with the two diode scheme, which may make it exceed maximum spec for
the uC as well as other components, depending on trace inductance, the
ESR of the caps from the +5V supply to GND, and other considerations.
Also, you may end up with a situation where your power supply voltage
regulator may have its output at a greater voltage than its input,
which can be a disaster for certain regulators.

Now, you have to attend to the layout here either way, so try to have
a good, low inductance, low resistance path for the ESD charge to
travel without creating a spike on the GND lines.  But from a
beginner, hobbyist perspective, a zener/cap combination with a series
resistor is a great start to preventing your uC from getting smoked by
a 100pF cap namely the user) discharging a few KV of potential to GND
through your project.