CA2009468C - Frequency control apparatus and method for a digital radio receiver - Google Patents
Frequency control apparatus and method for a digital radio receiverInfo
- Publication number
- CA2009468C CA2009468C CA002009468A CA2009468A CA2009468C CA 2009468 C CA2009468 C CA 2009468C CA 002009468 A CA002009468 A CA 002009468A CA 2009468 A CA2009468 A CA 2009468A CA 2009468 C CA2009468 C CA 2009468C
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- Prior art keywords
- frequency
- burst
- signal
- correction
- correction signal
- Prior art date
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Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03J—TUNING RESONANT CIRCUITS; SELECTING RESONANT CIRCUITS
- H03J7/00—Automatic frequency control; Automatic scanning over a band of frequencies
- H03J7/02—Automatic frequency control
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION, OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
- H03L7/00—Automatic control of frequency or phase; Synchronisation
- H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
- H03L7/08—Details of the phase-locked loop
- H03L7/085—Details of the phase-locked loop concerning mainly the frequency- or phase-detection arrangement including the filtering or amplification of its output signal
- H03L7/087—Details of the phase-locked loop concerning mainly the frequency- or phase-detection arrangement including the filtering or amplification of its output signal using at least two phase detectors or a frequency and phase detector in the loop
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B7/00—Radio transmission systems, i.e. using radiation field
- H04B7/24—Radio transmission systems, i.e. using radiation field for communication between two or more posts
- H04B7/26—Radio transmission systems, i.e. using radiation field for communication between two or more posts at least one of which is mobile
- H04B7/2643—Radio transmission systems, i.e. using radiation field for communication between two or more posts at least one of which is mobile using time-division multiple access [TDMA]
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
- H04L27/22—Demodulator circuits; Receiver circuits
- H04L27/227—Demodulator circuits; Receiver circuits using coherent demodulation
- H04L27/2271—Demodulator circuits; Receiver circuits using coherent demodulation wherein the carrier recovery circuit uses only the demodulated signals
- H04L27/2273—Demodulator circuits; Receiver circuits using coherent demodulation wherein the carrier recovery circuit uses only the demodulated signals associated with quadrature demodulation, e.g. Costas loop
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0044—Control loops for carrier regulation
- H04L2027/0053—Closed loops
- H04L2027/0057—Closed loops quadrature phase
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0044—Control loops for carrier regulation
- H04L2027/0063—Elements of loops
- H04L2027/0065—Frequency error detectors
Abstract
FREQUENCY CONTROL APPARATUS AND METNOD FOR A
DIGITAL RADIO RECEIVER
Abstract of the Invention A frequency control apparatus and method is disclosed in which a frequency offset between a received signal and the local oscillator (115) of a digital receiver is corrected in substantially one step.
DIGITAL RADIO RECEIVER
Abstract of the Invention A frequency control apparatus and method is disclosed in which a frequency offset between a received signal and the local oscillator (115) of a digital receiver is corrected in substantially one step.
Description
2~
FREQUENCY CONTROL APPARATUS AND METHOD
Background of the Iru~e~tion This invention relates generally to digital radio sy6tems and more particularly to the rapid correction of frequency error in a digital radio receiver which receives communication messages from a transmitter transmitting the communication 15 messages in bursts.
Radio receivers often utilize a circuit to automatically correct for discrepancies in frequency between the carrier frequency of the signal to be received and the frequency of a 2 0 local oscillator used in a superheterodyne receiver. Thi~ local oscillator converts the carrier frequency and associated information carrying sidebands of the received signal to a convenient intermediate frequency. The typical frequency correction process is performed over a relatively long period of 2 5 time, assuming that the carrier frequency (of the received signal) is continuously present. The carrier frequency may be tracked by means of amplitude detectors, discriminators, or the like to generate a frequency control signal. Some systems may use a pilot signal modulated onto the carrier frequency to 3 0 provide a reference in the receiver to derive the frequency control signal. tSuch a pilot control is further described in U.S.
Patent No. 4,541,118).
2 ~ 6 8 The frequency control signal i9 subsequently applied to the local oscillator from its initial frequency to a frequency which converts the received carrier frequency into an intermediate 5 frequency optimally placed within the selectivity of the intermediate frequency amplification and filtering stages.
Digital receivers present a new set of problems to such conventional automatic frequency control networks. One 10 approach to a fast re~eiver frequency control for a digital receiver has been disclosed in Canadian Patent Application No.
2~oo3~688-5~Digital Automatic Frequency Control on Pure Sine Waves", filed on December 16, 1988 on behalf of Borth et al.
Usually, digital receivers must process the received carrier frequency signal in a linear fashion. Such linear processing allows amplitude variation of the received signal to create further errors in the detection of frequency offset.
Furthermore, digital communication is often accomplished 2 0 using burst transmission techniques such as time division multiple access (TDMA). Burst transmissions do not provide a continuously available carrier or carrier plus pilot which can be employed by conventional frequency control techniques.
2 5 Summarv of the Invention Therefore, it is one object of the present invention to rapidly correct for frequency errors between a received signal and the frequency to which the receiver is tuned in a digital radio 3 0 receiver.
,' '~
3 CE0015~O9468 It is another object of the present invention to rapidly correct a receiver oscillator for frequency errors between a nominal frequency and a desired frequency.
It is another object of the present invention is to correct the 5 frequency error between the received signal and the correct receiver local oscillator frequency within a single frame of received data.
It is a further object of the present invention to determine the frequency error from a correction burst and directly correct 10 the local oscillator frequency.
Brief Dessriptior~ hQDrawin~
Figure 1 is a block diagram of a TDMA receiver which may 15 utilize the present invention.
Figure 2 shows the unwrapping process for both an increasing phase trajectory and a decreasing phase trajectory.
Figure 3 shows the result of a computer simulation.
Figure 4 shows a timing diagram relating the burst TDMA
2 0 transmissions to the frequency control of the present invention.
Detailed Descril)tion of the Preferred Embodiment A block diagram of a TDMA receiver which may utilize the 2 5 present invention i8 shown in Fig. 1. Such a receiver may receive burst communication signals in a timeslot from a TDMA transmitter 103 received on an antenna 104 and applied to quadrature demodulator 105 as a signal x(t). The quadrature demodulator 105 produces two quadraturely related 3 0 downconverted signals which are applied to conventional analog to digital converters 107 and 109 which digitize each of the two quadrature related signals before applying the digitized 4 CE0015~R ~9(~68 quadrature signals to a digital signal processor (DSP) function 111. In the preferred embodiment, the DSP is realized employing a DSP56001 available from Motorola, Inc. (or 5 equivalent). The DSP function 111 recovers a data communication signal using conventional TDMA techniques in a communication signal recovery function 113. Such signal recovery functions include channel estimation, channel equalization, and data demodulation. Also included in the DSP
1 0 function 111 is the recovery of a frequency control signal which allows the rapid adjustment of a voltage controlled oscillator 115 in the quadrature demodulator 105 so that the TDMA
receiver can be quickly brought into a condition where there is virtually no frequency error between the carrier frequency of 1 5 x(t) and the frequency of voltage controlled oscillator 115.
Because speed of adjustment is important, it is an important feature of the present invention that the adjustment of the frequency of voltage controlled oscillator 115 be substantially accomplished in one step without oscillator frequency hunting 2 0 or successive approximations to a final oscillator frequency.
Assume that the transmitted signal corresponding to a frequency correction burst (or pure sine wave (PSW)) is given by x(t) = A cos[~ct I cl)ot] (1) where:
2 5 A is the amplitude of the signal, = 2~1fC is the carrier frequency of the signal in radians/sec;
c~O = 27~fo is the PSW baseband signal frequency in radians/sec = 27~ (67.708...kHz).
3 0 In the absence of multipath noise, Rayleigh fading, etc., the received signal is given by (1). In general, the voltage controlled oscillator 115 (VCXO) frequency without frequ~e~y ~ , :
CE00158R 2~39 control will be given by (C~c + ~3) where ~CD corresponds to a frequency of~set error (in radians/sec). Assume that the VCXO output signal is therefore given by VI(t) = COS[(CdC + ~)t] . (2) After phase shifting vI(t) by -90 in the phase shifter 117, the resulting signal is given by VQ(t) = sin[(coC + ~o~)t] . (3) The output of the in-phase (I) mixer 117 (in the absence of any DC of~sets) is given by IA(t) = x(t) VI(t) = A cos [cl~ct + ~ot] cos[(c~c + ~c~)t]
= V2 A ( cos[(a~O - ~c3)t] + cos [(2coct + coO + ~c~)t] } (4) which, after low-pass filtering with filter 121 having a bandwidth of approximately 4 coO radians/sec, becomes Ig(t) = V2 A COS[(CDo - ~)t] . (5) Similarly the output of the quadrature-phase (Q) mixer 123 (in the absence of any DC o~sets) is given by QB(t) = X(t) VQ(t) = A cos [~ct + ~ot] sin[(cDC + ~ )t]
= V2 A ( sin [(2coct + CDo + ~c3)t] - sin [(coO - ~ )t] ) (6) which after low-pass filtering by filter 125 becomes Qg(t) = -V2 A sin [(~ ~)t] . (7) 5~009'~6~3 The two A/D converters (107, 109) convert Ig(t) and Qg(t) into their quantized, sampled equivalents, Ig(k) and Qg(k~, respectively. In the absence of any DC offsets: Ig(k) = I~(k) 5 and Qg(k) = Qc(k) Note that direct rneasurement of the frequency of~set ~CD via a frequency counter which counts zero crossings of the signals given in (5) and (7) will yield a measurement uncertainty of 0 freq. uncertainty = 1/measurement per;od = l/slot period = V0.58 msec = + 1724 Hz (8) which corresponds to an uncertainty of i 1.9 ppm at 900 MHz.
In certain radio systems, for example, a digital radiotelephone system for use in Europe defined in GSM documents GSM
05.01/3.1.0, "Physical Layer on the Radio Path: General Description" and GSM 5.10/3, "Radio Sub-System Synchronization", the required stability of the VCXO 115 must 2 0 be vithin + 0.1 ppm of the received signal coming from the base station. Therefore, an approach to measuring frequency errors which does not count zero-crossings must instead be employed. The apparatus and method of the present invention:
2 5 (1) translates the I and Q signals given by (5) and (7) by 67.7 kHz to DC (i.e., removes the ~0 term present in (5) and (7));
(2) computes the phase of the translated signal at regularly spaced time intervals;
(3) constructs the phase trajectory from the phase samples;
and -7 CE00158R 2~34~i (4) computes an estimate of the instantaneous frequency of the frequency-translated signals from the time-derivative of the phase trajectory.
I2C O~set Com~e~j~
The DSP function 111 accepts the quantized VQ signals and processes both the I and the Q signal through a direct current 10 (DC) subtraction process 129. An uncorrected DC offset is a result of mismatched I and Q channels and local oscillator leakage. An estimate of the DC offset of IB (k) is made by DC
estimate function 131 which averages the input signal the output of which is subtracted from IB (k) in digital adder 133 1 5 thereby producing IC (k), the I quadrature signal with the DC
removed. Likewise, an estimate of the direct current offset is made by DC estimate function 135 of the QB (k) signal and subtracted in digital adder 137 to produce Qc(k), the Q
quadrature signal ~,vith the DC removed.
In order to implement the above frequency control of the present invention in a fractional fixed point general purpose digital signal processor (DSP) such as the DSP56001, it is necessary to perform ~everal unique steps. Imperfections in 2 5 an implementation of the quadrature demodulator 105 necessarily results in DC offsets at the output of the quadrature demodulator 105 which must be compensated for in the frequency control.
3 0 The presence of DC offsets at the output of the quadrature demodulator 105 can severely limit the performance of the frequency control. To see why this iS the case, let Ig(t) and 2~ 68 Qg(t) in (5) and (7) contain DC o~set terms of magnitude C and D, respectively. Then Ig'(k)= 1/2Acos[((do~ )k] +C (5a) Qg'(k) = -1/2 A sin [(cdo - ~co)k] +D (7a) Then af~er frequency translation by ej~O k a8 described below 1 0 ID'(k) = Re {[Sg(k) + C + jD] ejcoo k}
= Re {[1/2 A ej(~30 - ~o~)k + C + jD] ei~l)O k) = 1/2 A cos ~G3k + C cos cook - D sin co QD'(k) = Im {LSB(k) ~ c + jD] eiC~O k}
1 5 = Im t[V2 A ei(~l)O ~ ~I))k + C + jD] ei~l)O k) = 1/2 A sin ~c~k + C sin c~ok + D cos ~ok .
The presence of the two quadrature components at a frequency cl~O prevent further downsampling (or decimation) 2 0 without introducing aliasing errors and can contribute to a significant error in the computation of tan~l ( QE(k) / IE(k) ) in (13) below. Removal of the two undesired frequency components in (17) below can be accomplished by two methods:
(1) Low-pass filtering of ID'(k) and QD'(k) to remove the sin 2 5 ~I)ok and cos ~I)ok components; or (2) Removal of the DC components prior to frequency translation by eJ~DO k.
The first approach requires a low-pass filter with a 3 0 bandwidth les~ than fo/10= 6.7kHz, for example, and thus having an impulse response of duration greater than 5/6.7 kHz 9 CE001~O9~68 = 0.73 msec, for example, an impulse response duration greater than the duration of a single time slot. Clearly this is an unacceptable approach.
The second approach simply computes the average DC
value of the two quadrature branches separately and subtracts these values from the respective branch signals:
J
IC(k) = Ig(k) - l/J ~ IB(k) k=1 J
QC~k) = Qg(k) - VJ ~, Qg(k) 1 5 k=l F re~a~n~y~anslati~n bv 67.7 ~Iz.
2 0 A TDMA receiver compatible with GSM specifications such as those mentioned above utilizes a frequency colrection signal which iB transmitted as a carrier shif~ of exactly 67.7 kHz during one timeslot of the TDMA transmission. A frequency translation of the I quadrature channel signal and the Q
2 5 channel quadrature signal by 67.7 l~Iz accomplishes the following result. The signal is translated to 0 Hz i ~c~/2~
thereby reducing the information data rate and permitting further decimination.
This i~ accomplished by a quantized 67.7 KHz signal in 3 0 quadrature, as generated by 67.7 KHz oscillator 139.
2~9~68 Translation is accomplished by complex quadrature mixer 141, which is a complex multiplication realized in the DSP.
Let SC(k) _ IC(k) + jQc(k) = V2 Ae~ o ~ )k (g) where IC(k) and Qc(k) are given above. After multiplication of SC(k) by e)~l)ok, we obtain SD(k) - ID(k) + jQD(k) = 1/2 A(ei~3k) = lt2 A cos~c~k + j 1/2 A sin~k (10) 1 0 i.e., ID(k) = 1/2 A cos~cdk QD(k) = V2 A sin~3k . (11) In the preferred embodiment, the TDMA modulation i8 1 5 conventionally processed by the transmitter in a manner known as GMSK which limits the occupied bandwidth of the transmitted signal. Since ~co will generally be small compared to the information bandwidth of the transmitted GMSK signal with random data (i.e., a normal burst instead of a frequency 2 0 co~Tection burst), SD(k) may be decimated by a factor of M to reduce the ~ignal processing complexity required for frequency control without reducing control accuracy. The decimated signal SE(k) is obtained from SD(k) by the operation 2 5 SE(k) = SD(Mk) (12) which is accomplished in function 143, a conventional decimator. I'hus, the M-1 samples of SD(k) between valid samples of SE(k) are simply deleted.
11 CE00158~0 Con~ut~tiQn of the pha~ of the decimated, tran~la~ ~ienal.
In accordance with known trigonometric identities, a 5 calculation of the phase relationship between the VCXO 115 output signal and x(t) can be made employing the decimated and translated I and Q signal samples. Thi~ phase relationship may be expressed as ~ (k) for the k-th decimated sample period. The phase samples ~(k) are computed from 1 0 SE(k) via the operation ~(k) = tan~1 ( QE(k) / IE(k) ) = tan~l ( sin~ k / cos~c~k ) = tan~l ( tan(~cl)k) ) 1 5 = A~lk (13) where the time interval of k is understood to be the decimated period.
Three problems arise in calculating tan~l ( Q} (k) / IE(k) ) in 2 û (13):
l. Division of QE(k) by IE(k) on a fractional fixed point machine is limited to the case where IE(k) ~ QE(k) > 0.
2. tan-l (x) is an inverse transcendental function requi~ng either a Taylor series expansion or a table-lookup.
2 5 3. The valid range of tan~l ( QE(k) / IE(k) ) extends beyond ~180l after phase unwrapping and hence the tan~1 function must be scaled appropriately for a fixed point machine.
All three problems may be solved by:
3 û (a) Using a small (256 word) ROM arctangent table over the range [0,45]. To use the table, a short (8 instruction cycle) division of QE(k) by IE(k) is required to yield an 8 bit ROM table 12 OE00158R 2~ 68 address. Note that over the arctangent ROM table range, the inequality IE(k) > QE(k) 2 0 is preserved.
(b) Extending the range of the arctangent table to _180 by (i) preserving the 8igns of QE(k) and IE(k);
(ii) noting whether I IE(k) I > i QE(k) I or vice-versa; and (iii) using a precomputed table of trigonometric identities to compute the arctangent function in each of the 4 quadrants.
(c) Scaling the ROM table values to the largest possible 1 0 range of the unwrapped phase ~o(k)~ This range is given by +V= ~[maximum allowable frequency error (Hz) duration of 1 time slot (sec) 360 + 180]
For example, with a VCXO 115 stability of +2.5ppm at 1 GHz, the maximum allowable frequency error is i:2.5kHz and the 1 5 phase range of ~o(k) i8 given by +[(2.5kHz)(0.58 msec) 360 + 180]
= i702 = iV .
All phase values must therefore be scaled by V=702 to prevent overflow in the phase unwrapping algorithm.
Construction of the ~hase tra-ectorv from the Dhase sam~les.
Provided that the signs of IE(k~ and QE(k) are preserved, tan-l ( QE(k) / IE(k) ) is defined only over the internal [-7~ ], or 2 5 equivalently only over the interval [-180, 180]. Since absolute phase i~ not known in the frequency control process, any gi~en sample of SE(k) will yield a phase sample ~(k) within this range. For example, if the output of the phase computation of ~(k) is used directly a~ the phase trajectory, and if ~(1) = 179, 3 0 ~(2) = -179, ~(3) = -177,.. , then determination of the instantaneou~ frequency offset via time differentiation of the phase trajectory of ~(k) would yield ambiguous reslllts due to 2~ 68 the apparent -368 phase jump between ~(1) and ~(2). In fact, if the phase were "unwrapped" in thi~ example by allowing the phase samples to take on values outside the range i 180~, it becomes obvious that the phase samples are increasing by +2 5 every phase sample.
In order to unwrap the phase samples, a phase unwrap process 147 is employed in the DSP of the present invention.
Conceptually, the unwrapping process for both an increasing 1 0 phase trajectory and a decreasing phase trajectory is shown in Fig. 2 and is accomplished by the following steps which are performed by the DSP 111 of the present invention:
Initialization: a(k), k=1,..., N from Step 2 Previou~ = ~(1) Sumphase = 0 Threshold = 90 DOk=2toN
Current = 0(k) + Sumphase 2û IF( I(Current-Previous)l <Threshold)THEN
~o(k) = 0(k) Previous = ~o(k) (No phase jump) ELSE IF ((Current-Previous)< - Threshold) THEN
Sumphase = Sumphase + 360 2 5 0O(k) = Current + 360 Previous = 0O(k) (-360 phase jump) E~SE
Sumphase = Sumphase - 360 ~o(k) = Current- 360 3 0 Previous = ~o(k) (+360 phase jump~
END ~? END DO
9~4i8 The array of unwrapped phase points Oo(k)~ k~ [1,Nl are employed in subsequent computation of the instantaneous frequency offset. Note that the unwrapped phase samples Oo(k) are just the values ~ k given by (13) where ~c~k may now 5 take on any value (including values outside the interval [-180, 180]).
2~ 8 ~5 CE00158R
Coml)utation of the in~tar tançD~ frequency of tlle fre~ tran~lated ~ienal~ froTn t~ time-derivative of the ~ha~e traiect~rxA
In the absence of any noise, quantization, or frequency instability, the instantaneous frequency error may be calculated from (13) using just the difference between any two phase sample~. In practice, all of these impairment~ result in measurement errors from employing just two ~amples.
1 0 Instead a least-squares linear fit to the phase trajectory i8 employed. The slope of such a least-squares linear fit curve is then proportional to the instantaneous frequency. From M.
Schwartz and L. Shaw Sienal Processine: Discrete SDectral Analvsis. Detection. and Estimation, McGraw-Hill, 1975, pp.
1 5 14-15, the slope (S) of the least-square linear fit line to a set of 2N+1 data points Oo(~N)~... 00(O),... Oo(N) i~ given by the following calculation performed by function 149:
2 0 n~0(n) n=-N
S= (14) N
~n2 n=-N
16 CE001i~
Assllme ~O(n) has the units of degrees. The instantaneous frequency error is therefore given by feITor (Hz) = d~/dt = S/360N L-270.833x103 (15 where L i8 the oversampling factor employed by the A/D
converter~ 107 and 109; i.e., the A/D sampling rate is L-270.833x103 samples/sec. It is important to note that the 1 0 frequency error is (a) independent of the input amplitude A and (b) can be calibrated directly in terms of Hz (see (15)).
Af~cer multiplication of the frequency error, given by (15), by the loop gain constant a in amplifier function 161 and 1 5 integrating this result in conventional integrator 1~3, the integrator output i8 converted to an analog voltage Vc (t) by D/A
converter 15~. The D/A converter 155 output analog voltage Vc (t) drives the VCXO 115, thereby closing the frequency control loop.
With proper selection of the loop gain constant ~ and knowledge of the VCXO Hztvolt gain constant, it is possible to lock the loop in just one time frame duration of the TDMA
signal since the frequency error given by (15) may be directly 2 5 calibrated in Hertz.
Generally ~peaking, the larger the number of phase sample points employed in the least-squares linear fit algorithm, the smaller the calculated frequency measurement 3 0 error. However, a large number of data points al~o requires a large amount of computation. To study the tradeoffbetween complexity and frequency mea~urement accuracy, a number of 2~ 8 computer simulation~ were run in which (a) the frequency offset was varied between ~500Hz in increments of 1.95Hz with one time slot of data at each frequency ~tep, (b) different values 5 of the decimation factor (M) and the number of measured data points employed in the least-squares linear fit algorithm were varied, keeping the product (2N+l)(M) _ 800 so that the same span of data points were used in each case.
A summary of these three simulations is given below.
Simulation M N 2N+1 A~8Q~
Fre~ue~y Error 1. 1 398 7g7 lHz 2. 20 19 39 11Hz 3. 40 9 19 27Hz Since a conventional 8 bit D/A converter 155 is employed in the frequency control loop and the stability of VCXO 115 is +2.5ppm 2 0 at 1 GHz implying an error step size of 19.5 Hz, the values of N
and M employed in simulation 2 will result in a frequency measurement error of approximately 1/2 LSB of the D/A 155 referenced to the VCXO 115 stability. The result of simulation 2 is shown in Fig. 3 in which Vc (t) is plotted as a function of 25 frequency off~et.
Note that once N i8 selected, the denominator in (14) can be pre-computed. For expediency, the denominator used in (14) can be selected to speed the frequency measurement process by selecting the denominator W to be ~ power of 2, i.e., let S'_~,n2S/W .
n=-N
Using the above-defined constants in (15), the expression for the instantaneous frequency error in terms of the measured parameter S' is ferror (Hz) = S/360-N L 270.8333 x 103 WV
N
~n2 1 5 n=-N
In the preferred embodiment, M=20 W=512 L=8 V=2048 N=19 N
~n2 =4940 n=-N
2 5 and thus f INST (Hz) = 63,875.S~ .
n=-N
n=-N
and thus 3 f error (Hz) = 63,875.S' .
L9 CEool58R2~9'-~8 A system which would employ a receiver employing the present invention i8 that which i9 specified for the Pan-European System in the aforementioned GSM docl~ments. A timing 5 diagram shown in Fig. 4 relates the burst TDMA transmissions to the frequency control of the present invention. In the exemplary transmission shown in Fig. 4 one transmission frame and part of a second transmission frame illustrated as blocks of infurmation transmission conveyed as frequency excursion~
1 0 about a center carrier frequency (fc). Each of these information blocks are conventional TDMA timeslots, each timeslot conveying information to a selected dif~erent receiver. In the preferred embodiment each frame lasts for a duration of time equal to 4.6ms. and each timeslot lasts for a time duration of 0.58ms.
In order to convey a frequency correction signal from the transmitter to the various receivers, the preferred embodiment employs a distinct carrier frequency shift during a predetermined timeslot, such as that shown at 401. (Thi~ carrier 2 0 shift has come to be called a "pure sine wave"(PSW)). The calTier shift is equal to 13MHz/192 _ 67.708kHz (within a tolerance range f t 0-05ppm = i 0.00338Hz). Regular TDMA communications may occur in subsequent timeslots, as indicated. The PSW need not be transmitted during each frame; the preferred embodiment 2 5 transmits the PSW at a rate of approximately 21 times per second.
The frequency control of the present invention processes the PSW as described previously. Upon receipt of the PSW, the 3 0 curve relating frequency offset (in Hertz) to the required frequency control signal (in millivolts) is interrogated based on the value of frequency offset calculated by the phase trajectory 2D CE00158R ;~0~ 8 computation. The single value of control signal (Vc (t)) corresponding to the frequency offset is determined. This singular control signal value is input to the VCX0 115 as a single step that places the VCX0 frequency within 0.03 parts per million 5 of the carrier frequency of the transmitted TDMA signal. Thi8 singular control signal value is maintained for the duration of the frames of signal until the next PSW is received. In this way, a rapid and accurate correction of frequency between the transmitted carrier frequency and the received frequency i8 1 0 obtained.
We Claim:
FREQUENCY CONTROL APPARATUS AND METHOD
Background of the Iru~e~tion This invention relates generally to digital radio sy6tems and more particularly to the rapid correction of frequency error in a digital radio receiver which receives communication messages from a transmitter transmitting the communication 15 messages in bursts.
Radio receivers often utilize a circuit to automatically correct for discrepancies in frequency between the carrier frequency of the signal to be received and the frequency of a 2 0 local oscillator used in a superheterodyne receiver. Thi~ local oscillator converts the carrier frequency and associated information carrying sidebands of the received signal to a convenient intermediate frequency. The typical frequency correction process is performed over a relatively long period of 2 5 time, assuming that the carrier frequency (of the received signal) is continuously present. The carrier frequency may be tracked by means of amplitude detectors, discriminators, or the like to generate a frequency control signal. Some systems may use a pilot signal modulated onto the carrier frequency to 3 0 provide a reference in the receiver to derive the frequency control signal. tSuch a pilot control is further described in U.S.
Patent No. 4,541,118).
2 ~ 6 8 The frequency control signal i9 subsequently applied to the local oscillator from its initial frequency to a frequency which converts the received carrier frequency into an intermediate 5 frequency optimally placed within the selectivity of the intermediate frequency amplification and filtering stages.
Digital receivers present a new set of problems to such conventional automatic frequency control networks. One 10 approach to a fast re~eiver frequency control for a digital receiver has been disclosed in Canadian Patent Application No.
2~oo3~688-5~Digital Automatic Frequency Control on Pure Sine Waves", filed on December 16, 1988 on behalf of Borth et al.
Usually, digital receivers must process the received carrier frequency signal in a linear fashion. Such linear processing allows amplitude variation of the received signal to create further errors in the detection of frequency offset.
Furthermore, digital communication is often accomplished 2 0 using burst transmission techniques such as time division multiple access (TDMA). Burst transmissions do not provide a continuously available carrier or carrier plus pilot which can be employed by conventional frequency control techniques.
2 5 Summarv of the Invention Therefore, it is one object of the present invention to rapidly correct for frequency errors between a received signal and the frequency to which the receiver is tuned in a digital radio 3 0 receiver.
,' '~
3 CE0015~O9468 It is another object of the present invention to rapidly correct a receiver oscillator for frequency errors between a nominal frequency and a desired frequency.
It is another object of the present invention is to correct the 5 frequency error between the received signal and the correct receiver local oscillator frequency within a single frame of received data.
It is a further object of the present invention to determine the frequency error from a correction burst and directly correct 10 the local oscillator frequency.
Brief Dessriptior~ hQDrawin~
Figure 1 is a block diagram of a TDMA receiver which may 15 utilize the present invention.
Figure 2 shows the unwrapping process for both an increasing phase trajectory and a decreasing phase trajectory.
Figure 3 shows the result of a computer simulation.
Figure 4 shows a timing diagram relating the burst TDMA
2 0 transmissions to the frequency control of the present invention.
Detailed Descril)tion of the Preferred Embodiment A block diagram of a TDMA receiver which may utilize the 2 5 present invention i8 shown in Fig. 1. Such a receiver may receive burst communication signals in a timeslot from a TDMA transmitter 103 received on an antenna 104 and applied to quadrature demodulator 105 as a signal x(t). The quadrature demodulator 105 produces two quadraturely related 3 0 downconverted signals which are applied to conventional analog to digital converters 107 and 109 which digitize each of the two quadrature related signals before applying the digitized 4 CE0015~R ~9(~68 quadrature signals to a digital signal processor (DSP) function 111. In the preferred embodiment, the DSP is realized employing a DSP56001 available from Motorola, Inc. (or 5 equivalent). The DSP function 111 recovers a data communication signal using conventional TDMA techniques in a communication signal recovery function 113. Such signal recovery functions include channel estimation, channel equalization, and data demodulation. Also included in the DSP
1 0 function 111 is the recovery of a frequency control signal which allows the rapid adjustment of a voltage controlled oscillator 115 in the quadrature demodulator 105 so that the TDMA
receiver can be quickly brought into a condition where there is virtually no frequency error between the carrier frequency of 1 5 x(t) and the frequency of voltage controlled oscillator 115.
Because speed of adjustment is important, it is an important feature of the present invention that the adjustment of the frequency of voltage controlled oscillator 115 be substantially accomplished in one step without oscillator frequency hunting 2 0 or successive approximations to a final oscillator frequency.
Assume that the transmitted signal corresponding to a frequency correction burst (or pure sine wave (PSW)) is given by x(t) = A cos[~ct I cl)ot] (1) where:
2 5 A is the amplitude of the signal, = 2~1fC is the carrier frequency of the signal in radians/sec;
c~O = 27~fo is the PSW baseband signal frequency in radians/sec = 27~ (67.708...kHz).
3 0 In the absence of multipath noise, Rayleigh fading, etc., the received signal is given by (1). In general, the voltage controlled oscillator 115 (VCXO) frequency without frequ~e~y ~ , :
CE00158R 2~39 control will be given by (C~c + ~3) where ~CD corresponds to a frequency of~set error (in radians/sec). Assume that the VCXO output signal is therefore given by VI(t) = COS[(CdC + ~)t] . (2) After phase shifting vI(t) by -90 in the phase shifter 117, the resulting signal is given by VQ(t) = sin[(coC + ~o~)t] . (3) The output of the in-phase (I) mixer 117 (in the absence of any DC of~sets) is given by IA(t) = x(t) VI(t) = A cos [cl~ct + ~ot] cos[(c~c + ~c~)t]
= V2 A ( cos[(a~O - ~c3)t] + cos [(2coct + coO + ~c~)t] } (4) which, after low-pass filtering with filter 121 having a bandwidth of approximately 4 coO radians/sec, becomes Ig(t) = V2 A COS[(CDo - ~)t] . (5) Similarly the output of the quadrature-phase (Q) mixer 123 (in the absence of any DC o~sets) is given by QB(t) = X(t) VQ(t) = A cos [~ct + ~ot] sin[(cDC + ~ )t]
= V2 A ( sin [(2coct + CDo + ~c3)t] - sin [(coO - ~ )t] ) (6) which after low-pass filtering by filter 125 becomes Qg(t) = -V2 A sin [(~ ~)t] . (7) 5~009'~6~3 The two A/D converters (107, 109) convert Ig(t) and Qg(t) into their quantized, sampled equivalents, Ig(k) and Qg(k~, respectively. In the absence of any DC offsets: Ig(k) = I~(k) 5 and Qg(k) = Qc(k) Note that direct rneasurement of the frequency of~set ~CD via a frequency counter which counts zero crossings of the signals given in (5) and (7) will yield a measurement uncertainty of 0 freq. uncertainty = 1/measurement per;od = l/slot period = V0.58 msec = + 1724 Hz (8) which corresponds to an uncertainty of i 1.9 ppm at 900 MHz.
In certain radio systems, for example, a digital radiotelephone system for use in Europe defined in GSM documents GSM
05.01/3.1.0, "Physical Layer on the Radio Path: General Description" and GSM 5.10/3, "Radio Sub-System Synchronization", the required stability of the VCXO 115 must 2 0 be vithin + 0.1 ppm of the received signal coming from the base station. Therefore, an approach to measuring frequency errors which does not count zero-crossings must instead be employed. The apparatus and method of the present invention:
2 5 (1) translates the I and Q signals given by (5) and (7) by 67.7 kHz to DC (i.e., removes the ~0 term present in (5) and (7));
(2) computes the phase of the translated signal at regularly spaced time intervals;
(3) constructs the phase trajectory from the phase samples;
and -7 CE00158R 2~34~i (4) computes an estimate of the instantaneous frequency of the frequency-translated signals from the time-derivative of the phase trajectory.
I2C O~set Com~e~j~
The DSP function 111 accepts the quantized VQ signals and processes both the I and the Q signal through a direct current 10 (DC) subtraction process 129. An uncorrected DC offset is a result of mismatched I and Q channels and local oscillator leakage. An estimate of the DC offset of IB (k) is made by DC
estimate function 131 which averages the input signal the output of which is subtracted from IB (k) in digital adder 133 1 5 thereby producing IC (k), the I quadrature signal with the DC
removed. Likewise, an estimate of the direct current offset is made by DC estimate function 135 of the QB (k) signal and subtracted in digital adder 137 to produce Qc(k), the Q
quadrature signal ~,vith the DC removed.
In order to implement the above frequency control of the present invention in a fractional fixed point general purpose digital signal processor (DSP) such as the DSP56001, it is necessary to perform ~everal unique steps. Imperfections in 2 5 an implementation of the quadrature demodulator 105 necessarily results in DC offsets at the output of the quadrature demodulator 105 which must be compensated for in the frequency control.
3 0 The presence of DC offsets at the output of the quadrature demodulator 105 can severely limit the performance of the frequency control. To see why this iS the case, let Ig(t) and 2~ 68 Qg(t) in (5) and (7) contain DC o~set terms of magnitude C and D, respectively. Then Ig'(k)= 1/2Acos[((do~ )k] +C (5a) Qg'(k) = -1/2 A sin [(cdo - ~co)k] +D (7a) Then af~er frequency translation by ej~O k a8 described below 1 0 ID'(k) = Re {[Sg(k) + C + jD] ejcoo k}
= Re {[1/2 A ej(~30 - ~o~)k + C + jD] ei~l)O k) = 1/2 A cos ~G3k + C cos cook - D sin co QD'(k) = Im {LSB(k) ~ c + jD] eiC~O k}
1 5 = Im t[V2 A ei(~l)O ~ ~I))k + C + jD] ei~l)O k) = 1/2 A sin ~c~k + C sin c~ok + D cos ~ok .
The presence of the two quadrature components at a frequency cl~O prevent further downsampling (or decimation) 2 0 without introducing aliasing errors and can contribute to a significant error in the computation of tan~l ( QE(k) / IE(k) ) in (13) below. Removal of the two undesired frequency components in (17) below can be accomplished by two methods:
(1) Low-pass filtering of ID'(k) and QD'(k) to remove the sin 2 5 ~I)ok and cos ~I)ok components; or (2) Removal of the DC components prior to frequency translation by eJ~DO k.
The first approach requires a low-pass filter with a 3 0 bandwidth les~ than fo/10= 6.7kHz, for example, and thus having an impulse response of duration greater than 5/6.7 kHz 9 CE001~O9~68 = 0.73 msec, for example, an impulse response duration greater than the duration of a single time slot. Clearly this is an unacceptable approach.
The second approach simply computes the average DC
value of the two quadrature branches separately and subtracts these values from the respective branch signals:
J
IC(k) = Ig(k) - l/J ~ IB(k) k=1 J
QC~k) = Qg(k) - VJ ~, Qg(k) 1 5 k=l F re~a~n~y~anslati~n bv 67.7 ~Iz.
2 0 A TDMA receiver compatible with GSM specifications such as those mentioned above utilizes a frequency colrection signal which iB transmitted as a carrier shif~ of exactly 67.7 kHz during one timeslot of the TDMA transmission. A frequency translation of the I quadrature channel signal and the Q
2 5 channel quadrature signal by 67.7 l~Iz accomplishes the following result. The signal is translated to 0 Hz i ~c~/2~
thereby reducing the information data rate and permitting further decimination.
This i~ accomplished by a quantized 67.7 KHz signal in 3 0 quadrature, as generated by 67.7 KHz oscillator 139.
2~9~68 Translation is accomplished by complex quadrature mixer 141, which is a complex multiplication realized in the DSP.
Let SC(k) _ IC(k) + jQc(k) = V2 Ae~ o ~ )k (g) where IC(k) and Qc(k) are given above. After multiplication of SC(k) by e)~l)ok, we obtain SD(k) - ID(k) + jQD(k) = 1/2 A(ei~3k) = lt2 A cos~c~k + j 1/2 A sin~k (10) 1 0 i.e., ID(k) = 1/2 A cos~cdk QD(k) = V2 A sin~3k . (11) In the preferred embodiment, the TDMA modulation i8 1 5 conventionally processed by the transmitter in a manner known as GMSK which limits the occupied bandwidth of the transmitted signal. Since ~co will generally be small compared to the information bandwidth of the transmitted GMSK signal with random data (i.e., a normal burst instead of a frequency 2 0 co~Tection burst), SD(k) may be decimated by a factor of M to reduce the ~ignal processing complexity required for frequency control without reducing control accuracy. The decimated signal SE(k) is obtained from SD(k) by the operation 2 5 SE(k) = SD(Mk) (12) which is accomplished in function 143, a conventional decimator. I'hus, the M-1 samples of SD(k) between valid samples of SE(k) are simply deleted.
11 CE00158~0 Con~ut~tiQn of the pha~ of the decimated, tran~la~ ~ienal.
In accordance with known trigonometric identities, a 5 calculation of the phase relationship between the VCXO 115 output signal and x(t) can be made employing the decimated and translated I and Q signal samples. Thi~ phase relationship may be expressed as ~ (k) for the k-th decimated sample period. The phase samples ~(k) are computed from 1 0 SE(k) via the operation ~(k) = tan~1 ( QE(k) / IE(k) ) = tan~l ( sin~ k / cos~c~k ) = tan~l ( tan(~cl)k) ) 1 5 = A~lk (13) where the time interval of k is understood to be the decimated period.
Three problems arise in calculating tan~l ( Q} (k) / IE(k) ) in 2 û (13):
l. Division of QE(k) by IE(k) on a fractional fixed point machine is limited to the case where IE(k) ~ QE(k) > 0.
2. tan-l (x) is an inverse transcendental function requi~ng either a Taylor series expansion or a table-lookup.
2 5 3. The valid range of tan~l ( QE(k) / IE(k) ) extends beyond ~180l after phase unwrapping and hence the tan~1 function must be scaled appropriately for a fixed point machine.
All three problems may be solved by:
3 û (a) Using a small (256 word) ROM arctangent table over the range [0,45]. To use the table, a short (8 instruction cycle) division of QE(k) by IE(k) is required to yield an 8 bit ROM table 12 OE00158R 2~ 68 address. Note that over the arctangent ROM table range, the inequality IE(k) > QE(k) 2 0 is preserved.
(b) Extending the range of the arctangent table to _180 by (i) preserving the 8igns of QE(k) and IE(k);
(ii) noting whether I IE(k) I > i QE(k) I or vice-versa; and (iii) using a precomputed table of trigonometric identities to compute the arctangent function in each of the 4 quadrants.
(c) Scaling the ROM table values to the largest possible 1 0 range of the unwrapped phase ~o(k)~ This range is given by +V= ~[maximum allowable frequency error (Hz) duration of 1 time slot (sec) 360 + 180]
For example, with a VCXO 115 stability of +2.5ppm at 1 GHz, the maximum allowable frequency error is i:2.5kHz and the 1 5 phase range of ~o(k) i8 given by +[(2.5kHz)(0.58 msec) 360 + 180]
= i702 = iV .
All phase values must therefore be scaled by V=702 to prevent overflow in the phase unwrapping algorithm.
Construction of the ~hase tra-ectorv from the Dhase sam~les.
Provided that the signs of IE(k~ and QE(k) are preserved, tan-l ( QE(k) / IE(k) ) is defined only over the internal [-7~ ], or 2 5 equivalently only over the interval [-180, 180]. Since absolute phase i~ not known in the frequency control process, any gi~en sample of SE(k) will yield a phase sample ~(k) within this range. For example, if the output of the phase computation of ~(k) is used directly a~ the phase trajectory, and if ~(1) = 179, 3 0 ~(2) = -179, ~(3) = -177,.. , then determination of the instantaneou~ frequency offset via time differentiation of the phase trajectory of ~(k) would yield ambiguous reslllts due to 2~ 68 the apparent -368 phase jump between ~(1) and ~(2). In fact, if the phase were "unwrapped" in thi~ example by allowing the phase samples to take on values outside the range i 180~, it becomes obvious that the phase samples are increasing by +2 5 every phase sample.
In order to unwrap the phase samples, a phase unwrap process 147 is employed in the DSP of the present invention.
Conceptually, the unwrapping process for both an increasing 1 0 phase trajectory and a decreasing phase trajectory is shown in Fig. 2 and is accomplished by the following steps which are performed by the DSP 111 of the present invention:
Initialization: a(k), k=1,..., N from Step 2 Previou~ = ~(1) Sumphase = 0 Threshold = 90 DOk=2toN
Current = 0(k) + Sumphase 2û IF( I(Current-Previous)l <Threshold)THEN
~o(k) = 0(k) Previous = ~o(k) (No phase jump) ELSE IF ((Current-Previous)< - Threshold) THEN
Sumphase = Sumphase + 360 2 5 0O(k) = Current + 360 Previous = 0O(k) (-360 phase jump) E~SE
Sumphase = Sumphase - 360 ~o(k) = Current- 360 3 0 Previous = ~o(k) (+360 phase jump~
END ~? END DO
9~4i8 The array of unwrapped phase points Oo(k)~ k~ [1,Nl are employed in subsequent computation of the instantaneous frequency offset. Note that the unwrapped phase samples Oo(k) are just the values ~ k given by (13) where ~c~k may now 5 take on any value (including values outside the interval [-180, 180]).
2~ 8 ~5 CE00158R
Coml)utation of the in~tar tançD~ frequency of tlle fre~ tran~lated ~ienal~ froTn t~ time-derivative of the ~ha~e traiect~rxA
In the absence of any noise, quantization, or frequency instability, the instantaneous frequency error may be calculated from (13) using just the difference between any two phase sample~. In practice, all of these impairment~ result in measurement errors from employing just two ~amples.
1 0 Instead a least-squares linear fit to the phase trajectory i8 employed. The slope of such a least-squares linear fit curve is then proportional to the instantaneous frequency. From M.
Schwartz and L. Shaw Sienal Processine: Discrete SDectral Analvsis. Detection. and Estimation, McGraw-Hill, 1975, pp.
1 5 14-15, the slope (S) of the least-square linear fit line to a set of 2N+1 data points Oo(~N)~... 00(O),... Oo(N) i~ given by the following calculation performed by function 149:
2 0 n~0(n) n=-N
S= (14) N
~n2 n=-N
16 CE001i~
Assllme ~O(n) has the units of degrees. The instantaneous frequency error is therefore given by feITor (Hz) = d~/dt = S/360N L-270.833x103 (15 where L i8 the oversampling factor employed by the A/D
converter~ 107 and 109; i.e., the A/D sampling rate is L-270.833x103 samples/sec. It is important to note that the 1 0 frequency error is (a) independent of the input amplitude A and (b) can be calibrated directly in terms of Hz (see (15)).
Af~cer multiplication of the frequency error, given by (15), by the loop gain constant a in amplifier function 161 and 1 5 integrating this result in conventional integrator 1~3, the integrator output i8 converted to an analog voltage Vc (t) by D/A
converter 15~. The D/A converter 155 output analog voltage Vc (t) drives the VCXO 115, thereby closing the frequency control loop.
With proper selection of the loop gain constant ~ and knowledge of the VCXO Hztvolt gain constant, it is possible to lock the loop in just one time frame duration of the TDMA
signal since the frequency error given by (15) may be directly 2 5 calibrated in Hertz.
Generally ~peaking, the larger the number of phase sample points employed in the least-squares linear fit algorithm, the smaller the calculated frequency measurement 3 0 error. However, a large number of data points al~o requires a large amount of computation. To study the tradeoffbetween complexity and frequency mea~urement accuracy, a number of 2~ 8 computer simulation~ were run in which (a) the frequency offset was varied between ~500Hz in increments of 1.95Hz with one time slot of data at each frequency ~tep, (b) different values 5 of the decimation factor (M) and the number of measured data points employed in the least-squares linear fit algorithm were varied, keeping the product (2N+l)(M) _ 800 so that the same span of data points were used in each case.
A summary of these three simulations is given below.
Simulation M N 2N+1 A~8Q~
Fre~ue~y Error 1. 1 398 7g7 lHz 2. 20 19 39 11Hz 3. 40 9 19 27Hz Since a conventional 8 bit D/A converter 155 is employed in the frequency control loop and the stability of VCXO 115 is +2.5ppm 2 0 at 1 GHz implying an error step size of 19.5 Hz, the values of N
and M employed in simulation 2 will result in a frequency measurement error of approximately 1/2 LSB of the D/A 155 referenced to the VCXO 115 stability. The result of simulation 2 is shown in Fig. 3 in which Vc (t) is plotted as a function of 25 frequency off~et.
Note that once N i8 selected, the denominator in (14) can be pre-computed. For expediency, the denominator used in (14) can be selected to speed the frequency measurement process by selecting the denominator W to be ~ power of 2, i.e., let S'_~,n2S/W .
n=-N
Using the above-defined constants in (15), the expression for the instantaneous frequency error in terms of the measured parameter S' is ferror (Hz) = S/360-N L 270.8333 x 103 WV
N
~n2 1 5 n=-N
In the preferred embodiment, M=20 W=512 L=8 V=2048 N=19 N
~n2 =4940 n=-N
2 5 and thus f INST (Hz) = 63,875.S~ .
n=-N
n=-N
and thus 3 f error (Hz) = 63,875.S' .
L9 CEool58R2~9'-~8 A system which would employ a receiver employing the present invention i8 that which i9 specified for the Pan-European System in the aforementioned GSM docl~ments. A timing 5 diagram shown in Fig. 4 relates the burst TDMA transmissions to the frequency control of the present invention. In the exemplary transmission shown in Fig. 4 one transmission frame and part of a second transmission frame illustrated as blocks of infurmation transmission conveyed as frequency excursion~
1 0 about a center carrier frequency (fc). Each of these information blocks are conventional TDMA timeslots, each timeslot conveying information to a selected dif~erent receiver. In the preferred embodiment each frame lasts for a duration of time equal to 4.6ms. and each timeslot lasts for a time duration of 0.58ms.
In order to convey a frequency correction signal from the transmitter to the various receivers, the preferred embodiment employs a distinct carrier frequency shift during a predetermined timeslot, such as that shown at 401. (Thi~ carrier 2 0 shift has come to be called a "pure sine wave"(PSW)). The calTier shift is equal to 13MHz/192 _ 67.708kHz (within a tolerance range f t 0-05ppm = i 0.00338Hz). Regular TDMA communications may occur in subsequent timeslots, as indicated. The PSW need not be transmitted during each frame; the preferred embodiment 2 5 transmits the PSW at a rate of approximately 21 times per second.
The frequency control of the present invention processes the PSW as described previously. Upon receipt of the PSW, the 3 0 curve relating frequency offset (in Hertz) to the required frequency control signal (in millivolts) is interrogated based on the value of frequency offset calculated by the phase trajectory 2D CE00158R ;~0~ 8 computation. The single value of control signal (Vc (t)) corresponding to the frequency offset is determined. This singular control signal value is input to the VCX0 115 as a single step that places the VCX0 frequency within 0.03 parts per million 5 of the carrier frequency of the transmitted TDMA signal. Thi8 singular control signal value is maintained for the duration of the frames of signal until the next PSW is received. In this way, a rapid and accurate correction of frequency between the transmitted carrier frequency and the received frequency i8 1 0 obtained.
We Claim:
Claims (10)
1. A frequency control apparatus for a burst-mode radio communications system employing a frequency correction signal transmitted as a burst to enable correction of frequency differences between the frequency of the radio carrier of a subsequent communication burst and the frequency of reception by a radio receiver, the frequency control apparatus characterized by:
means, including a variable frequency oscillator, for receiving a frequency correction signal burst and for processing a communication burst having a frequency difference between said variable frequency oscillator and the radio carrier frequency of said communication burst; and means, coupled to said means for receiving said frequency correction signal burst, for generating a singular control signal value and for applying said singular control signal value to said variable frequency oscillator to correct said frequency difference.
means, including a variable frequency oscillator, for receiving a frequency correction signal burst and for processing a communication burst having a frequency difference between said variable frequency oscillator and the radio carrier frequency of said communication burst; and means, coupled to said means for receiving said frequency correction signal burst, for generating a singular control signal value and for applying said singular control signal value to said variable frequency oscillator to correct said frequency difference.
2. A frequency control apparatus in accordance with claim 1 wherein said means for generating is further characterized by means for calculating at least one phase magnitude related to said frequency difference and wherein said means for processing a communication burst is further characterized by means for converting said frequency correction signal burst into I and Q quadrature frequency correction burst signals.
3 A frequency control apparatus in accordance with claim 2 wherein said means for generating is further characterized by:
means for translating the frequency of at least some of said I and Q quadrature frequency correction burst signals by a predetermined frequency;
means for sampling said I and Q quadrature frequency correction burst signals to generate I and Q correction signal samples;
means for generating a phase trajectory from at least some of said I and Q correction signal samples; and means for compensating a direct current offset.
means for translating the frequency of at least some of said I and Q quadrature frequency correction burst signals by a predetermined frequency;
means for sampling said I and Q quadrature frequency correction burst signals to generate I and Q correction signal samples;
means for generating a phase trajectory from at least some of said I and Q correction signal samples; and means for compensating a direct current offset.
4. A frequency control apparatus in accordance with claim 3 wherein said means for calculating at least one phase magnitude is further characterized by means for calculating the arctangent of the quotient of a Q correction signal sample divided by an I correction signal sample and wherein said means for generating a phase trajectory is further characterized by means for calculating a least squares linear fit curve of a first calculation of an arctangent to a second calculation of an arctangent, thereby producing a frequency difference versus control signal value relationship from which one singular control signal value for each frequency difference is derived.
5. A frequency control apparatus in accordance with claim 1 wherein said means for generating is further characterized by means for producing a frequency difference versus control signal value relationship from which one singular control signal value for each frequency difference is derived and wherein said means for generating and applying said singular control signal value to correct said frequency difference applies said singular control signal value within one frame of a plurality of communication bursts.
6. An automatic frequency adjustment circuit for a radio characterized by:
oscillator means for producing a signal at a predetermined nominal frequency, means for receiving a communications signal at substantially the frequency of the oscillator means, and a reference signal having a fixed frequency offset to the communications signal frequency, and digital means responsive to the reference signal to produce a singular correction signal to the oscillator means, to cause the oscillator means to produce a signal at the frequency of the communications signal.
oscillator means for producing a signal at a predetermined nominal frequency, means for receiving a communications signal at substantially the frequency of the oscillator means, and a reference signal having a fixed frequency offset to the communications signal frequency, and digital means responsive to the reference signal to produce a singular correction signal to the oscillator means, to cause the oscillator means to produce a signal at the frequency of the communications signal.
7. An automatic frequency adjustment circuit as set forth in claim 6 wherein the means for producing the singular correction signal is characterized by means for digitally generating a calibrated S-curve of a correction value as a function of the frequency offset and which is substantially insensitive to amplitude variations of the reference signal.
8. An automatic frequency adjustment circuit as set forth in claim 6 wherein the singular correction signal causes the oscillator means to produce a signal within one part per million of the frequency of the communications signal in one step within ten system frames.
9. A method of frequency control for a receiver in a burst-mode radio communications system employing a frequency correction signal transmitted as a burst to enable correction of frequency differences between the frequency of the radio carrier of a subsequent communication burst and the frequency of reception by the radio receiver, the method of frequency control characterized by the steps of:
receiving a frequency correction signal burst and processing a communication burst having a frequency difference between a variable frequency oscillator and the radio carrier frequency of said communication burst; and generating a singular control signal value in response to said receiving of said frequency correction signal burst and for applying said singular control signal value to said variable frequency oscillator to correct said frequency difference.
receiving a frequency correction signal burst and processing a communication burst having a frequency difference between a variable frequency oscillator and the radio carrier frequency of said communication burst; and generating a singular control signal value in response to said receiving of said frequency correction signal burst and for applying said singular control signal value to said variable frequency oscillator to correct said frequency difference.
10. A method in accordance with claim 9 wherein said processing a communication burst step is further characterized by the step of converting said frequency correction signal burst into I and Q quadrature frequency correction burst signals wherein said calculating at least one phase magnitude step is further characterized by the steps of calculating the arctangent of the quotient of a Q correction signal sample divided by an I
correction signal sample, and calculating a least squares linear fit curve of a first calculation of an arctangent to a second calculation of an arctangent, thereby producing a frequency difference versus control signal value relationship from which one singular control signal value for each frequency difference is derived; and wherein said generating step is further characterized by the steps of:
(a) calculating at least one phase magnitude related to said frequency difference, (b) translating the frequency of at least some of said I
and Q quadrature frequency correction burst signals by a predetermined frequency, (c) sampling said I and Q quadrature frequency correction burst signals to generate I and Q correction signal samples, (d) generating a phase trajectory from at least some of said I and Q correction signal samples, and (e) compensating a direct current offset.
correction signal sample, and calculating a least squares linear fit curve of a first calculation of an arctangent to a second calculation of an arctangent, thereby producing a frequency difference versus control signal value relationship from which one singular control signal value for each frequency difference is derived; and wherein said generating step is further characterized by the steps of:
(a) calculating at least one phase magnitude related to said frequency difference, (b) translating the frequency of at least some of said I
and Q quadrature frequency correction burst signals by a predetermined frequency, (c) sampling said I and Q quadrature frequency correction burst signals to generate I and Q correction signal samples, (d) generating a phase trajectory from at least some of said I and Q correction signal samples, and (e) compensating a direct current offset.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US331,557 | 1989-03-31 | ||
| US07/331,557 US4887050A (en) | 1989-03-31 | 1989-03-31 | Frequency control apparatus and method for a digital radio receiver |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CA2009468A1 CA2009468A1 (en) | 1990-09-30 |
| CA2009468C true CA2009468C (en) | 1994-02-01 |
Family
ID=23294456
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002009468A Expired - Fee Related CA2009468C (en) | 1989-03-31 | 1990-02-07 | Frequency control apparatus and method for a digital radio receiver |
Country Status (15)
| Country | Link |
|---|---|
| US (1) | US4887050A (en) |
| EP (1) | EP0389974B1 (en) |
| JP (1) | JP3275178B2 (en) |
| CN (1) | CN1016034B (en) |
| AR (1) | AR244924A1 (en) |
| AT (1) | ATE154481T1 (en) |
| AU (1) | AU5161190A (en) |
| CA (1) | CA2009468C (en) |
| DE (1) | DE69030892T2 (en) |
| DK (2) | DK0389974T3 (en) |
| FI (1) | FI905571A0 (en) |
| IE (1) | IE62289B1 (en) |
| MX (1) | MX167064B (en) |
| NO (1) | NO905055D0 (en) |
| WO (1) | WO1990012457A1 (en) |
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-
1989
- 1989-03-31 US US07/331,557 patent/US4887050A/en not_active Expired - Lifetime
-
1990
- 1990-02-07 CA CA002009468A patent/CA2009468C/en not_active Expired - Fee Related
- 1990-02-20 AU AU51611/90A patent/AU5161190A/en not_active Abandoned
- 1990-02-20 WO PCT/US1990/000822 patent/WO1990012457A1/en active Application Filing
- 1990-03-12 IE IE87190A patent/IE62289B1/en unknown
- 1990-03-13 AR AR90316367A patent/AR244924A1/en active
- 1990-03-13 JP JP06395190A patent/JP3275178B2/en not_active Expired - Lifetime
- 1990-03-13 MX MX019875A patent/MX167064B/en unknown
- 1990-03-23 DK DK90105506.1T patent/DK0389974T3/en active
- 1990-03-23 AT AT90105506T patent/ATE154481T1/en active
- 1990-03-23 DE DE69030892T patent/DE69030892T2/en not_active Expired - Fee Related
- 1990-03-23 EP EP90105506A patent/EP0389974B1/en not_active Expired - Lifetime
- 1990-03-30 CN CN90101796A patent/CN1016034B/en not_active Expired
- 1990-11-09 FI FI905571A patent/FI905571A0/en not_active IP Right Cessation
- 1990-11-22 NO NO905055A patent/NO905055D0/en unknown
- 1990-11-27 DK DK281790A patent/DK281790D0/en not_active Application Discontinuation
Also Published As
| Publication number | Publication date |
|---|---|
| NO905055L (en) | 1990-11-22 |
| JPH02288447A (en) | 1990-11-28 |
| CN1016034B (en) | 1992-03-25 |
| IE900871L (en) | 1990-09-30 |
| DK281790A (en) | 1990-11-27 |
| CN1046074A (en) | 1990-10-10 |
| AR244924A1 (en) | 1993-11-30 |
| FI905571A0 (en) | 1990-11-09 |
| ATE154481T1 (en) | 1997-06-15 |
| EP0389974A1 (en) | 1990-10-03 |
| IE62289B1 (en) | 1995-01-25 |
| DK281790D0 (en) | 1990-11-27 |
| MX167064B (en) | 1993-03-01 |
| CA2009468A1 (en) | 1990-09-30 |
| US4887050A (en) | 1989-12-12 |
| DE69030892D1 (en) | 1997-07-17 |
| DE69030892T2 (en) | 1998-01-02 |
| JP3275178B2 (en) | 2002-04-15 |
| AU5161190A (en) | 1990-11-05 |
| NO905055D0 (en) | 1990-11-22 |
| WO1990012457A1 (en) | 1990-10-18 |
| EP0389974B1 (en) | 1997-06-11 |
| DK0389974T3 (en) | 1997-12-29 |
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| EEER | Examination request | ||
| MKLA | Lapsed |