CA2054513C

CA2054513C - Constant frequency resonant dc/dc converter

Info

Publication number: CA2054513C
Application number: CA002054513A
Authority: CA
Inventors: Praveen K. Jain
Original assignee: Northern Telecom Ltd
Current assignee: Nortel Networks Ltd
Priority date: 1990-12-13
Filing date: 1991-10-30
Publication date: 1997-08-19
Anticipated expiration: 2011-10-30
Also published as: CA2054513A1; US5157593A; JP3326754B2; JPH05168252A

Abstract

A constant frequency full resonant mode dc/dc converter is provided, suitable for use in low voltage, high current dc power distribution systems. The converter comprises a full-bridge inverter, a resonant circuit, an output transformer, an output diode rectifier, a dc output filter, and a controlcircuit. The resonant circuit consists of a series branch and a parallel branch.The components of the resonant circuit are selected such that a sinusoidal voltage waveform is achieved across the parallel branch, and an overall lagging current is obtained at the output of the full-bridge inverter for a desired range of output voltage control and no-load to full-load operating conditions.

Description

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CONSTANT FREQUENCY RESONANT DC/DC CON~/kK l ~K

Field of the Invention This invention generally relates to resonant dc/dc converters.
More specifically, it is directed to pulse width modulation (PWM) resonant mode dc/dc converters with constant and very high operating frequency.

Ba~gloulld of the Il~v~l~lion In many advanced telecommunication systems, for example, such as Fiber World Products, high performance dc/dc power converters with low voltage, high current output are required. Some of the basic requirements for these power converters are:
a) high ~ffil~iency;
b) high power density;
c) high reliability;
d) fast dynamic response;
e) low mass densit,v;
fl low EMI (both conducted and radiated);
g) regulated output dc voltage with low ripple; and h) constant frequenc,v operation.

It is believed that the above features of the converters will 25 ~ignifil~ntly enh~nce overall ~elroll.,ance of the system. Consequently, there is likely to be a trend toward use of such converters in the future.

Various topologies of half-bridge resonant converters are studied and compared in IEEE Trans. on Power Electronics, Vol. 3, No. 2, April 1988, 30 pp. 174-182 "A Comparison of Half-Brldge Resonant Converter Topologies"
by R. Steigerwald. A half-bridge inverter is coupled with a series, a parallel or a series-parallel resonant circuit and their ~uelfullLlances are compared.

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In an article by the present inventor, "Performance Comparison of Pulse Width Modulated Resonant Mode dc/dc Converters for Space Applications", IEEE Industry Applications Society Conference Record, Oct.
1989, pp. 1106-1114, full-bridge resonant converters are described.

PWM series-parallel resonant converters are further studied in "Fixed Frequency PWM Series-Parallel Resonant Converter", by A.K.S. Bhat, IEEE Industry Applications Society Conference Record, Oct. 1989, pp. 111 1121.

U.S. Patent 4,857,822, issued August 15, 1989 in the name of Tabisz et al, describes a zero-voltage-switched multi-resonant converter in which a multi-resonant switch forms three different resonant circuits for improved performance at higher freq1len~i~s U.S. Patent 4,833,584, issued May 23, 1989 in the name of Divan, on the other hand, teaches a high frequency quasi-resonant current mode static power conv~llel in which the output voltage is controlled by varying the operating frequency. It is therefore not suitable for the application where 20 frequency locking/synchronization is required.

In the above articles, parallel resonant and series-parallel resonant circuits contain only parallel capacitors as their parallel resonant component.
Objects of the Invention It is therefore an object of the present invention to provide resonant dc/dc CO11V~

2 ~ 3 ~ cij It is another object of the present invention to provide pulse width modulated (PWM) resonant dc/dc converters by using phase-shift control within the inverter.

It is a further object of the present invention to provide PWM
resonant dc/dc converters including a series and a parallel resonant circuit.

It is a further object of the present invention to provide a resonant dc/dc converter with near zero turn-on, turn-off, and snubber losses 10 while operating at constant and very high frequency.

Summary of the Invention Briefly stated, the present invention is directed to a constant 15 frequency full resonant mode dc/dc converter which comprises a full bridge inverter, a resonant circuit means connected to the full bridge inverter, a transformer connecte-3 to the resonant circuit means, and a rectifying circuit means col necte~ to the` transformer. The resonant circuit means includes a parallel resonant branch which has a parallel inductor and a parallel capacitor,20 and a series resonant branch which has a series inductor.

Brief Des.~ Iion of the Dld~. ings For a more complete underst~n~ing of the present invention 25 and for further objects and advantages thereof, reference may now be made to the following description, taken in conjunction with the accompanying drawings, in which:
Figure 1 is a diagrammatic illustration of a prior art dc/dc converter;
Figure 2 is a diagrammatic illustration of a resonant dc/dc converter according to one embodiment of the present invention;

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Figures 3(a) and 3(b) are graphs showing various waveforms of signals on a corresponding time scale;
Figure 4 is a diagrammatic illustration of a resonsant dc/dc converter according to a further embodiment of the present invention; and Figures 5 and 6 are graphs showing various wavefo~ s of signals according to the embodiment shown in Figure 4.

Detailed Descl;~tioll of the P~ d Embodiments o A large number of resonant mode dc/dc converter topologies have been reported in the literature. However, it has been shown, for example, in the aforementioned articles that the series-parallel resonant converter is most suitable for applications with low voltage, high current regulated dc outputs.

Figure 1 illustrates diagrammatically such a known dc/dc converter circuit. In the figure, a dc voltage source 2 supplies the power to a full-bridge in~rerter 4 whose output, voltage Vs and current Is is fed to a resonant circuit means 6. The resonant circuit means consists of a series resonant branch 8 and.a parallel resonant branch 10. A transformer 12 is connected to the resonant circuit means. The secondary windings of the transformer is provided with a full-wave rectifying circuit means 14 which feeds its output to a load 16.

It has been well known that the full-bridge inverter contains switching elements, e.g. FETs Sl, S2, S3 and S4 and their parallel or anti-parallel diodes. Snubber capacitors and resistors are also provided across theirrespective switching elements. The full-wave rectifying circuit means includes rectifying diodes and a filter consisting of a capacitor and an inductor.

In the circuit shown, the output voltage of the converter can be regulated by varying either (a) switching frequency of the converter with respect to the resonant frequency at constant pulse width, or (b) pulse width ofthe converter at constant switching frequency. Between the two, it has been 5 shown that the pulse width modulation (PWM) at constant switching frequency performs better than the variable frequency method, particularly in systems with frequency locking requirement. However, the PWM method also has the following disadvantages:
(1) turn-on losses (for either switches S1 and S4 or switches S2 and S3) and turn-o~f losses (for either switches S2 and S3 or switches S1 and S4) are present which are dnmin~nt at higher o~elatil~g freq~enri~si (2) lossy snubbers are required across switches S1 through S4.
These losses increase as the operating frequency of the converter is increased;

(3) either pair of switches S1 and S4 or S2 and S3 experience high`di/dt and dv/dt switching stresses; and (4) fast recovery diodes are required across either pair of switches S2 and S3 or Sl and S4~
", All the above disadvantages of the pulse width modulation limit the operation of the converter at m~dillm freql~enries (up to approximAtely 40 kHz).

Referring to Figure 2, one of the plefelled embodiments is diagrAmmAtically illustrated. The full-bridge inverter 22 contains four switches (Sl to S4), four diodes (D1 to D4) and four snubber capacitors (C1 to C4).
In this embo~limen~, high frequency MOSFET switches are employed for S1 to S4 and no external diode is required across the switch as body diodes of the 30 MOSFETs serve the function for D~ to D4. The function of this bridge is to , : .

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convert dc input voltage to high frequency quasi-square waveform voltage Vs at its own output.

The resonant circuit means 24 consists of a series branch 26 and a 5 parallel branch 28. The series and parallel branches of the resonant circuit means each comprise an inductor and a capacitor. Thus, a series capacitor Cs and a series inductor Ls are in series in the series branch and a parallel capacitor Cp and a parallel inductor Lp are in parallel in the parallel branch.
The components of the resonant circuit means are selected so that a close-to-o sinusoidal voltage waveform Vp across the parallel branch, and a near-sinusoidal current waveform Is with lagging power factor through the series branch is obtained.

The high frequency transformer 30 provides matching and 15 isolation for the output voltage of the converter. The rectifying circuit means 32 includes rectifying diodes ~RI and DR2~ and a filter consisting of an inductor Lo and capacitor CO so that an essentially ripple-free dc output voltage VO is applied to the load 34.

In another embodiment, the parallel inductor Lp of the parallel branch of the resonant circuit can be included as an integral part of the high frequency transformer 30 by adjusting the parameters of the transformer, such as air gap, turns, etc.

In this converter, a high frequency quasi-square voltage waveform is produced at the output of the full-bridge inverter 22 using phase-shift control within the inverter. The components of the series branch are tuned to the operating frequency so that this branch provides a zero impedance at the fundamental operating frequency. The components of the 30 parallel branch are off-tuned such that this branch provides an effective inductive impedance at the filnli~m~nt~l operating frequency.

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The application of the high frequency quasi-square voltage waveform across the combination of series and parallel branches results in a near-sinusoidal curren~ waveform with lagging power factor through the series branch and a close-to-sinusoidal voltage waveform across the parallel 5 branch. Voltage and current wave sh~ping obtained in this way achieves the following:
a) load-independent output voltage resulting in a narrow range of phase-shift control;
b) maximum voltage gain of the converter resulting in lower conduction losses;
c) turn-on and turn-off of the MOSFET switches under zero voltage, resulting in near zero switching losses;
d) loss-less snubbers across the switches;
e) elimination of Miller's effect, res-llting in lower gate drive current and losses at MOSFET switches;
f) mAxim1lm duty cycle (50%) for each rectifying diode of the output re~ ing circuit means, resulting in a lower value of the output filter in~1llctor; and low voltage stress across the output rectifying diode, res~lting in the use of schottky diode with lower ~o~ard voltage drop and hence co~duction losses.

The sinusoidal voltage waveform Vp across the parallel branch also appears at the primary winding of the transformer 30. The rectification , 25 achon of the output rectifying diodes converts this sinusoidal voltage ~vdve~",. into unidirectional output voltage waveform VDR at the output of ; ~ the output filter. The required voltage is now filtered by the output filter to provide a low ripple output dc voltage VO. The output voltage of the col~vl:lter is m~intained constant at a desired level against any variation in ~. ~
30 input voltage and output load by controlling the phase shift angle of the ~, ~

.
- :, converter. This converter performs at the operating frequency up to several hundred kHz.

A description of the operation of the dc/dc converter shown in 5 Figure 2 will now be given with reference to Figure 3.

For a steady-state cycle, there exists eight distinct intervals in which the converter of Figure 2 operates. For each interval, the operation of the converter is described below, where gating signals igl, ig2, ig3 and ig4 are10 control signals applied to the gates of the MOSFET switches S1, S2, S3 and S4 respectively.

Interval I:
During this interval, MOS~ET switches Sl and S~ are conducting 15 and carrying the positive resonant current Is~ Simultaneous conduction of S1 and S2 provides a zero voltage interval for the output voltage of the inverter (Vs)~ The zero voltage interval is needed to control power flow from the input to the output in order to regulate and/or vary the output DC voltage.

20 Interval II:
At the beginning of thls interval, gating signal igl applied to the gate of switch Sl has been removed. Switch Sl starts to turn-off, and the snubber capacitor Cl starts to charge. At the same time, because of the positiver~son~nt current Is~ the snubber capacitor C4 across switch S4 starts to discharge 25 to the reson~nt circuit. Once voltage across the snubber capacitor C4 reacheszero, the positive reso~ ~nt current forces anti-parallel diode D4 across switchS4 to conduct. Simultaneous conduction of diode D4 and switch S2 provides a negative voltage Vs at the output of the inverter. Since energy stored in the snubber capacitor is discharged to the resonant circuit, a larger value of the ~3~ 3 ~

snubber capacitor Cl is employed to provide a slow rise of the voltage across switch Sl. This ensures a close to zero turn-off loss in the switch.

Interval III:
At the beginning of this interval, gating signal ig4 has been applied at the gate of switch S4. The resonant current Is is now transferred to switch S4 from anti-parallel diode D4. Since diode D4 was conducting prior to the conduction of switch S4, turn-on of switch S4 is, therefore, obtained under zero voltage. This results in zero turn-on loss for the switch. During this 10 interval, switches S2 and S4 conduct and a negative voltage Vs appears at theoutput of the inverter. In this interval, power flows from input DC source to the resonant circuit.

Interval IV:
: 15 At the beginning of this interval, gating signal i~2 from the gate of switch S2 has been removed. Switch S2 starts to turn-off, and the snubber capacitor C2 starts to charge. At the same time, because of the negative resonant current Is~ the snubber capacitor C3 across switch S3, starts to discharge to the resonAnt circuit. Once voltage across snubber capacitor C3 reaches zero, the negative rçson~nt current forces anti-parallel diode D3 to conduct. Simultaneous conduction of switch S4 and anti-parallel diode D3 provides a zero voltage interval for the inverter output voltage Vs. A large value of snubber capacitor C2 across switch S2 provides a slow rise of voltage across switch S2, thereby ~liminA~ing turn-off loss in the switch.

Interval V:
At the beginning of this interval, gating signal ig3 has been applied at the gate of switch S3. The resonant current Is is now transferred to switch S3 from anti-parallel diode D3. Since diode D3 was conducting prior to the conduction of switch S3, turn-on of the switch is, therefore, obtained ~, ~

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under zero voltage which results in zero turn-on loss for the switch~ During this interval, switches S4 and S3 are conducting and providing a zero voltage interval for the inverter output voltage V

5 Interval VI:
At the beginning of this interval, gating signal ig4 from the gate of switch S4 has been removed. Switch S4 starts to turn-off and snubber capacitor C4 starts to charge. At the same time, because of negative resonant current Is~ the snubber capacitor Cl across switch S1 starts to discharge to theresonant circuit. Once voltage across snubber capacitor C1 reaches to zero, the negative resonant current forces anti-parallel diode D1 to conduct.
Simultaneous conduction of diode D1 and switch S3 provides a positive voltage Vs at the output of the inverter. A large value of snubber capacitor C4 provides a slow rise of voltage across switch S4, thereby eliminating turn-off 5 loss in the switch.

Interval VII:
At the beginning of this interval, signal ig1 has been applied at the gate of switch S1. The resonant current Is is now transferred to switch S1 20 from anti-parallel diode D1. Since diode D1 was conducting prior to the conduction of switch S1, turn-on of switch S1 is achieved under zero voltage, thereby elimin~ting turn-on loss for the switch. During this interval, switches S1 and S3 are con~l-cfing and providing a negative voltage Vs at the output of the inverter.
Interval VIII:
At the beginning of this interval, gating signal i~;3 from the gate of switch S3 has been removed. Switch S3 starts to turn-off, and snubber C3 starts to charge. At the same time, because of positive resonant current Is~ the30 snubber capacitor C2 starts to discharge to the resonant circuit. Once voltage across snubber capacitor C2 reaches zero, the positive resonant current forces anti-parallel diode D~ to conduct. Simultaneous conduction of switch S1 and diode D2 provides a zero voltage interval for the inverter output voltage Vs.
A large value of snubber capacitor C3 provides a slow rise of voltage across 5 switch S3, thereby eliminating turn-off loss in the switch.

At the position where ~ot = 2~, ~O is the filnd~mPntal operating frequency in radian/second, gating signal ig2 is applied at the gate of switch S2.
Switch S1 and S2 now conduct simultaneously and a new cycle begins.

The above description of the convelle operation reveals that the switches are always turned-on and turned-off under near zero voltage, thereby eliminating the switching losses. Further, the snubber capacitors always discharge to the resol ~nt circuit, which also eliminates the snubber 15 losses. As can be seen in the above from the above discussion with reference to the operation of the converter, all this is possible because of the lagging nature of the reson~nt current Is and large values of snubber ~ra~itors. The resonant circuit employed in this converter provides the desired type of output current Is.
The table below shows effi.i~ncies for the converter circuits of Figures 1 (prior art) and 2 (invention) as a function of operating frequency. Ascan be seen from this table, the P~ iency of the converter circuit of Figure 1 decreases as the ope,ating frequency increases, while the efficiency of the 25 convellel circuit of Figure 2 is independent of the operating frequency and is always higher than the other circuit. This gives the obvious advantages of the conve~ circuit of Figure 2 over the collv~l~er circuit of Figure 1 for high - ~ frequency operation in terms of higher power density and lower conversion losses.
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OPERATING EFFI~ NC:Y OPERATING
r~ PARAMETERS
PRIOR ART~ hv ~ n ~ lON
2S kHz 87 . 0% B7 .1% Oul.~u~.
Voltage - 5V
50 kHz 85.0~ 87.1% Oul~ul. Power ~ 500W
100 kHs 83 . o% 87 .1% Input Vo1t-~IgQ ~ 48V
250 XHZ 76.3% a7.1 500 k~z 67 . 29~ ~7 . 19 1000 kHz s5. o% 87 . 19~
Referring now to Figure 4, a further embodiment of the present invention is diagrammatically illustrated. In this figure, the resonant circuit 5 means 40 consists of a series branch 42 cont~ining a series inductor Ls and a parallel branch 44 cont~ining a combination of a parallel inductor Lp and a parallel capacitor Cp. The components of the parallel branch are s~lecte~l such that they are tuned to the operating frequency of the converter. However, the value of the series inductor Ls is chosen such that the reson~nt frequency of Ls10 and Cp is about 1.7 times the operating frequency of the converter. The series in~ ctor Ls has two function~: (a) it provides high impe~nces for harmonic components which are present in output voltage of the full-bridge inverter, and (b) it limits discharging from the parallel capacitor to the switches of thefull-bridge il~Vel Lel during zero voltage interval of the output voltage of the5 full-bridge il~veller. The parallel inductor Lp and parallel capacitor Cp of the parallel branch also have two functions: (a) they provide a near-sinusoidal stiff voltage source across the output transformer, and (b) they allow the circ~ tin~ current of the resonant circuit to be confined primarily within the parallel branch.

In this converter, a high frequency quasi-square voltage ~averorll~ is produced at the input of the resonant circuit means using phase-shift modulation within the inverter. This voltage contains f~m~l~ental 2 ~j 3 ~

and odd harmonic components. The series inductor Ls of the series resonant branch offers a low impedance for the flln~lAnl~n~l, and high impe~l~ncPs for the harmonic components of the output voltage. The parallel inductor Lp and parallel capacitor Cp, which are tuned to the operating frequency, provide 5 a high impedance for the f1ln~1Ament~l, and low impe-l~nces for the harmonic components of the output voltage. Applic~tion of the high frequency quasi-square voltage waveform across the combination of series and parallel branches results in a resonant current waveform which has near exporlenti~lly rising and falling edges with lagging power factor through the 10 series inductor and a near-sinusoidal voltage waveform cross the parallel branch. Voltage and current wave shaping obtained in this way achieves the following characteristics, in addition to some features discussed heretofor with respect to the embodiment shown in Figure 2:
a) circulating currents are cot fine-l within the parallel 5 branch, resulting in high convelle~ Pffi~iPncy from full-load to reduced-load; and b) turn~on and turn-off of the MOSFET switches under zero voltage over a wide range of phase-shift control, resulting in near zero switching losses.

It is therefore evident that the switches are always turned on and turned off under near zero voltage, thereby eliminating the switching losses.
Further, the cnllhbPr capacitors always discharge to the resonant circuit, whichalso elimin~tes the snubber losses. In achieving near zero switching and 25 snubber losses, shape of the output current waveform Is plays a major role.
The nature of current Is should be such that it must force the anti-parallel diode across the switch to conduct before the switch can be turned on. This will ensure discharging of the snubber capacitors to the resonant circuit, and ~ ~turn-on of the switches under zero voltage. As explained earlier, the ; ~30 components Lp and Cp of the parallel branch are tuned to the operating frequency. However, the value of the series inductor is chosen such that the ~:

2 i.~

resonant frequency of Ls and Cp is higher than the operating frequency of the converter. The resonant frequency of Ls and Cp has a signific~nt effect on the shape of resonant current waveform Is as shown in Figures 5 and 6. As can be seen from these figures, depending on the values of the resonant frequency, 5 the current Is may have multiple or single zero crossings in each half cycle.
Over a wide range of control of the pulse width angle (~), the current waveform with multiple zero crossings does not always ensure conduction of the anti-parallel diode before the switch across it is turned on. Therefore, theresonant frequency of Ls and Cp should be such that the resultant output 10 current waveform has only single zero crossing in each half cycle over a widerange of the angle ~ control. Normally the resonant frequency of Ls and Cp is chosen about 1.7 times the operating frequency of the converter.

Claims

1. A constant frequency full resonant mode dc/dc converter operating at an operating frequency comprising:
a full bridge inverter;
a resonant circuit means connected to said full bridge inverter;
a transformer connected to said resonant circuit means; and a rectifying circuit means connected to said transformer in that:
said resonant circuit means comprises a parallel resonant branch having a parallel inductor and a parallel capacitor, and a series resonant branch having a series inductor.

2. The constant frequency full resonant mode dc/dc converter according to claim 1, wherein said series resonant branch further includes a series capacitor, and said full bridge inverter comprises at least four semiconductor switches and at least four snubber capacitors.

3. The constant frequency full resonant mode dc/dc converter according to claim 2, wherein said snubber capacitors and said resonant circuit means are chosen as to their values with respect to said operating frequency so that a near-sinusoidal current waveform with a lagging power factor is applied through said series resonant branch and a close-to-sinusoidal voltage waveform is applied across said parallel resonant branch.

4. The constant frequency full resonant mode dc/dc converter according to claim 3, wherein said series resonant branch is tuned to an operating frequency and said parallel resonant branch is tuned to a frequency different from said operating frequency.

5. The constant frequency full resonant mode dc/dc converter according to claim 4, wherein said full bridge inverter further comprises at least four diodes and said semiconductor switches are MOSFETs.

6. The constant frequency full resonant mode dc/dc converter according to claim 2, wherein the parallel inductor of the parallel resonant branch is an integral part of the transformer.

7. The constant frequency full resonant mode dc/dc converter according to claim 3, wherein the parallel inductor of the parallel resonant branch is an integral part of the transformer.

8. The constant frequency full resonant mode dc/dc converter according to claim 4, wherein the parallel inductor of the parallel resonant branch is an integral part of the transformer.

9. The constant frequency full resonant mode dc/dc converter according to claim 5, wherein the parallel inductor of the parallel resonant branch is an integral part of the transformer.

10. The constant frequency full resonant mode dc/dc converter according to claim 1, wherein said full bridge inverter comprises at least four semiconductor switches and at least four snubber capacitors.

11. The constant frequency full resonant mode dc/dc converter according to claim 10, wherein said snubber capacitors and said resonant circuit means are chosen as to their values with respect to said operating frequency so that a resonant current waveform having near exponentially rising and falling edges with a lagging power factor is applied through said series inductor and a near-sinusoidal voltage waveform is applied across said parallel branch.

12. The constant frequency full resonant mode dc/dc converter according to claim 11, wherein said parallel resonant branch is tuned to said operating frequency and the value of said series inductor is chosen so that the resonant frequency of said series inductor and said parallel capacitor is about 1.7 times said operating frequency.

13. The constant frequency full resonant mode dc/dc converter according to claim 12, wherein said full bridge inverter further comprises at least four diodes and said semiconductor switches are MOSFETs.

14. The constant frequency full resonant mode dc/dc converter according to claim 10, wherein the parallel inductor of the parallel resonant branch is an integral part of the transformer.

15. The constant frequency full resonant mode dc/dc converter according to claim 11, wherein the parallel inductor of the parallel resonant branch is an integral part of the transformer.

16. The constant frequency full resonant mode dc/dc converter according to claim 12, wherein the parallel inductor of the parallel resonant branch is an integral part of the transformer.

17. The constant frequency full resonant mode dc/dc converter according to claim 13, wherein the parallel inductor of the parallel resonant branch is an integral part of the transformer.