CA2087566A1 - D. c. chopper regulating method and apparatus incorporating bilateral regulating voltage path - Google Patents
D. c. chopper regulating method and apparatus incorporating bilateral regulating voltage pathInfo
- Publication number
- CA2087566A1 CA2087566A1 CA002087566A CA2087566A CA2087566A1 CA 2087566 A1 CA2087566 A1 CA 2087566A1 CA 002087566 A CA002087566 A CA 002087566A CA 2087566 A CA2087566 A CA 2087566A CA 2087566 A1 CA2087566 A1 CA 2087566A1
- Authority
- CA
- Canada
- Prior art keywords
- regulator
- voltage
- signal
- chopped
- switch
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Abandoned
Links
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/337—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
- H02M3/3372—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration of the parallel type
- H02M3/3374—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration of the parallel type with preregulator, e.g. current injected push-pull
Abstract
D.C. CHOPPER REGULATING METHOD AND APPARATUS INCORPORATING
BILATERAL REGULATING VOLTAGE PATH
ABSTRACT OF THE DISCLOSURE
An unregulated D.C. input voltage (VIN) is processed by a high power chopper (34,36) to produce a first chopped signal having the same voltage as the input (VIN), and by a low power chopper (56,58) to produce a second chopped signal in synchronism with and having a lower voltage than the first chopped signal. The first and second chopped signals are algebraically combined by transformers (52,54) to produce an A.C. output signal which may be converted to D.C. (VOUT). The second chopped signal may be controlled to have the same polarity relative to the first chopped signal in which case the voltages add to increase the output voltage (VOUT), or to have the opposite polarity relative to the first chopped signal in which case the voltages subtract to decrease the output voltage (VOUT). The periods of addition and subtraction are alternated within the period of the first chopped signal. The ratio of addition and subtraction of the second chopped signal may be varied to adjust the ratio of output voltage to input voltage over a wide range. An energy storage inductor (62) is provided in the high power chopper path to average the voltage addition and subtraction. The high power chopper operation may be intermittently interrupted to limit input current flow during startup, and switches (102,104) provided to discharge energy stored in the inductor (62) into the transformers (52,54) and thereby contribute to the output voltage (VOUT).
BILATERAL REGULATING VOLTAGE PATH
ABSTRACT OF THE DISCLOSURE
An unregulated D.C. input voltage (VIN) is processed by a high power chopper (34,36) to produce a first chopped signal having the same voltage as the input (VIN), and by a low power chopper (56,58) to produce a second chopped signal in synchronism with and having a lower voltage than the first chopped signal. The first and second chopped signals are algebraically combined by transformers (52,54) to produce an A.C. output signal which may be converted to D.C. (VOUT). The second chopped signal may be controlled to have the same polarity relative to the first chopped signal in which case the voltages add to increase the output voltage (VOUT), or to have the opposite polarity relative to the first chopped signal in which case the voltages subtract to decrease the output voltage (VOUT). The periods of addition and subtraction are alternated within the period of the first chopped signal. The ratio of addition and subtraction of the second chopped signal may be varied to adjust the ratio of output voltage to input voltage over a wide range. An energy storage inductor (62) is provided in the high power chopper path to average the voltage addition and subtraction. The high power chopper operation may be intermittently interrupted to limit input current flow during startup, and switches (102,104) provided to discharge energy stored in the inductor (62) into the transformers (52,54) and thereby contribute to the output voltage (VOUT).
Description
2087~66 D.C. CHOPPER REGULATING MET~OD AND APPARATUS INCORPORATING
BILATERAL REGUL~TING VOLTAGE PATH
BACKGROUND OF THE INVENTION
Field of the Invention ~ he present invention generally relates to theregulation or conversion of an unregulated D.C. input voltage into a regulated D.C. or A.C. output voltage, and more specifically to a D.C. chopper regulator and regulat-ing method incorporating a bilateral regulating voltage path.
Descri~tion of the Related Art D.C. to D.C. regulators or converters are utili7ed in applications in which a precisely regulated D.C. voltage is required, but the available voltage from a primary D.C.
power supply such as a battery varies over a range which is unacceptably large. Many regulator topologies have been proposed in the past, including the basic forward ~onverter or "buck" regulator 10 illustrated in FIG. 1. An unregu-lated D.C. voltage VIN from a battery 12 is applied through a switch 14 and energy storage inductor 16 to a load 18.
The cathode of a diode 20 is connected to the junction of the switch 14 and inductor 16, with the anode of the diode 18 being connected to ground. A smoothing capacitor 22 is connected across the load 18. A control unit 24 periodi-cally closes and opens the switch 14 to "chop" the input voltage VI~ applied to the inductor 16.
When the switch 14 is closed, current flows from the battery 12 through the switch 14 and inductor 16 to the load 18 in the direction indicated by an arrow 26. The diode 20 is reverse biased and does not conduct. When the switch 14 is opened, the inductor 16 is disconnected from the battery 12. The back electromotive force (EMF) in the inductor 16 forward biases the diode 20 and allows current to continue to flow through the inductor 16 (the energy stored in the inductor 16 causes it to act as a current source) to the load 18 in the direction of the arrow 26, with the forward biased diode 20 completing a closed circuit path to ground.
The inductor 16 and diode 20 constitute a "flywheel"
arrangement which causes current to flow continuously from the inductor 16 into the load 18 in the direction of the arrow 26. The load 18 dissipates current continuously, regardless of whether the switch 14 is open or closed. The capacitor 22 charges to the time averaged value of voltage across the load 18, providing a smoothed or regulated D.C.
output voltage VOUT' The output voltage V0ur can be regulated by varying the duty cycle (proportion of time the switch 14 is closed during each cycle) of the switch 1~. This operation is known in the art as pulse width modulation (PWM). The output voltage VOUT will approach VI~ as the duty cycle is increased toward 100~, and will approach zero as the duty cycle approaches 0%. Regulation of the output voltage VO.J~
to a predetermined value is accomplished by sensing the actual value of VOUTI increasing the duty cycle if VOUI drops below the predetermined value, and decreasing the duty cycle if VOUT increases above the predetermined value. In this configuration, the output voltage V0u~ cannot exceed the input voltage VIN-The regulator 10 is disadvantageous in that the inductor 16 must accommodate the entire current flow through the load 18, and must necessarily be large and 2087~66 heavy. This is unacceptable in applications where space and weight must ~e strictly conserved.
The regulator 10 also 6uffers from inefficiency since the entire load current flows through the diode 20 when the switch 14 is open. The diode 20 introduces a voltage drop of approximately one volt into the circuit, which dissi-pates power equal to the voltage drop multiplied by the current flow. The dissipative power loss is inversely proportional to the output voltage, and can become exces-sive in low voltage applications.
FIG. 2 illustrates a regulator 30 including the buckregulator of FIG. 1 combined with a "chopper" in a "push-pull", or more accurately a "push-push" arrangement. Liké
elements are designated by the same reference numerals. In the regulator 30, the inductor 16 is connected to the center tap of a primary winding of a transformer 32, with the opposite ends of the primary winding being connected through chopper switches 34 and 36 to ground. The center tap of a secondary winding of the transformer 32 is grounded, with the opposite ends of the secondary winding being connected to the anodes of rectifier diodes 38 and 40. The cathodes of the diodes 38 and 40 are connected together across the capacitor 22 and ioad 18.
The switches 34 and 36 constitute a chopper, and are alternatively driven with typically 50% duty cycle. More specifically, the switch 34 is closed when the switch 36 is open and vice-versa. This causes current to be induced in the ~econdary winding of the transformer 32 in alternating directions, which is rectified and converted to pulsating D.C. by the diodes 38 and 40. The pulsating D.C. is converted to smoothed D.C. by the capacitor 22.
The regulator 32 is capable of producing an output voltage V0u~ which is higher than the input voltage VI~ due to the provision of the transformer 32. However, the entire load current must still flow through the inductor 16 .
.
208756~
and diode 20, and there are two switches in the input current path as opposed to one switch in the basic regula-tor 10. Typically, the switches 14, 34 and 36 are embodied as switching transi6tors, which introduce voltage drops into the circuit in a manner similar to diodes. This can result in excessive power loss At low input voltages. Most prior art converters have been unable to achieve efficien-cies in excess of 88%.
FIG. 3 illustrates a regulator 50 which is disclosed in U.S. patent no. 4,943,903, entitled "POWER SUPPLY IN
WHICH REGULATION IS ACHIEVED BY PROCESSING A SMALL PORTION
OF APPLIED POWER THROUGH A SWITCHING REGULATOR", issued July 24, 1990 to Gilbert I. Cardwell, Jr., who is also one of the present inventors. In the regulator 50, a first transformer 52 has a center-tapped primary winding includ-ing two sections 52a and 52b and a center-tapped secondary winding having two sections 52c and 52d. The center tap of the primary winding sections 52a and 52b is connected to the inductor 16, whereas the opposite ends of the sections 52a and 52b are connected through switches 56 and 58 respectively to ground. The center tap of the secondary winding sections 52c and 52d is connected to the battery 12.
A second transformer 54 has two primary windings 54a and 54b and a center tapped secondary winding which is connected to the diodes 38 and 40 in the same manner as in the regulator 30. One end of the winding 52a is connected to ground through the switch 34, whereas one end of the winding 52b is connected to ground through the switch 36.
The opposite ends of the windings 54a and 54b are connected to the opposite ends of the windings 52a and 52b respec-tively.
In operation, the switches 34,56 are driven alternat-ingly with the switches 36,58 at typically 50~ duty cycle.
The switch 14 is driven as described above with a variable . .
-:
duty cycle to regulate the output voltage VOUT~ This is accomplished by producing a voltage at the center tap of the first transformer 52 which increases as the duty cycle of the switch 14 increases.
Closing of the switch 34 or 36 causes voltage VIN to be applied across the series combination of windings 54a, 52C
or 54b,52d respectively. If the switch 14 i6 driven with 0% duty cycle, the voltage at the center tap of the primary winding sections 52a and 52b will be zero, and the voltage across the cections 52a and 52b will also be zero. ~ue to transformer action, the voltage across the secondary winding sections 52C and 52d will also be zero. The voltage across the windings 54a or 54b will be equal to ViN.
Assuming that the turns ratio of the transformer 54 is unity, V0U~ will be equal to VIN.
If the switch 14 is driven at a duty cycle approaching 100%, the voltage at the center tap of the primary winding sections 52a and 52b will be substantially equal to VIN.
Due to transformer action, a voltage will be induced across the winding sections 52c and 52d equal to VIN divided by the turns ratio of the transformer 52. Assuming an exemplary turns ratio of 5:1, the induced voltage across the winding sections 52c and 52d will be VIN/5. The voltage appearing across the primary winding section 54a or 54b is thereby equal to V~N (from the battery 12) plus VIN/5 (from the winding section 52c or 52d) or 1.2 x VIN.
The transformer 52 adds a fraction of the voltage produced by the buck regulator 10, which is continuously variable and increases as the duty cycle of the switch 14 increases, to the input voltage VIN. The regulator 50 is advantageous over the regulator 30 in that the main portion of the input current flows through the windings 54a,52c or 54b,52d which are inductive elements and not subject to dissipative loss. The input current flows through only one switch 34 or 36 in the main current path. Only a fraction .
. - ~. . . ~ . ~
- ~ ~
.
2087~66 of the input current (in this case a maximum of 17%) flows through the auxiliary or regulating current path including the regulator 10 and is thereby subject to dissipative losses as described above.
Due to its inherent construction, the regulator 50 is only capable of unilateral regulation. In other words, it i6 capable of adding a regulating voltage to the input voltage, but not subtracting a regulating voltage there-from. The turns ratio of the transformer 52 is selected such that the switch 14 will be driven at 50% duty cycle and the regulating voltage appearing across the winding section 52c or 52d will be one-half the maximum regulating voltage when the input and output voltages VIN and VWI are at their predetermined (nominal, average or center of range) design values. The turns ratio of the transformer 54 is selected such that, with the predetermined design value of VI~ plus one-half the maximum regulating voltage applied across the winding section 54a or 54b, the prede-termined design value of VO~ will appear across the load 18.
In other words, when VIN and VOUT are at the center of their ranges, the regulating voltage will be at the center of its range.
This design is able to achieve efficiencies of 95%, but only if the system has a small input voltage range (e.g. + 5%).
SUMMARY OF THE INVENTION
In a bilateral D.C. chopper voltage regulating method and apparatus embodying the present invention, an unregu-lated D.C. input voltage is processed by a high power chopper to produce a first chopped signal having the same voltage as the input, and by a low power chopper to produce a second or regulating chopped signal in synchronism with and having a lower voltage than the first chopped signal.
The first and second chopped signals are algebraically 2~87~66 combined by transformers to produce an A.C. output signal which may be converted to D.C.
The second chopped signal may be controlled to have the same polarity relative to the first chopped signal in which case the voltages add to ~ncrease the output voltage, or to have the opposite polarity relative to the first - chopped signal in which case the voltages subtract to decrease the output voltage. The second chopped signal will normally be controlled to add part of the time and subtract for the remainder of the time. In this way, the power converter can change the ratio of input voltage to output voltage over a wide range.
An energy storage inductor is provided in the high power chopper path to average the voltage addition and subtraction. The high power chopper operation may be intermittently interrupted to limit input current flow during startup, and switches are provided in the embodiment of FIG. 7 to allow interruption without causing dangerous voltage surges. In addition, the added switches of FIG. 7 discharge energy stored in the inductor into the output transformer and thereby contribute to the output voltage.
The present regulator improves over the prior art regulator illustrated in FIG. 3 in that the energy storage inductor can be smaller than in the prior art regulator for the same regulation range. This is because the present regulator provides bilateral regulation (both addition and subtraction modes), and thereby twice the range of regula-tion as the prior art regulator using corresponding circuit elements.
The turns ratio of the transformer which produces the regulating voltage may also be one-half that of the corre~ponding transformer in the prior art regulator, since twice the regulating range can be obtained with the same turns ratio in accordance with the present invention. This enables the size and weight of the transformer to be reduced over the prior art, or the designer may chose to use the same size and weight and achieve wider regulating range.
These and other features and advantages of the present invention will be apparent to those skilled in the art from the following detailed description, taken together with the accompanying drawings, in which like reference numerals refer to like parts.
DESCRIPTION OF THE DRAWINGS
FIG. 1 is an electrical schematic diagram illustrating a basic prior art voltage regulator or converter;
FIG. 2 is similar to FIG. 1, illustrating another prior art regulator;
15FIG. 3 is also similar to FIG. 2, illustrating a prior art regulator to which the present invention constitutes a direct improvement;
FIG. 4 is an electrical schematic diagram illustrating ~ a voltage regulator or converter embodying the present invention;
FIG. 5 is similar to FIG. 4, but illustrates the regulator as including transistor switches and anti-parallel diodes;
FIG. 6 is a timing diagram illustrating the operation of the regulator of FIG. 4; and FIG. 7 is also similar to FIG. 4, but illustrates the regulator as including a non-dissipative startup switch arrangement.
DETAILED DESCRIPTION OF THE INVENTION
A D.C. chopper voltage regulator or converter embody-ing the present invention is illustrated in FIG. 4 and generally designated as 60. As discussed above, the present regulator 60 is a direct improvement over the prior art regulator 50 illustrated in FIG. 3. In order to enable comparison and appreciation of the present improvements, like elements are designated in FIG. 4 by the same refer-ence numerals used in FIG. 3. It will be understood, however, that the manner in which the elements are inter-connected in accordance with the present invention causes the regulator 60 to operate in a fundamentally different manner than the regulator 50.
In the regulator 60, the buck regulator 10 is omitted, and the battery 12 is connected directly to the center tap of the primary winding sections 52a and 52b of the first transformer 52. An energy storage inductor 62 is connected between the battery 12 and the center tap of the secondary windinq sections 52c and 52d of the transformer 52. It will be understood that the inductor 16 of the prior art regulator 50 is part of the buck regulator 10 thereof, and is not included in the regulator 60. The inductor 62 of the present regulator 60 does not have a corresponding circuit element in the prior art regulator 50.
The switches 34, 36, 56 and 58 are controlled by a control unit 64 which senses the output voltage VOUT, input voltage VIN~ output current and/or input current, compares the sensed parameter with a command voltage Vc or other command value and controls regulation in accordance therewith. In some applications, the output voltage VOUT is able to vary in response to variations in the command voltage Vc. In this case, the output voltage VOUT is compared to the command voltage Vc for regulation. In other applications, the output voltage V0u~ is desired to be constant in the presence of a varying input voltage VIN. In this case, the output voltage is compared to a reference voltage. In yet other applications, the output current may be the controlling parameter which is to be regulated and the control law built into the control unit can accommodate these variations.
The switches 34 and 36 are typically driven alternat-~.
,~ ~
ingly with 50% duty cycle, and constitute a first chopper which periodically interrupts the input voltage VIN to produce a first chopped signal in the form of a square wave across the series combination of the winding sections 54a or 54b, 52c or 52d and inductor 62.
The switches 56 and 58 are driven in synchronism (like frequency, variable phase) with the switches 34 and 36, and constitute a second chopper which periodically interrupts the input voltage VIN to produce a second chopped signal in the form of a square wave across the primary winding section 52a or 52b respectively. The second chopped signal is transformed by the transformer 52 and appears across the winding section 52c or 52d as a third chopped signal which is also in synchronism with the first chopped signal, but has a smaller voltage and a polarity which can be the same or opposite relative to the polarity of the first chopped signal.
A simplified manner of viewing the second chopper is to disregard the signals across the primary winding sections 52a or 52b, and consider the switches 56 and 58 in combination with the transformer 52 as producing a second chopped signal across the winding section 52c or 52d. In this case, the second chopped signal corresponds to the third chopped signal described above.
In either case, the signal chopped by the switches 34 and 36 constitutes a main power signal for supply to the load 18. The signal chopped by the switches 56 and 58 constitutes a relatively low power regulating signal, and is algebraically combined with the main power signal by the transformer 52 to provide the regulated output voltage VO~T.
The polarities of the winding sections of the trans-formers 52 and 54 are indicated by the dot winding conven-tion. In the following description, the "opposite end" of a winding section is construed to mean the end of the winding section opposite the dot end thereof.
2087~66 The regulator 60 is operated in an addition mode by closing the switches 34,58 or 36,56 together, or in a subtraction mode by closing the switches 34,56 or 36,58 together. The operation will be described for the four possible cases as follows.
Case I - addition mode, switches 34,58 closed and switches 36,56 open.
Current flows from the battery 12 to ground through the inductor 62, winding sections 52c and 54a and switch 34. This current flows into the dot end of the secondary winding section 52c, causing induced current to flow out of the dot end of the primary winding section 52b to ground through the switch 58. Since the dot end of the section 52b is negative (ground) with respect to the opposite end (battery 12) of the section 52b, current flows from the battery 12 through the section 52b to ground through the switch 58. Current flows through the section 52c from the battery 12 to ground. This causes the current flow out of the battery 12 to be larger than the average value.
The center tap of the sections 52a and 52b (the opposite end of the section 52b) receives VIN from the battery 12. Thus, the dot end of the section 52b is negative relative to the opposite end. Due to the action of the transformer 52, the dot end of the secondary winding section 52c is negative relative to the opposite end thereof. The result is that a voltage is induced in the section 52c having a value equal to VIN divided by the turns ratio of the transformer 52. Assuming an exemplary turns ratio of 5:1, the voltage induced in the section 52c will be VIN/5 Th~s voltage has the same polarity as VI~. Thus, the voltage applied to the primary winding section 54a of the transformer 54 is equal to VIN ( from the battery 12) plus VI~/5 (across the secondary winding section 52c) or 1.2 X VI~ minus the voltage across the inductor 62. The voltage across the inductor 62 causes the inductor current to increase.
Case II - subtraction mode, switches 34,56 closed and switches 36,58 open.
Current flows from the battery 12 to ground through the inductor 62, winding sections 52c and 54a and switch 34. This current flows into the dot end of the secondary winding section 52c, urging induced current to flow from ground, through the switch 56, and out of the dot end of the primary winding section 52a into the battery 12.
Although the dot end of the section 52a is positive (battery 12) with respective to the opposite end (ground) of the section 52a, current flows through the section 52c from ground into the battery 12, opposite to the direction of applied voltage. This causes the current flow out of the battery 12 to be smaller than the average value.
The switch 56 connects the opposite end of the section 52a to ground. The center tap of the sections 52a and 52b (the dot end of the section 52a) receives VIN from the battery 12. Thus, the dot end of the section 52a is positive relative to the opposite end. Due to the action of the transformer 52, the dot end of the secondary winding section 52c is positive relative to the opposite end thereof. The result is that a voltage is induced in the section 52c having a value equal to VIN/5l and the opposite polarity relative to VIN. Thus, the voltage applied to the primary winding section 54a of the transformer 54 is equal to VIN (from the battery 12) minus VIN/5 (across the section 52c) or 0.8 x VIN plus the voltage across the inductor 62.
The voltage across the inductor causes the inductor current to decrease.
Case III - addition mode, switches 36, 56 closed and switches 34,58 open.
Current flows from the battery 12 to ground through the inductor 62, winding sections 52d and 54b and switch 36. This current flows out of the dot end of the secondary ; ~ :
winding section 52d, urging induced current to flow into the dot end of the primary winding section s2a to ground through the switch 56. Since the dot end of the section 52a is positive ~battery 12) with respective to the opposite end (ground) of the section 52a, current flows from the battery 12 through the section 52a to ground through the switch 56. Current flows through the section 52d from the battery 12 to ground. This causes the current flow out of the battery 12 to be larger than the average value, similar to Case I.
The center tap of the sections 52a and 52b (the dot end of the section 52a) receives VIN from the battery 12.
Thus, the dot end of the section 52a is positive relative to the opposite end. Due to the action of the transformer -52, the opposite end of the secondary winding section 52d is clamped negative relative to the dot end thereof. A
voltage is induced in the section 52d having a value equal to VIN/5~ and the same polarity as VIN. Thus, the voltage applied to the primary winding section 54b of the trans-former 54 is equal to 1.2 x VIN minus the voltage across theinductor 62. The voltage across the inductor 62 causes the inductor current to increase in the same manner as in Case I.
Case IV - subtraction mode, switches 36,58 closed and switches 34,56 open.
Current flows from the battery 12 to ground through the inductor 62, winding sections 52d and 54b and switch 36. This current flows out of the dot end of the section 52d, urging induced current to flow from ground, through the switch 58, and out of the opposite end of the primary winding section 52b into the battery 12. Although the dot end of the section 52b is negative (ground) with respective to the opposite end (battery 12) of the section 52b, current flows through the section 52d from ground to the battery 12. This causes the current flow out of the " " ., :. ~
:
20~7566 battery 12 to be smaller than the average value, similar to Case II.
The center tap of the sections 52a and 52b (the opposite end of the section 52b) receives VIN from the battery 12. Thus, the dot end of the section 52b is negative relative to the opposite end. Due to the action of the transformer 52, the dot end of the 6econdary winding section 52d is clamped negative relative to the opposite end thereof. The result is that a voltage is induced in the section s2d having a value equal to V~N/5~ and the opposite polarity relative to VI~. Thus, the voltage applied to the primary winding section 54b of the trans-former 54 is egual 0.8 x VIN plus the voltage across the inductor 62. The voltage across the inductor 62 causes the inductor current to decrease, similar to Case II.
As described above, the switches 34 and 36 are alternatingly driven with S0% duty cycle. During the times the switch 34 is closed, the switch 58 is closed and the switch 56 opened to produce addition mode (Case I), and subsequently the switch 56 is closed and the switch 58 opened to produce subtraction mode (Case II). Voltage requlation is performed by varying the relative duty cycles of the addition and subtraction modes. When the sensed output voltage VOUT (or other controlled parameter) drops below the predetermined design value, the control unit 64 causes the switch 58 (addition mode) to be closed for a relatively longer time than the switch 56 (subtraction mode), such that the regulator 60 operates in addition mode longer than in subtraction mode and the output voltage vOU~
increases. ~he duty cycle is increased as the amount of deviation of the sensed voltage from the predetermined design voltage increases to provide a higher correction for larger amounts of deviation.
When the sensed output voltage V0u~ (or input voltage 35 VI~;) increases above the predetermined design value, the . :
"
2087~66 control unit 64 causes the switch 58 (addition mode) to be closed for a relatively shorter time than the switch 56 (subtraction mode), such that the regulator 60 operates in subtraction mode longer than in addition mode. The operation is essentially similar during the times the switch 36 is closed, with the switch 56 being closed for addition mode (Case III) and the switch 58 being closed for subtraction mode (Case IV).
The voltage induced across the secondary winding of the transformer 54 is A.C. In applications where an A.C.
output voltage is desired, the diodes 38 and 40 and capacitor 22 are omitted and the output voltage is consti-tuted by this A.C. voltage. The diodes 38 and 40 rectify the A.C. voltage to produce a pulsating D.C. voltage, which is smoothed by the capacitor 22 to produce the smooth D.C.
output voltage VOUI' FIG. 4 further illustrates how switches 66 and 68 may be provided to enable alternative methods of driving the regulator 60. The switches 66 and 68 are connected across the winding sections 52a and 52b respectively, and con-trolled by the control unit 64 in a desired manner. The switch 66 and/or 68 can be closed to short out one or both of the sections 52a and 52b and enable a mode of operation in which neither addition nor subtraction occurs. The inductor loss terms are halved when this option can be used. It is further possible to open both switches 56 and 58 simultaneously to completely open the primary winding sections 52a and 52b.
In the regulator 60 it is necessary to close two switches simultaneously to produce an addition mode or a subtraction mode. FIG. 5 illustrates a regulator 70 in which the switches 34, 36, 56 and 58 are replaced by NPN
bipolar transistors 72, 74, 76 and 78 respectively. Diodes 80 and 82 are connected across the emitter and collector of the transistors 76 and 78 respectively in anti-parallel relation. This arrangement simplifies the control of the switchea since it i6 necessary to close only switch 72 or 74 to produce a subtraction mode. It will be noted that the bipolar transistors 76 and 78 can be replaced by metal-oxide-semiconductor field-effect transistors (MOSFETs) which inherently include anti-parallel diodes, in which case the diodes 80 and 82 can be omitted.
The operation of the regulator 70 will be described with reference also being made to the timing diagram of FIG. 6. The drive signals applied by a control unit 90 to the bases of the transistors 72, 74, 76 and 78 are desig-nated as D72, D74, D76 and D78 respectively. The voltages at the collectors of the transistors 72, 74, 76 and 78 are designated as V72, V74, V76 and V78 respectively. Further illustrated are the voltage v84 at the center tap 84 of the secondary winding of the transformer 52, the current I62 through the inductor 62, the current I86 through the center tap 86 of the primary winding of the transformer 52, the current I88 flowing out of the battery 12 and the current I76 flowing through the collector of the transistor 76.
Case,I - addition mode, D72 and D78 high.
The transistors 72 and 78 are turned on. Current induced from the secondary winding section 52c into the primary winding section 52b flows out of the dot end of the section 52b through the transistor 78 to ground. The dot end of the section 52b is connected to ground, causing a voltage with the same polarity as VIN to be induced in the ~ection 52c. This operation is equivalent to closing the switches 34 and 58 of the regulator 60.
Case II - subtraction mode, D72 high.
Subtraction mode is produced by turning on only the transistor 72. A voltage is induced across the section 52a such that the dot end is positive relative to the opposite end. Increasing current flow through the section 52c causes the induced voltage in the section 52a to increase :, ;-,: ` . ~
.
:
. ~
- ~ : -208756~
until the opposite end of the section S2a i8 one diode dropmore negative than ground. This forward biases the diode 80, enabling current induced from the section 52c to flow from ground through the diode 80 and section 52a into the battery 12.
This causes the dot end of the section S2c to be positive relative to the opposite end, such that the dot ends of the sections 52a and 52b are also positive relative to the opposite ends. Current flows into the dot end of the section 5~c, and must flow out of the dot end of the section 52a or into the opposite end of the section 52b.
Since the transistor 78 is turned off and the diode 82 is reverse biased, current cannot flow throuqh the section 52b.
The opposite end of the section 52a is effectively connected to ground through the diode 80, causing a voltage with the dot having the same polarity as VIN to be induced in the section 52c. This operation is equivalent to closing the switches 34 and 56 of the regulator 60, with the diode 80 performing the function of the switch 56.
Case III - addition mode, D74 and D76 high.
The transistors 74 and 76 are turned on. Current induced from the secondary winding section 52d into the primary winding section 52a flows from out of the opposite - 25 end of the winding 52a through the transistor 76 to ground.
The opposite end of the section 52a is connected to ground, causing a voltage with the same polarity as VIN to be induced in the section 52d. This operation is eguivalent to closing the switches 36 and 56 of the regulator 60.
Case IV - subtraction mode, D74 high.
Subtraction mode is produced by turning on only the transistor 74. A voltage is induced across the section 52b such that the dot end is negative relative to the opposite end. Increasing current flow through the section 52d causes the induced voltage to increase until the opposite , ~ , .
~ :;
end of the section 52b is one diode drop more negative than ground. This forward biases the diode 82, enabling current induced from the section 52d to flow from ground through the d~ode 82 and section 52b into the battery 12.
This causes the dot end of the section 52d to be negative relative to the opposite end, such that the dot ends of the sections 52a and 52b are also negative relative to the opposite ends. Current flows out of the dot end of the section 52d, and must flow out of the dot end of the section 52a or into the dot end of the section 52b. Since the transistor 76 i6 turned off and the diode 80 is reverse biased, current cannot flow through the section 52a.
The dot end of the section 52b is effectively connect-ed to ground through the diode 82, causing a voltage with the opposite polarity relative to VIN to be induced in the section 52d. This operation is equivalent to closing the switches 36 and 58 of the regulator 60, with the diode ~2 performing the function of the switch 58.
It will be noted that the collector voltages of the transistors 76 and 78 vary only slightly between the addition and subtraction modes. In addition mode the voltage is ground, whereas in subtraction mode the voltage is one diode drop (approximately one volt) negative of ground. This small difference eliminates one term of switching loss in the regulator 70.
The present regulator 60 improves over the prior art regulator 50 in that the energy storage inductor 62 need only be half as large as the inductor 16 in the regulator 50. This is because the present regulator 60 provides bilateral regulation (both addition and subtraction modes), and thereby twice the range of regulation as the regulator 50 using corresponding circuit elements. The smaller and lighter inductor 62 enables the regulator 60 to be used in applications in which size and weight constraints would preclude the use of the regulator 50.
: : ., . ', ' -2087~66 The turns ratio of the transformer 52 which produces the regulating voltage may also be one-half that of the corresponding transformer in the regulator 50, since twice the regulating range can be obtained with the same turns ratio in accordance with the present invention. This enables the 6ize and weight of the transformer 52 to be reduced over the prior art.
The present regulator 60 also eliminates the buck regulator 10 including the diode 20 which is an inherent element in the prior art regulator 50, thereby eliminating the dissipative loss associated therewith.
A problem inherent in the basic regulator 60 or 70 is that it is difficult to control the initial supply of input voltage VI~ . When the battery 12 i6 first connected to the regulator 60 or 70, zero voltage is reflected from the load 18 to the primary winding of the transformer 54. The entire input voltage VI~ is effectively applied across the inductor 16 and primary winding sections 54a and 54b which could cause the input current to rapidly increase to a level high enough to destroy the switches 34 and 36. The switches 34 and 36 must both be opened before this occurs in order to limit the input current. ~owever, the back EMF
in the inductor 62 is then applied across the switches 34 and 36 which is also high enough to cause severe damage FIG. 7 illustrates a regulator 100 in which additional elements are added to the basic regulator 60 to prevent excessive current flow during startup. Switches 102 and 104 are connected between the junction of 54a,34 and the battery 12, and between the junction of 54b,36 and the battery 12 respectively. Diodes 106 and 108 are connected in series with the switches 102 and 104 respectively, with the cathodes of the diodes 106 and 108 being connected to the battery 12. A startup control unit 110 is additionally provided which cooperates with the control unit 64 to control the switches 102 and 104.
~, .
2087~66 During startup, the switches 34, 36, 56 and 58 are alternatingly controlled to operate exclusively in subtrac-tion mode to reduce the input current flow. ~he switches 34, 36, 56 and 58 are closed for a length of time which is insufficient for the input current to increase to an excessive level. The diodes 106 and 108 are reverse biased during this operation, preventing current from flowing through the switches 102 and 104.
The switches 34, 36, 56 and 58 are then opened and the one of the switches 102 and 104 is closed. Assuming that the switch 102 is closed, the back EMF of the inductor 62 will cause current flow through the sections 52c and 54a, switch 102 and diode 106 in the same direction as the input current. This current is induced into the secondary winding of the transformer 54 and supplied to the load 18.
Power dissipation by the load 18 reduces the current flow to a level low enough that the switches 34, 36, 38 and 40 can be closed again to supply mor~ energy into the regula-tor 100 and increase the output voltage V0ul toward the predetermined design level. The operation is essentially similar with the switch 104 closed.
The operation of the switches 34, 36, 56 and 58 alternates with the operation of the switches 102 and 104 to supply a safe incremental amount of energy into the inductor 62, and then discharge energy from the inductor 62 into the load 18. The timing of closing and opening the switches 102 and 104 is variable over a large range.
Typically, the switch 102 will be closed when the switch 34 is opened, and the switch 104 will be closed with the switch 36 is opened. However, if the switches 102 and 104 are turned off slower than the switches 34, 36, 56 and 58, various reactive currents such as induced in the leakage inductance of the sections 54a and 54b will be discharged into the inductor 62 or load 18, thereby providing a non-dissipative snubbing function.
The arrangement of FIG. 7 is desirable in that substantially no power is dissipated in the regulator 100 during startup. All of the input power (excluding low order losses) is either supplied directly to the load 18, or stored in the inductor 62 and 6ubseguently supplied therefrom to the load 18.
While several illustrative embodiments of the inven-tion have been shown and described, numerous variations and alternate embodiments will occur to those skilled in the art, without departing from the spirit and 6cope of the invention. Accordingly, it is intended that the present invention not be limited 601ely to the 6pecifically described illustrative embodiments. Various modifications are contemplated and can be made without departing from the spirit and scope of the invention as defined by the appended claims.
... . ~
BILATERAL REGUL~TING VOLTAGE PATH
BACKGROUND OF THE INVENTION
Field of the Invention ~ he present invention generally relates to theregulation or conversion of an unregulated D.C. input voltage into a regulated D.C. or A.C. output voltage, and more specifically to a D.C. chopper regulator and regulat-ing method incorporating a bilateral regulating voltage path.
Descri~tion of the Related Art D.C. to D.C. regulators or converters are utili7ed in applications in which a precisely regulated D.C. voltage is required, but the available voltage from a primary D.C.
power supply such as a battery varies over a range which is unacceptably large. Many regulator topologies have been proposed in the past, including the basic forward ~onverter or "buck" regulator 10 illustrated in FIG. 1. An unregu-lated D.C. voltage VIN from a battery 12 is applied through a switch 14 and energy storage inductor 16 to a load 18.
The cathode of a diode 20 is connected to the junction of the switch 14 and inductor 16, with the anode of the diode 18 being connected to ground. A smoothing capacitor 22 is connected across the load 18. A control unit 24 periodi-cally closes and opens the switch 14 to "chop" the input voltage VI~ applied to the inductor 16.
When the switch 14 is closed, current flows from the battery 12 through the switch 14 and inductor 16 to the load 18 in the direction indicated by an arrow 26. The diode 20 is reverse biased and does not conduct. When the switch 14 is opened, the inductor 16 is disconnected from the battery 12. The back electromotive force (EMF) in the inductor 16 forward biases the diode 20 and allows current to continue to flow through the inductor 16 (the energy stored in the inductor 16 causes it to act as a current source) to the load 18 in the direction of the arrow 26, with the forward biased diode 20 completing a closed circuit path to ground.
The inductor 16 and diode 20 constitute a "flywheel"
arrangement which causes current to flow continuously from the inductor 16 into the load 18 in the direction of the arrow 26. The load 18 dissipates current continuously, regardless of whether the switch 14 is open or closed. The capacitor 22 charges to the time averaged value of voltage across the load 18, providing a smoothed or regulated D.C.
output voltage VOUT' The output voltage V0ur can be regulated by varying the duty cycle (proportion of time the switch 14 is closed during each cycle) of the switch 1~. This operation is known in the art as pulse width modulation (PWM). The output voltage VOUT will approach VI~ as the duty cycle is increased toward 100~, and will approach zero as the duty cycle approaches 0%. Regulation of the output voltage VO.J~
to a predetermined value is accomplished by sensing the actual value of VOUTI increasing the duty cycle if VOUI drops below the predetermined value, and decreasing the duty cycle if VOUT increases above the predetermined value. In this configuration, the output voltage V0u~ cannot exceed the input voltage VIN-The regulator 10 is disadvantageous in that the inductor 16 must accommodate the entire current flow through the load 18, and must necessarily be large and 2087~66 heavy. This is unacceptable in applications where space and weight must ~e strictly conserved.
The regulator 10 also 6uffers from inefficiency since the entire load current flows through the diode 20 when the switch 14 is open. The diode 20 introduces a voltage drop of approximately one volt into the circuit, which dissi-pates power equal to the voltage drop multiplied by the current flow. The dissipative power loss is inversely proportional to the output voltage, and can become exces-sive in low voltage applications.
FIG. 2 illustrates a regulator 30 including the buckregulator of FIG. 1 combined with a "chopper" in a "push-pull", or more accurately a "push-push" arrangement. Liké
elements are designated by the same reference numerals. In the regulator 30, the inductor 16 is connected to the center tap of a primary winding of a transformer 32, with the opposite ends of the primary winding being connected through chopper switches 34 and 36 to ground. The center tap of a secondary winding of the transformer 32 is grounded, with the opposite ends of the secondary winding being connected to the anodes of rectifier diodes 38 and 40. The cathodes of the diodes 38 and 40 are connected together across the capacitor 22 and ioad 18.
The switches 34 and 36 constitute a chopper, and are alternatively driven with typically 50% duty cycle. More specifically, the switch 34 is closed when the switch 36 is open and vice-versa. This causes current to be induced in the ~econdary winding of the transformer 32 in alternating directions, which is rectified and converted to pulsating D.C. by the diodes 38 and 40. The pulsating D.C. is converted to smoothed D.C. by the capacitor 22.
The regulator 32 is capable of producing an output voltage V0u~ which is higher than the input voltage VI~ due to the provision of the transformer 32. However, the entire load current must still flow through the inductor 16 .
.
208756~
and diode 20, and there are two switches in the input current path as opposed to one switch in the basic regula-tor 10. Typically, the switches 14, 34 and 36 are embodied as switching transi6tors, which introduce voltage drops into the circuit in a manner similar to diodes. This can result in excessive power loss At low input voltages. Most prior art converters have been unable to achieve efficien-cies in excess of 88%.
FIG. 3 illustrates a regulator 50 which is disclosed in U.S. patent no. 4,943,903, entitled "POWER SUPPLY IN
WHICH REGULATION IS ACHIEVED BY PROCESSING A SMALL PORTION
OF APPLIED POWER THROUGH A SWITCHING REGULATOR", issued July 24, 1990 to Gilbert I. Cardwell, Jr., who is also one of the present inventors. In the regulator 50, a first transformer 52 has a center-tapped primary winding includ-ing two sections 52a and 52b and a center-tapped secondary winding having two sections 52c and 52d. The center tap of the primary winding sections 52a and 52b is connected to the inductor 16, whereas the opposite ends of the sections 52a and 52b are connected through switches 56 and 58 respectively to ground. The center tap of the secondary winding sections 52c and 52d is connected to the battery 12.
A second transformer 54 has two primary windings 54a and 54b and a center tapped secondary winding which is connected to the diodes 38 and 40 in the same manner as in the regulator 30. One end of the winding 52a is connected to ground through the switch 34, whereas one end of the winding 52b is connected to ground through the switch 36.
The opposite ends of the windings 54a and 54b are connected to the opposite ends of the windings 52a and 52b respec-tively.
In operation, the switches 34,56 are driven alternat-ingly with the switches 36,58 at typically 50~ duty cycle.
The switch 14 is driven as described above with a variable . .
-:
duty cycle to regulate the output voltage VOUT~ This is accomplished by producing a voltage at the center tap of the first transformer 52 which increases as the duty cycle of the switch 14 increases.
Closing of the switch 34 or 36 causes voltage VIN to be applied across the series combination of windings 54a, 52C
or 54b,52d respectively. If the switch 14 i6 driven with 0% duty cycle, the voltage at the center tap of the primary winding sections 52a and 52b will be zero, and the voltage across the cections 52a and 52b will also be zero. ~ue to transformer action, the voltage across the secondary winding sections 52C and 52d will also be zero. The voltage across the windings 54a or 54b will be equal to ViN.
Assuming that the turns ratio of the transformer 54 is unity, V0U~ will be equal to VIN.
If the switch 14 is driven at a duty cycle approaching 100%, the voltage at the center tap of the primary winding sections 52a and 52b will be substantially equal to VIN.
Due to transformer action, a voltage will be induced across the winding sections 52c and 52d equal to VIN divided by the turns ratio of the transformer 52. Assuming an exemplary turns ratio of 5:1, the induced voltage across the winding sections 52c and 52d will be VIN/5. The voltage appearing across the primary winding section 54a or 54b is thereby equal to V~N (from the battery 12) plus VIN/5 (from the winding section 52c or 52d) or 1.2 x VIN.
The transformer 52 adds a fraction of the voltage produced by the buck regulator 10, which is continuously variable and increases as the duty cycle of the switch 14 increases, to the input voltage VIN. The regulator 50 is advantageous over the regulator 30 in that the main portion of the input current flows through the windings 54a,52c or 54b,52d which are inductive elements and not subject to dissipative loss. The input current flows through only one switch 34 or 36 in the main current path. Only a fraction .
. - ~. . . ~ . ~
- ~ ~
.
2087~66 of the input current (in this case a maximum of 17%) flows through the auxiliary or regulating current path including the regulator 10 and is thereby subject to dissipative losses as described above.
Due to its inherent construction, the regulator 50 is only capable of unilateral regulation. In other words, it i6 capable of adding a regulating voltage to the input voltage, but not subtracting a regulating voltage there-from. The turns ratio of the transformer 52 is selected such that the switch 14 will be driven at 50% duty cycle and the regulating voltage appearing across the winding section 52c or 52d will be one-half the maximum regulating voltage when the input and output voltages VIN and VWI are at their predetermined (nominal, average or center of range) design values. The turns ratio of the transformer 54 is selected such that, with the predetermined design value of VI~ plus one-half the maximum regulating voltage applied across the winding section 54a or 54b, the prede-termined design value of VO~ will appear across the load 18.
In other words, when VIN and VOUT are at the center of their ranges, the regulating voltage will be at the center of its range.
This design is able to achieve efficiencies of 95%, but only if the system has a small input voltage range (e.g. + 5%).
SUMMARY OF THE INVENTION
In a bilateral D.C. chopper voltage regulating method and apparatus embodying the present invention, an unregu-lated D.C. input voltage is processed by a high power chopper to produce a first chopped signal having the same voltage as the input, and by a low power chopper to produce a second or regulating chopped signal in synchronism with and having a lower voltage than the first chopped signal.
The first and second chopped signals are algebraically 2~87~66 combined by transformers to produce an A.C. output signal which may be converted to D.C.
The second chopped signal may be controlled to have the same polarity relative to the first chopped signal in which case the voltages add to ~ncrease the output voltage, or to have the opposite polarity relative to the first - chopped signal in which case the voltages subtract to decrease the output voltage. The second chopped signal will normally be controlled to add part of the time and subtract for the remainder of the time. In this way, the power converter can change the ratio of input voltage to output voltage over a wide range.
An energy storage inductor is provided in the high power chopper path to average the voltage addition and subtraction. The high power chopper operation may be intermittently interrupted to limit input current flow during startup, and switches are provided in the embodiment of FIG. 7 to allow interruption without causing dangerous voltage surges. In addition, the added switches of FIG. 7 discharge energy stored in the inductor into the output transformer and thereby contribute to the output voltage.
The present regulator improves over the prior art regulator illustrated in FIG. 3 in that the energy storage inductor can be smaller than in the prior art regulator for the same regulation range. This is because the present regulator provides bilateral regulation (both addition and subtraction modes), and thereby twice the range of regula-tion as the prior art regulator using corresponding circuit elements.
The turns ratio of the transformer which produces the regulating voltage may also be one-half that of the corre~ponding transformer in the prior art regulator, since twice the regulating range can be obtained with the same turns ratio in accordance with the present invention. This enables the size and weight of the transformer to be reduced over the prior art, or the designer may chose to use the same size and weight and achieve wider regulating range.
These and other features and advantages of the present invention will be apparent to those skilled in the art from the following detailed description, taken together with the accompanying drawings, in which like reference numerals refer to like parts.
DESCRIPTION OF THE DRAWINGS
FIG. 1 is an electrical schematic diagram illustrating a basic prior art voltage regulator or converter;
FIG. 2 is similar to FIG. 1, illustrating another prior art regulator;
15FIG. 3 is also similar to FIG. 2, illustrating a prior art regulator to which the present invention constitutes a direct improvement;
FIG. 4 is an electrical schematic diagram illustrating ~ a voltage regulator or converter embodying the present invention;
FIG. 5 is similar to FIG. 4, but illustrates the regulator as including transistor switches and anti-parallel diodes;
FIG. 6 is a timing diagram illustrating the operation of the regulator of FIG. 4; and FIG. 7 is also similar to FIG. 4, but illustrates the regulator as including a non-dissipative startup switch arrangement.
DETAILED DESCRIPTION OF THE INVENTION
A D.C. chopper voltage regulator or converter embody-ing the present invention is illustrated in FIG. 4 and generally designated as 60. As discussed above, the present regulator 60 is a direct improvement over the prior art regulator 50 illustrated in FIG. 3. In order to enable comparison and appreciation of the present improvements, like elements are designated in FIG. 4 by the same refer-ence numerals used in FIG. 3. It will be understood, however, that the manner in which the elements are inter-connected in accordance with the present invention causes the regulator 60 to operate in a fundamentally different manner than the regulator 50.
In the regulator 60, the buck regulator 10 is omitted, and the battery 12 is connected directly to the center tap of the primary winding sections 52a and 52b of the first transformer 52. An energy storage inductor 62 is connected between the battery 12 and the center tap of the secondary windinq sections 52c and 52d of the transformer 52. It will be understood that the inductor 16 of the prior art regulator 50 is part of the buck regulator 10 thereof, and is not included in the regulator 60. The inductor 62 of the present regulator 60 does not have a corresponding circuit element in the prior art regulator 50.
The switches 34, 36, 56 and 58 are controlled by a control unit 64 which senses the output voltage VOUT, input voltage VIN~ output current and/or input current, compares the sensed parameter with a command voltage Vc or other command value and controls regulation in accordance therewith. In some applications, the output voltage VOUT is able to vary in response to variations in the command voltage Vc. In this case, the output voltage VOUT is compared to the command voltage Vc for regulation. In other applications, the output voltage V0u~ is desired to be constant in the presence of a varying input voltage VIN. In this case, the output voltage is compared to a reference voltage. In yet other applications, the output current may be the controlling parameter which is to be regulated and the control law built into the control unit can accommodate these variations.
The switches 34 and 36 are typically driven alternat-~.
,~ ~
ingly with 50% duty cycle, and constitute a first chopper which periodically interrupts the input voltage VIN to produce a first chopped signal in the form of a square wave across the series combination of the winding sections 54a or 54b, 52c or 52d and inductor 62.
The switches 56 and 58 are driven in synchronism (like frequency, variable phase) with the switches 34 and 36, and constitute a second chopper which periodically interrupts the input voltage VIN to produce a second chopped signal in the form of a square wave across the primary winding section 52a or 52b respectively. The second chopped signal is transformed by the transformer 52 and appears across the winding section 52c or 52d as a third chopped signal which is also in synchronism with the first chopped signal, but has a smaller voltage and a polarity which can be the same or opposite relative to the polarity of the first chopped signal.
A simplified manner of viewing the second chopper is to disregard the signals across the primary winding sections 52a or 52b, and consider the switches 56 and 58 in combination with the transformer 52 as producing a second chopped signal across the winding section 52c or 52d. In this case, the second chopped signal corresponds to the third chopped signal described above.
In either case, the signal chopped by the switches 34 and 36 constitutes a main power signal for supply to the load 18. The signal chopped by the switches 56 and 58 constitutes a relatively low power regulating signal, and is algebraically combined with the main power signal by the transformer 52 to provide the regulated output voltage VO~T.
The polarities of the winding sections of the trans-formers 52 and 54 are indicated by the dot winding conven-tion. In the following description, the "opposite end" of a winding section is construed to mean the end of the winding section opposite the dot end thereof.
2087~66 The regulator 60 is operated in an addition mode by closing the switches 34,58 or 36,56 together, or in a subtraction mode by closing the switches 34,56 or 36,58 together. The operation will be described for the four possible cases as follows.
Case I - addition mode, switches 34,58 closed and switches 36,56 open.
Current flows from the battery 12 to ground through the inductor 62, winding sections 52c and 54a and switch 34. This current flows into the dot end of the secondary winding section 52c, causing induced current to flow out of the dot end of the primary winding section 52b to ground through the switch 58. Since the dot end of the section 52b is negative (ground) with respect to the opposite end (battery 12) of the section 52b, current flows from the battery 12 through the section 52b to ground through the switch 58. Current flows through the section 52c from the battery 12 to ground. This causes the current flow out of the battery 12 to be larger than the average value.
The center tap of the sections 52a and 52b (the opposite end of the section 52b) receives VIN from the battery 12. Thus, the dot end of the section 52b is negative relative to the opposite end. Due to the action of the transformer 52, the dot end of the secondary winding section 52c is negative relative to the opposite end thereof. The result is that a voltage is induced in the section 52c having a value equal to VIN divided by the turns ratio of the transformer 52. Assuming an exemplary turns ratio of 5:1, the voltage induced in the section 52c will be VIN/5 Th~s voltage has the same polarity as VI~. Thus, the voltage applied to the primary winding section 54a of the transformer 54 is equal to VIN ( from the battery 12) plus VI~/5 (across the secondary winding section 52c) or 1.2 X VI~ minus the voltage across the inductor 62. The voltage across the inductor 62 causes the inductor current to increase.
Case II - subtraction mode, switches 34,56 closed and switches 36,58 open.
Current flows from the battery 12 to ground through the inductor 62, winding sections 52c and 54a and switch 34. This current flows into the dot end of the secondary winding section 52c, urging induced current to flow from ground, through the switch 56, and out of the dot end of the primary winding section 52a into the battery 12.
Although the dot end of the section 52a is positive (battery 12) with respective to the opposite end (ground) of the section 52a, current flows through the section 52c from ground into the battery 12, opposite to the direction of applied voltage. This causes the current flow out of the battery 12 to be smaller than the average value.
The switch 56 connects the opposite end of the section 52a to ground. The center tap of the sections 52a and 52b (the dot end of the section 52a) receives VIN from the battery 12. Thus, the dot end of the section 52a is positive relative to the opposite end. Due to the action of the transformer 52, the dot end of the secondary winding section 52c is positive relative to the opposite end thereof. The result is that a voltage is induced in the section 52c having a value equal to VIN/5l and the opposite polarity relative to VIN. Thus, the voltage applied to the primary winding section 54a of the transformer 54 is equal to VIN (from the battery 12) minus VIN/5 (across the section 52c) or 0.8 x VIN plus the voltage across the inductor 62.
The voltage across the inductor causes the inductor current to decrease.
Case III - addition mode, switches 36, 56 closed and switches 34,58 open.
Current flows from the battery 12 to ground through the inductor 62, winding sections 52d and 54b and switch 36. This current flows out of the dot end of the secondary ; ~ :
winding section 52d, urging induced current to flow into the dot end of the primary winding section s2a to ground through the switch 56. Since the dot end of the section 52a is positive ~battery 12) with respective to the opposite end (ground) of the section 52a, current flows from the battery 12 through the section 52a to ground through the switch 56. Current flows through the section 52d from the battery 12 to ground. This causes the current flow out of the battery 12 to be larger than the average value, similar to Case I.
The center tap of the sections 52a and 52b (the dot end of the section 52a) receives VIN from the battery 12.
Thus, the dot end of the section 52a is positive relative to the opposite end. Due to the action of the transformer -52, the opposite end of the secondary winding section 52d is clamped negative relative to the dot end thereof. A
voltage is induced in the section 52d having a value equal to VIN/5~ and the same polarity as VIN. Thus, the voltage applied to the primary winding section 54b of the trans-former 54 is equal to 1.2 x VIN minus the voltage across theinductor 62. The voltage across the inductor 62 causes the inductor current to increase in the same manner as in Case I.
Case IV - subtraction mode, switches 36,58 closed and switches 34,56 open.
Current flows from the battery 12 to ground through the inductor 62, winding sections 52d and 54b and switch 36. This current flows out of the dot end of the section 52d, urging induced current to flow from ground, through the switch 58, and out of the opposite end of the primary winding section 52b into the battery 12. Although the dot end of the section 52b is negative (ground) with respective to the opposite end (battery 12) of the section 52b, current flows through the section 52d from ground to the battery 12. This causes the current flow out of the " " ., :. ~
:
20~7566 battery 12 to be smaller than the average value, similar to Case II.
The center tap of the sections 52a and 52b (the opposite end of the section 52b) receives VIN from the battery 12. Thus, the dot end of the section 52b is negative relative to the opposite end. Due to the action of the transformer 52, the dot end of the 6econdary winding section 52d is clamped negative relative to the opposite end thereof. The result is that a voltage is induced in the section s2d having a value equal to V~N/5~ and the opposite polarity relative to VI~. Thus, the voltage applied to the primary winding section 54b of the trans-former 54 is egual 0.8 x VIN plus the voltage across the inductor 62. The voltage across the inductor 62 causes the inductor current to decrease, similar to Case II.
As described above, the switches 34 and 36 are alternatingly driven with S0% duty cycle. During the times the switch 34 is closed, the switch 58 is closed and the switch 56 opened to produce addition mode (Case I), and subsequently the switch 56 is closed and the switch 58 opened to produce subtraction mode (Case II). Voltage requlation is performed by varying the relative duty cycles of the addition and subtraction modes. When the sensed output voltage VOUT (or other controlled parameter) drops below the predetermined design value, the control unit 64 causes the switch 58 (addition mode) to be closed for a relatively longer time than the switch 56 (subtraction mode), such that the regulator 60 operates in addition mode longer than in subtraction mode and the output voltage vOU~
increases. ~he duty cycle is increased as the amount of deviation of the sensed voltage from the predetermined design voltage increases to provide a higher correction for larger amounts of deviation.
When the sensed output voltage V0u~ (or input voltage 35 VI~;) increases above the predetermined design value, the . :
"
2087~66 control unit 64 causes the switch 58 (addition mode) to be closed for a relatively shorter time than the switch 56 (subtraction mode), such that the regulator 60 operates in subtraction mode longer than in addition mode. The operation is essentially similar during the times the switch 36 is closed, with the switch 56 being closed for addition mode (Case III) and the switch 58 being closed for subtraction mode (Case IV).
The voltage induced across the secondary winding of the transformer 54 is A.C. In applications where an A.C.
output voltage is desired, the diodes 38 and 40 and capacitor 22 are omitted and the output voltage is consti-tuted by this A.C. voltage. The diodes 38 and 40 rectify the A.C. voltage to produce a pulsating D.C. voltage, which is smoothed by the capacitor 22 to produce the smooth D.C.
output voltage VOUI' FIG. 4 further illustrates how switches 66 and 68 may be provided to enable alternative methods of driving the regulator 60. The switches 66 and 68 are connected across the winding sections 52a and 52b respectively, and con-trolled by the control unit 64 in a desired manner. The switch 66 and/or 68 can be closed to short out one or both of the sections 52a and 52b and enable a mode of operation in which neither addition nor subtraction occurs. The inductor loss terms are halved when this option can be used. It is further possible to open both switches 56 and 58 simultaneously to completely open the primary winding sections 52a and 52b.
In the regulator 60 it is necessary to close two switches simultaneously to produce an addition mode or a subtraction mode. FIG. 5 illustrates a regulator 70 in which the switches 34, 36, 56 and 58 are replaced by NPN
bipolar transistors 72, 74, 76 and 78 respectively. Diodes 80 and 82 are connected across the emitter and collector of the transistors 76 and 78 respectively in anti-parallel relation. This arrangement simplifies the control of the switchea since it i6 necessary to close only switch 72 or 74 to produce a subtraction mode. It will be noted that the bipolar transistors 76 and 78 can be replaced by metal-oxide-semiconductor field-effect transistors (MOSFETs) which inherently include anti-parallel diodes, in which case the diodes 80 and 82 can be omitted.
The operation of the regulator 70 will be described with reference also being made to the timing diagram of FIG. 6. The drive signals applied by a control unit 90 to the bases of the transistors 72, 74, 76 and 78 are desig-nated as D72, D74, D76 and D78 respectively. The voltages at the collectors of the transistors 72, 74, 76 and 78 are designated as V72, V74, V76 and V78 respectively. Further illustrated are the voltage v84 at the center tap 84 of the secondary winding of the transformer 52, the current I62 through the inductor 62, the current I86 through the center tap 86 of the primary winding of the transformer 52, the current I88 flowing out of the battery 12 and the current I76 flowing through the collector of the transistor 76.
Case,I - addition mode, D72 and D78 high.
The transistors 72 and 78 are turned on. Current induced from the secondary winding section 52c into the primary winding section 52b flows out of the dot end of the section 52b through the transistor 78 to ground. The dot end of the section 52b is connected to ground, causing a voltage with the same polarity as VIN to be induced in the ~ection 52c. This operation is equivalent to closing the switches 34 and 58 of the regulator 60.
Case II - subtraction mode, D72 high.
Subtraction mode is produced by turning on only the transistor 72. A voltage is induced across the section 52a such that the dot end is positive relative to the opposite end. Increasing current flow through the section 52c causes the induced voltage in the section 52a to increase :, ;-,: ` . ~
.
:
. ~
- ~ : -208756~
until the opposite end of the section S2a i8 one diode dropmore negative than ground. This forward biases the diode 80, enabling current induced from the section 52c to flow from ground through the diode 80 and section 52a into the battery 12.
This causes the dot end of the section S2c to be positive relative to the opposite end, such that the dot ends of the sections 52a and 52b are also positive relative to the opposite ends. Current flows into the dot end of the section 5~c, and must flow out of the dot end of the section 52a or into the opposite end of the section 52b.
Since the transistor 78 is turned off and the diode 82 is reverse biased, current cannot flow throuqh the section 52b.
The opposite end of the section 52a is effectively connected to ground through the diode 80, causing a voltage with the dot having the same polarity as VIN to be induced in the section 52c. This operation is equivalent to closing the switches 34 and 56 of the regulator 60, with the diode 80 performing the function of the switch 56.
Case III - addition mode, D74 and D76 high.
The transistors 74 and 76 are turned on. Current induced from the secondary winding section 52d into the primary winding section 52a flows from out of the opposite - 25 end of the winding 52a through the transistor 76 to ground.
The opposite end of the section 52a is connected to ground, causing a voltage with the same polarity as VIN to be induced in the section 52d. This operation is eguivalent to closing the switches 36 and 56 of the regulator 60.
Case IV - subtraction mode, D74 high.
Subtraction mode is produced by turning on only the transistor 74. A voltage is induced across the section 52b such that the dot end is negative relative to the opposite end. Increasing current flow through the section 52d causes the induced voltage to increase until the opposite , ~ , .
~ :;
end of the section 52b is one diode drop more negative than ground. This forward biases the diode 82, enabling current induced from the section 52d to flow from ground through the d~ode 82 and section 52b into the battery 12.
This causes the dot end of the section 52d to be negative relative to the opposite end, such that the dot ends of the sections 52a and 52b are also negative relative to the opposite ends. Current flows out of the dot end of the section 52d, and must flow out of the dot end of the section 52a or into the dot end of the section 52b. Since the transistor 76 i6 turned off and the diode 80 is reverse biased, current cannot flow through the section 52a.
The dot end of the section 52b is effectively connect-ed to ground through the diode 82, causing a voltage with the opposite polarity relative to VIN to be induced in the section 52d. This operation is equivalent to closing the switches 36 and 58 of the regulator 60, with the diode ~2 performing the function of the switch 58.
It will be noted that the collector voltages of the transistors 76 and 78 vary only slightly between the addition and subtraction modes. In addition mode the voltage is ground, whereas in subtraction mode the voltage is one diode drop (approximately one volt) negative of ground. This small difference eliminates one term of switching loss in the regulator 70.
The present regulator 60 improves over the prior art regulator 50 in that the energy storage inductor 62 need only be half as large as the inductor 16 in the regulator 50. This is because the present regulator 60 provides bilateral regulation (both addition and subtraction modes), and thereby twice the range of regulation as the regulator 50 using corresponding circuit elements. The smaller and lighter inductor 62 enables the regulator 60 to be used in applications in which size and weight constraints would preclude the use of the regulator 50.
: : ., . ', ' -2087~66 The turns ratio of the transformer 52 which produces the regulating voltage may also be one-half that of the corresponding transformer in the regulator 50, since twice the regulating range can be obtained with the same turns ratio in accordance with the present invention. This enables the 6ize and weight of the transformer 52 to be reduced over the prior art.
The present regulator 60 also eliminates the buck regulator 10 including the diode 20 which is an inherent element in the prior art regulator 50, thereby eliminating the dissipative loss associated therewith.
A problem inherent in the basic regulator 60 or 70 is that it is difficult to control the initial supply of input voltage VI~ . When the battery 12 i6 first connected to the regulator 60 or 70, zero voltage is reflected from the load 18 to the primary winding of the transformer 54. The entire input voltage VI~ is effectively applied across the inductor 16 and primary winding sections 54a and 54b which could cause the input current to rapidly increase to a level high enough to destroy the switches 34 and 36. The switches 34 and 36 must both be opened before this occurs in order to limit the input current. ~owever, the back EMF
in the inductor 62 is then applied across the switches 34 and 36 which is also high enough to cause severe damage FIG. 7 illustrates a regulator 100 in which additional elements are added to the basic regulator 60 to prevent excessive current flow during startup. Switches 102 and 104 are connected between the junction of 54a,34 and the battery 12, and between the junction of 54b,36 and the battery 12 respectively. Diodes 106 and 108 are connected in series with the switches 102 and 104 respectively, with the cathodes of the diodes 106 and 108 being connected to the battery 12. A startup control unit 110 is additionally provided which cooperates with the control unit 64 to control the switches 102 and 104.
~, .
2087~66 During startup, the switches 34, 36, 56 and 58 are alternatingly controlled to operate exclusively in subtrac-tion mode to reduce the input current flow. ~he switches 34, 36, 56 and 58 are closed for a length of time which is insufficient for the input current to increase to an excessive level. The diodes 106 and 108 are reverse biased during this operation, preventing current from flowing through the switches 102 and 104.
The switches 34, 36, 56 and 58 are then opened and the one of the switches 102 and 104 is closed. Assuming that the switch 102 is closed, the back EMF of the inductor 62 will cause current flow through the sections 52c and 54a, switch 102 and diode 106 in the same direction as the input current. This current is induced into the secondary winding of the transformer 54 and supplied to the load 18.
Power dissipation by the load 18 reduces the current flow to a level low enough that the switches 34, 36, 38 and 40 can be closed again to supply mor~ energy into the regula-tor 100 and increase the output voltage V0ul toward the predetermined design level. The operation is essentially similar with the switch 104 closed.
The operation of the switches 34, 36, 56 and 58 alternates with the operation of the switches 102 and 104 to supply a safe incremental amount of energy into the inductor 62, and then discharge energy from the inductor 62 into the load 18. The timing of closing and opening the switches 102 and 104 is variable over a large range.
Typically, the switch 102 will be closed when the switch 34 is opened, and the switch 104 will be closed with the switch 36 is opened. However, if the switches 102 and 104 are turned off slower than the switches 34, 36, 56 and 58, various reactive currents such as induced in the leakage inductance of the sections 54a and 54b will be discharged into the inductor 62 or load 18, thereby providing a non-dissipative snubbing function.
The arrangement of FIG. 7 is desirable in that substantially no power is dissipated in the regulator 100 during startup. All of the input power (excluding low order losses) is either supplied directly to the load 18, or stored in the inductor 62 and 6ubseguently supplied therefrom to the load 18.
While several illustrative embodiments of the inven-tion have been shown and described, numerous variations and alternate embodiments will occur to those skilled in the art, without departing from the spirit and 6cope of the invention. Accordingly, it is intended that the present invention not be limited 601ely to the 6pecifically described illustrative embodiments. Various modifications are contemplated and can be made without departing from the spirit and scope of the invention as defined by the appended claims.
... . ~
Claims (22)
1. A regulator for receiving an unregulated D.C.
input signal and producing a regulated output signal, comprising:
first chopper means for periodically interrupting the input signal to produce a first chopped signal:
second chopper means for periodically interrupt-ing the input signal to produce a second chopped signal which is substantially in synchronism with, and selectively has the same or opposite polarity relative to the first chopped signal;
combining means for algebraically combining the first and second chopped signals to produce the output signal; and control means for controlling the second chopper means to select said relative polarity of the second chopped signal and thereby regulate the output signal.
input signal and producing a regulated output signal, comprising:
first chopper means for periodically interrupting the input signal to produce a first chopped signal:
second chopper means for periodically interrupt-ing the input signal to produce a second chopped signal which is substantially in synchronism with, and selectively has the same or opposite polarity relative to the first chopped signal;
combining means for algebraically combining the first and second chopped signals to produce the output signal; and control means for controlling the second chopper means to select said relative polarity of the second chopped signal and thereby regulate the output signal.
2. A regulator as in claim 1, further comprising energy storage inductor means connected in series circuit with the first chopper means and the combining means.
3. A regulator as in claim 1, in which the second chopper means produces the second chopped signal with a smaller voltage than the first chopped signal.
4. A regulator as in claim 1, in which the control means further controls the second chopper means to vary the duty cycle of the second chopped signal and thereby regulate the output signal.
5. A regulator as in claim 1, in which the combining means comprises transformer means having primary winding means for combining the first and second chopped signals and secondary winding means for transforming said combined first and second chopped signals to produce the output signal as an A.C. signal.
6. A regulator as in claim 1, in which the combining means comprises transformer means for combining and transforming the first and second chopped signals to produce an A.C. voltage, rectifier means for rectifying the A.C. voltage to produce a pulsating D.C. voltage and filter means for smoothing the pulsating D.C. voltage to produce the output signal as a smooth D.C. signal.
7. A regulator as in claim 1, in which:
the combining means comprises transformer means having primary winding means and secondary winding means;
and the first and second chopper means are connected to apply the first and second chopped signals in series circuit to the primary winding means.
the combining means comprises transformer means having primary winding means and secondary winding means;
and the first and second chopper means are connected to apply the first and second chopped signals in series circuit to the primary winding means.
8. A regulator as in claim 7, further comprising energy storage inductor means connected in series circuit with the primary winding means.
9. A regulator as in claim 1, in which the control means further comprises startup means for intermittently interrupting operation of the first chopper means to limit current flow into the regulator and discharging energy stored in the inductor means into the primary winding of the transformer means during startup of the regulator.
10. A regulator as in claim 1, in which the first and second chopper means and the combining means have a symmetrical push-push configuration.
11. A regulator as in claim 1, in which:
the combining means comprises:
first transformer means including a center-tapped primary winding having first and second sections and a center-tapped secondary winding having first and second sections; and second transformer means including a center-tapped primary winding having first and second sections and a secondary winding;
the first chopper means comprises:
first switch means connected in series circuit with the first section of the secondary winding of the first transformer means and the first section of the primary winding of the second transformer means for receiving the input signal; and second switch means connected in series circuit with the second section of the secondary winding of the first transformer means and the second section of the primary winding of the second transformer means for receiving the input signal; and the second chopper means comprises:
third switch means connected in series circuit with the first section of the primary winding of the first transformer means for receiving the input signal;
and fourth switch means connected in series circuit with the second section of the primary winding of the first transformer means for receiving the input signal.
the combining means comprises:
first transformer means including a center-tapped primary winding having first and second sections and a center-tapped secondary winding having first and second sections; and second transformer means including a center-tapped primary winding having first and second sections and a secondary winding;
the first chopper means comprises:
first switch means connected in series circuit with the first section of the secondary winding of the first transformer means and the first section of the primary winding of the second transformer means for receiving the input signal; and second switch means connected in series circuit with the second section of the secondary winding of the first transformer means and the second section of the primary winding of the second transformer means for receiving the input signal; and the second chopper means comprises:
third switch means connected in series circuit with the first section of the primary winding of the first transformer means for receiving the input signal;
and fourth switch means connected in series circuit with the second section of the primary winding of the first transformer means for receiving the input signal.
12. A regulator as in claim 11, further comprising energy storage inductor means connected in series circuit with said center tap of the secondary winding of the first transformer means for receiving the input signal.
13. A regulator as in claim 12, in which the third and fourth switch means each comprise:
switching transistor means; and diode means connected across the switching transistor means in anti-parallel relation.
switching transistor means; and diode means connected across the switching transistor means in anti-parallel relation.
14. A regulator as in claim 12, in which the control means comprises means for:
periodically opening and closing the first switch means;
periodically opening and closing the second switch means substantially out of synchronism with the first switch means;
selectively opening and closing the third switch means substantially in synchronism with the first switch means for causing the second chopper means to produce the second chopped signal with the same polarity relative to the first chopped signal;
selectively opening and closing the third switch substantially in synchronism with the second switch means for causing the second chopper means to produce the second chopped signal with the opposite polarity relative to the first chopped signal;
selectively opening and closing the fourth switch means substantially in synchronism with the second switch means for causing the second chopper means to produce the second chopped signal with the same polarity relative to the first chopped signal; and selectively opening and closing the fourth switch substantially in synchronism with the first switch means for causing the second chopper means to produce the second chopped signal with the opposite polarity relative to the first chopped signal.
periodically opening and closing the first switch means;
periodically opening and closing the second switch means substantially out of synchronism with the first switch means;
selectively opening and closing the third switch means substantially in synchronism with the first switch means for causing the second chopper means to produce the second chopped signal with the same polarity relative to the first chopped signal;
selectively opening and closing the third switch substantially in synchronism with the second switch means for causing the second chopper means to produce the second chopped signal with the opposite polarity relative to the first chopped signal;
selectively opening and closing the fourth switch means substantially in synchronism with the second switch means for causing the second chopper means to produce the second chopped signal with the same polarity relative to the first chopped signal; and selectively opening and closing the fourth switch substantially in synchronism with the first switch means for causing the second chopper means to produce the second chopped signal with the opposite polarity relative to the first chopped signal.
15. A regulator as in claim 14, in which:
the control means further comprises startup means for intermittently opening both the first and second switch means substantially in synchronism with each other to limit current flow into the regulator during startup thereof; and the regulator further comprises startup switch means controlled by the startup means for discharging energy stored in the inductor means into the primary winding of the second transformer means when both the first and second switch means are open.
the control means further comprises startup means for intermittently opening both the first and second switch means substantially in synchronism with each other to limit current flow into the regulator during startup thereof; and the regulator further comprises startup switch means controlled by the startup means for discharging energy stored in the inductor means into the primary winding of the second transformer means when both the first and second switch means are open.
16. A regulator as in claim 15, in which:
the startup switch means comprises:
fifth switch means connected in series circuit with the inductor means, the first section of the secondary winding of the first transformer means and the first section of the primary winding of the second trans-former means; and sixth switch means connected in series circuit with the inductor means, the second section of the secondary winding of the first transformer means and the second section of the primary winding of the second transformer means; and the startup means comprises means for:
selectively closing the fifth switch means for discharging said energy stored in the inductor means into the first section of the primary winding of the second transformer means; and selectively closing the fifth switch means for discharging said energy stored in the inductor means into the second section of the primary winding of the second transformer means.
the startup switch means comprises:
fifth switch means connected in series circuit with the inductor means, the first section of the secondary winding of the first transformer means and the first section of the primary winding of the second trans-former means; and sixth switch means connected in series circuit with the inductor means, the second section of the secondary winding of the first transformer means and the second section of the primary winding of the second transformer means; and the startup means comprises means for:
selectively closing the fifth switch means for discharging said energy stored in the inductor means into the first section of the primary winding of the second transformer means; and selectively closing the fifth switch means for discharging said energy stored in the inductor means into the second section of the primary winding of the second transformer means.
17. A regulator as in claim 12, in which the control means further comprises fifth switch means for selectively short circuiting at least one of the first and second sections of the primary winding of the first transformer means.
18. A regulator as in claim 12, in which the first transformer means has a turns ratio which is predetermined such that the second chopped signal has a smaller voltage than the first chopped signal.
19. A regulator as in claim 1, in which the control means further comprises means for sensing the voltage of the output signal and controlling the second chopper means to select said relative polarity of the second chopped signal such as to cause said voltage to approach a prede-termined value.
20. A regulator as in claim 1, in which the control means further comprises means for sensing the current of the output signal and controlling the second chopper means to select said relative polarity of the second chopped signal such as to cause said current to approach a prede-termined value.
21. A method of converting an unregulated D.C. input voltage into a regulated output voltage, comprising the steps of:
(a) periodically interrupting the input voltage to produce a first chopped signal;
(b) periodically interrupting the input voltage to produce a second chopped signal which is substantially in synchronism with, and selectively has the same or opposite polarity relative to the first chopped signal;
(c) transforming the second chopped signal to produce a third chopped signal which has the same polarity as the second chopped signal and a smaller voltage than the first chopped signal;
(d) algebraically combining the first and third chopped signals to produce the output voltage:
(e) sensing the polarity of deviation of the output voltage from a predetermined value; and (f) selecting said relative polarity of the second chopped signal in accordance with said sensed polarity such that the output voltage approaches the predetermined value.
(a) periodically interrupting the input voltage to produce a first chopped signal;
(b) periodically interrupting the input voltage to produce a second chopped signal which is substantially in synchronism with, and selectively has the same or opposite polarity relative to the first chopped signal;
(c) transforming the second chopped signal to produce a third chopped signal which has the same polarity as the second chopped signal and a smaller voltage than the first chopped signal;
(d) algebraically combining the first and third chopped signals to produce the output voltage:
(e) sensing the polarity of deviation of the output voltage from a predetermined value; and (f) selecting said relative polarity of the second chopped signal in accordance with said sensed polarity such that the output voltage approaches the predetermined value.
22. A method as in claim 21, in which:
step (e) further comprises sensing the amount of deviation of the output voltage from the predetermined value; and step (f) further comprises increasing the duty cycle of the second chopped signal as said sensed amount of deviation increases and vice-versa.
step (e) further comprises sensing the amount of deviation of the output voltage from the predetermined value; and step (f) further comprises increasing the duty cycle of the second chopped signal as said sensed amount of deviation increases and vice-versa.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US845,074 | 1992-03-03 | ||
| US07/845,074 US5218522A (en) | 1992-03-03 | 1992-03-03 | D.C. chopper regulating method and apparatus incorporating bilateral regulating voltage path |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2087566A1 true CA2087566A1 (en) | 1993-09-04 |
Family
ID=25294336
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002087566A Abandoned CA2087566A1 (en) | 1992-03-03 | 1993-01-19 | D. c. chopper regulating method and apparatus incorporating bilateral regulating voltage path |
Country Status (5)
| Country | Link |
|---|---|
| US (1) | US5218522A (en) |
| EP (1) | EP0559009B1 (en) |
| JP (1) | JP3369621B2 (en) |
| CA (1) | CA2087566A1 (en) |
| DE (1) | DE69301040T2 (en) |
Families Citing this family (24)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US5517400A (en) * | 1992-02-03 | 1996-05-14 | Ulrich Schwan | Method for synchronizing converters of an inductive element for reversible energy transmission |
| GB9206020D0 (en) * | 1992-03-19 | 1992-04-29 | Astec Int Ltd | Transition resonant convertor |
| US5327337A (en) * | 1992-09-01 | 1994-07-05 | Broadcast Electronics, Inc. | Resonant push-pull switching power amplifier |
| US5295058A (en) * | 1992-10-13 | 1994-03-15 | Recoton Corporation | Universal DC to DC power converter |
| DE4328556C2 (en) * | 1993-08-25 | 1995-06-08 | Ant Nachrichtentech | Push-pull converter with pre-regulator |
| JP2767781B2 (en) * | 1993-09-17 | 1998-06-18 | 東光株式会社 | AC-DC converter |
| JPH0795774A (en) * | 1993-09-20 | 1995-04-07 | Asia Denshi Kogyo Kk | Low-noise converter |
| FR2713030B1 (en) * | 1993-11-24 | 1996-01-12 | Merlin Gerin | Uninterruptible feed through neutral through, comprising a double chopper-elevator. |
| US6297724B1 (en) | 1994-09-09 | 2001-10-02 | The Whitaker Corporation | Lighting control subsystem for use in system architecture for automated building |
| US5907482A (en) * | 1995-11-30 | 1999-05-25 | Toko, Inc. | Power supply control device |
| US7269034B2 (en) | 1997-01-24 | 2007-09-11 | Synqor, Inc. | High efficiency power converter |
| US6055162A (en) * | 1998-03-12 | 2000-04-25 | Northrop Grumman Corporation | Self-oscillating DC-DC converter apparatus and method especially adaptable for VHF operation |
| US5969955A (en) * | 1998-12-29 | 1999-10-19 | General Electric Company | Push-pull power converter with crowbar circuit for very fast output voltage turn-off |
| DE19943575A1 (en) * | 1999-09-13 | 2001-03-22 | Mannesmann Vdo Ag | DC converter |
| US6198640B1 (en) * | 2000-06-01 | 2001-03-06 | Hughes Electronics Corporation | Three-switch add/subtract DC to DC converter |
| US6836414B1 (en) | 2002-10-17 | 2004-12-28 | University Of Central Florida | PWM half-bridge converter with dual-equally adjustable control signal dead-time |
| US6806662B1 (en) * | 2003-05-28 | 2004-10-19 | The Boeing Company | Multiple mode universal power source utilizing a rotating machine |
| US7339809B2 (en) * | 2004-11-30 | 2008-03-04 | The Boeing Company | Systems and methods for electrical power regulation and distribution in aircraft |
| US20070002593A1 (en) * | 2005-06-30 | 2007-01-04 | Dinh James S | Isolated DCX converter |
| JP2008138820A (en) | 2006-12-05 | 2008-06-19 | Toyota Motor Corp | Hydraulic control device of belt continuously variable transmission |
| US7859870B1 (en) * | 2008-07-29 | 2010-12-28 | Lockheed Martin Corporation | Voltage clamps for energy snubbing |
| TWI435536B (en) * | 2010-11-18 | 2014-04-21 | Univ Nat Formosa | Alternating-current chopper circuit with low noise |
| US10199950B1 (en) | 2013-07-02 | 2019-02-05 | Vlt, Inc. | Power distribution architecture with series-connected bus converter |
| WO2019014682A1 (en) * | 2017-07-14 | 2019-01-17 | Alpha Technologies Inc. | Voltage regulated ac power supply systems and methods |
Family Cites Families (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPS58218877A (en) * | 1982-06-14 | 1983-12-20 | Chiyoda:Kk | Electric power source device with controlled pulse width |
| ATE35882T1 (en) * | 1983-09-15 | 1988-08-15 | Ibm | SWITCHING POWER SUPPLY WITH OVERCURRENT PROTECTION. |
| US4698742A (en) * | 1986-08-01 | 1987-10-06 | The United States Of America As Represented By The Secretary Of The Air Force | High-voltage milberger slip slide power conditioner |
| US4943903A (en) * | 1989-03-14 | 1990-07-24 | Cardwell Jr Gilbert I | Power supply in which regulation is achieved by processing a small portion of applied power through a switching regulator |
-
1992
- 1992-03-03 US US07/845,074 patent/US5218522A/en not_active Expired - Lifetime
-
1993
- 1993-01-19 CA CA002087566A patent/CA2087566A1/en not_active Abandoned
- 1993-02-18 EP EP93102521A patent/EP0559009B1/en not_active Expired - Lifetime
- 1993-02-18 DE DE69301040T patent/DE69301040T2/en not_active Expired - Lifetime
- 1993-03-03 JP JP04290593A patent/JP3369621B2/en not_active Expired - Fee Related
Also Published As
| Publication number | Publication date |
|---|---|
| JP3369621B2 (en) | 2003-01-20 |
| EP0559009B1 (en) | 1995-12-20 |
| US5218522A (en) | 1993-06-08 |
| JPH066974A (en) | 1994-01-14 |
| DE69301040T2 (en) | 1996-07-25 |
| EP0559009A1 (en) | 1993-09-08 |
| DE69301040D1 (en) | 1996-02-01 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| US5218522A (en) | D.C. chopper regulating method and apparatus incorporating bilateral regulating voltage path | |
| US4559590A (en) | Regulated DC to DC converter | |
| EP1813012B1 (en) | An acdc converter | |
| US7272021B2 (en) | Power converter with isolated and regulated stages | |
| US10658934B2 (en) | Quasi-resonant converter with efficient light-load operation and method therefor | |
| US6664762B2 (en) | High voltage battery charger | |
| US6538909B2 (en) | Universal high efficiency power converter | |
| US4628426A (en) | Dual output DC-DC converter with independently controllable output voltages | |
| US6201719B1 (en) | Controller for power supply and method of operation thereof | |
| US6344768B1 (en) | Full-bridge DC-to-DC converter having an unipolar gate drive | |
| EP0262812A2 (en) | Buck-boost parallel resonant converter | |
| US6320764B1 (en) | Regulation circuit for a power converter and method of operation thereof | |
| US11757365B2 (en) | Dynamic transient control in resonant converters | |
| US7388761B1 (en) | High efficiency parallel post regulator for wide range input DC/DC converter | |
| US7050309B2 (en) | Power converter with output inductance | |
| US6504735B2 (en) | Regulated voltage reducing high-voltage isolated DC/DC converter system | |
| JPH1169802A (en) | Synchronous rectification circuit | |
| JPH0646535A (en) | Charger | |
| Sachin et al. | Constant and Green Mode Operation Technique in Flyback and Push-Pull DC-DC Converter | |
| Hasan et al. | Bidirectional isolated dc-dc converter with unity gain for low power applications | |
| EP4054064A1 (en) | An electrical switched mode power converter and operative procedure thereof | |
| JP2795229B2 (en) | DC-DC converter | |
| JPS6321190Y2 (en) | ||
| Kazimierczuk | Series resonant converter with phase-controlled synchronous rectifier | |
| JPH0591744A (en) | Switching regulator |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| FZDE | Discontinued |