CA2123477A1 - Delta-sigma fractional-n frequency synthesizer and frequency discriminator suitable for use therein - Google Patents
Delta-sigma fractional-n frequency synthesizer and frequency discriminator suitable for use thereinInfo
- Publication number
- CA2123477A1 CA2123477A1 CA002123477A CA2123477A CA2123477A1 CA 2123477 A1 CA2123477 A1 CA 2123477A1 CA 002123477 A CA002123477 A CA 002123477A CA 2123477 A CA2123477 A CA 2123477A CA 2123477 A1 CA2123477 A1 CA 2123477A1
- Authority
- CA
- Canada
- Prior art keywords
- frequency
- signal
- digital
- output
- error signal
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Abandoned
Links
- 238000012546 transfer Methods 0.000 claims abstract description 13
- 238000005070 sampling Methods 0.000 claims description 6
- 230000000087 stabilizing effect Effects 0.000 claims description 4
- 230000003068 static effect Effects 0.000 abstract description 2
- 230000007812 deficiency Effects 0.000 abstract 1
- 230000035945 sensitivity Effects 0.000 abstract 1
- 238000006243 chemical reaction Methods 0.000 description 4
- 238000013139 quantization Methods 0.000 description 3
- 230000009467 reduction Effects 0.000 description 3
- 238000000034 method Methods 0.000 description 2
- 101100134298 Bacillus subtilis (strain 168) nucA gene Proteins 0.000 description 1
- 208000007542 Paresis Diseases 0.000 description 1
- 238000010276 construction Methods 0.000 description 1
- 239000013078 crystal Substances 0.000 description 1
- 230000001934 delay Effects 0.000 description 1
- 238000013461 design Methods 0.000 description 1
- GWUSZQUVEVMBPI-UHFFFAOYSA-N nimetazepam Chemical compound N=1CC(=O)N(C)C2=CC=C([N+]([O-])=O)C=C2C=1C1=CC=CC=C1 GWUSZQUVEVMBPI-UHFFFAOYSA-N 0.000 description 1
- 208000012318 pareses Diseases 0.000 description 1
- 238000012545 processing Methods 0.000 description 1
- 238000007493 shaping process Methods 0.000 description 1
- 230000006641 stabilisation Effects 0.000 description 1
- 238000011105 stabilization Methods 0.000 description 1
Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION, OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
- H03L7/00—Automatic control of frequency or phase; Synchronisation
- H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
- H03L7/16—Indirect frequency synthesis, i.e. generating a desired one of a number of predetermined frequencies using a frequency- or phase-locked loop
- H03L7/18—Indirect frequency synthesis, i.e. generating a desired one of a number of predetermined frequencies using a frequency- or phase-locked loop using a frequency divider or counter in the loop
- H03L7/197—Indirect frequency synthesis, i.e. generating a desired one of a number of predetermined frequencies using a frequency- or phase-locked loop using a frequency divider or counter in the loop a time difference being used for locking the loop, the counter counting between numbers which are variable in time or the frequency divider dividing by a factor variable in time, e.g. for obtaining fractional frequency division
- H03L7/1974—Indirect frequency synthesis, i.e. generating a desired one of a number of predetermined frequencies using a frequency- or phase-locked loop using a frequency divider or counter in the loop a time difference being used for locking the loop, the counter counting between numbers which are variable in time or the frequency divider dividing by a factor variable in time, e.g. for obtaining fractional frequency division for fractional frequency division
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION, OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
- H03L7/00—Automatic control of frequency or phase; Synchronisation
- H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
- H03L7/08—Details of the phase-locked loop
- H03L7/10—Details of the phase-locked loop for assuring initial synchronisation or for broadening the capture range
- H03L7/113—Details of the phase-locked loop for assuring initial synchronisation or for broadening the capture range using frequency discriminator
Abstract
A fractional - N frequency synthesizer comprises a voltage-controlled oscillator(VCO) for generating an output signal in response to a control voltage derived by a digital-to-analog converter from a digital error signal. The error signal is derived by a differencing device which subtracts a digital signal representing a desired frequency of the output signal. The digital signal representing the output signal frequency is derived by a frequency discrimination device which determines the instant frequency of the analog output signal and provides a corresponding digital representation with zero static frequency error. In preferred embodiments, the frequency discrimination device is a delta-sigma frequency synthesizer is combination with a decimator. This frequency synthesizer configuration avoids deficiencies due to non-linearity and noise sensitivity of analog phase detectors. The invention also provides a frequency discriminator comprising a feedforward path including a multimodulus divider means for dividing the frequency of a digital signal (F0) in dependence upon a division ratio control signal, comparison means for comparing the divided frequency signal with a reference frequency signal to provide an error signal, and a feedback path comprising a transfer function of 2-Z-1 and responsive to the error signal and the reference frequency signal to provide the division ratio control signal.
Description
' 1 TITLE Delta-Sigma Fractional-N Frequency Synthesizer and Frequency Di~çrimin~tor for use therein.
FIELD OF THE INVENTION
The invention relates to frequency synth~si7~rs and especially fractional - N
frequency synthesi7~rs. The invention also relates to frequency discrimin~tors especially suitable for use in such frequency synthesizers.
BACKGROUND
Known fractional - N frequency synthesizers, such as that disclosed in US patentNo. 4,965,531, use an analog phase detector to detect the error signal or difference between the frequency being generated and a desired frequency. Such frequency synthesizers are not entirely satisfactory because analog phase detectors are not perfectly linear, can be a source of noise, and can themselves be sensitive to noise/intelrelence.
SUMMARY OF THE INVENTION:
According to one aspect of the invention, there is provided a fractional - N
synthesizer comprising a voltage controlled oscillator responsive to a control voltage (Vc) to generate an output signal (Fo) having a particular frequency, frequency discrimin~tor 20 means responsive to the output signal (Fo) and a reference signal (Fref) having a predetermined reference frequency to generate a digital signal (Do) representing the output signal frequency, differencing means responsive to the digital represent~tion signal and to a digital input signal representing a desired output frequency (Fd) to provide an error signal (e), and digital-to-analog conversion means for converting the error signal 2 5 to provide said control voltage (Vc).
With such a configuration, the combined transfer function of the frequency synthesizer can be such that q~l~nti7~tion noise generated by the frequency discrimin~tor is subst~nti~lly elimin~ted from the output signal (Fo)~
Preferably, filter means is provided between the digital-to-analog conversion 3 o means and the voltage controlled oscillator. This filter means may comprise an analog integl~tor and remove some quantization noise.
Filter means may be provided between the differencing means and the digital-to-analog conversion means. This filter means may comprise a digital integrator such that the output signal frequency (Fo) is phased locked to the reference signal frequency (FrCf).
Where both the first-mentioned filter means and the second-mentioned filter means are included, a stabilizing zero means will generally be included to ensure stability of the frequency synthesizer circuit. The stabilizing zero means may comprise a feed 5 forward loop and s--mming means provided in one of the filter means, conveniently that provided between hte differencing means and the digital-to-analog conversion means.
The frequency discrimin~tion means may comprise a frequency discrimin~tor connected to receive the output signal from the voltage-controlled oscillator and decimation filter provided between the output of the frequency discrimin~tor and the lo differencing means. The decimation filter will reduce the sampling rate of the frequency discrimin~tor, for example from 10 MHz to 1 MHz, contributing, in the process, to reduction of quantization noise.
The frequency discrimin~tion means may comprise a frequency discrimin~tor comprising a feedforward path including a multimodulus divider means for dividing the 15 frequency of the output signal (Fo) in dependence upon a division ratio control signal, comparison means for comI)~ring the divided signal with the reference signal to provide a second error signal, and a feedback path comprising a circuit having a transfer function of 2-Z-I and responsive to the error signal and the reference signal to provide the division ratio control signal.
According to a second aspect of the invention, there is provided a frequency discrimin~tor comprising a fee Irol~vard path including a multimodulus divider means for dividing the frequency of a digital signal (Fo) in dependence upon a division ratio control signal, co~p~ on means for comparing the divided frequency signal with a reference frequency signal to provide an error signal, and a feedback path comprising a transfer 25 function of 2-Z-I and responsive to the error signal and the reference frequency signal to provide the division ratio control signal.
The various objects, features, aspects and advantages of the present invention will become more appalellt from the following detailed description, in conjunction with the accompanying drawings in which:
Figure 1 illustrates a digital frequency syntht ci7er according to one aspect of the invention and Figure 2 illustrates a frequency ~is~rimin~t~r of the frequency synthesizer and in accordance with a second aspect of the invention.
. - 3 Referring first to Figure 1, a frequency synthpsi7~r comprises differencing means in the form of a subtracter 103, which receives at its "plus" input a ten bit digital word rt;plesenling a desired frequency (Fd) which the frequency synthPsi7Pr is required to generate. At its "minus" input, the differencing means 103 receives a digital signal Do 5 r~resellling the actual output frequency of the frequency synthesi7er in a manner which will become al)pa~ent later. This digital signal is a ten bit word The output of the differencing means 103 i.e. the error signal e, also ten bit word, is applied via a filter 104 to the input of a twelve bit digital-to-analog convertor 105. The digital to analog convertor 105 converts the twelve bit word to a corresponding control voltage Vc and 10 applies it via an analog filter 106 to the control input of a voltage controlled oscillator 107. A suitable voltage controlled oscillator 107 is available as part No. VCO-P-800 from Synergy Microwave Inc. The output of the voltage controlled oscillator 107 is the output signal (Fo) of the frequency synthesizer and has a frequency of about 800 MHz.
This output signal Fo also is applied to one input of frequency discrimination means 15 comprising a frequency (li.ccrimin~tor 101 and decimation means 102. A reference signal Fref which has a frequency of 10 MHz, is applied to a reference input of the frequency discrimin~tor 101. The drawing shows the reference signal Fref em~n~tin~ from a crystal oscillator 108. It will be appreciated, however, that any suitable source could be used and could be internal to the frequency discrimin~tor 101. The output of the frequency 20 ~ çrimin~tor 101 is a one bit bitstream the density of the ones representing the output frequency Fo~ This bitstream from the frequency discrimin~tor 101 is supplied todecimation filter 102 which converts it to ten bit words at a rate of 1 MHz, in the process contributing to qu~nti7~tion noise reduction. The output of the decimation filter 102 is applied to the "minus" input of dirre ~ncing means 103 as the digital signal Do 2 5 representing the measured output frequency.
The transfer functions of the various components are selected so that the complete 1QP con~titutes a phased locked loop with a bandwidth of around 50 KHz. The transfer functions of the three filters and the other components in the circuit all contribute towards a combined or overall circuit transfer function which reduces qu~nti7~tion noise 30 created by the frequency discrimin~tor 101 to an acceptable level in the control voltage Vc and hence in the phase error of the output signal Fo While various kinds of filter can be used, the specific embodiment uses an analog filter 106 is represented as an analog integrator ~0/S providing a one volt output to the VCO 107.
The filter 104 comprises an integrator formed by a summin~ device 140 and delay 5 109. The delay 109 is connected between the output of the summing device 108 and one of its inputs. The output of differencing means 103 is applied to the other input of ~u~ in~ device 108 and an input of a second sl-mming device 110, the other input of which receives the output of first summing device 108. The output of the second s~mming device 110 is the output of the filter 104. The second summing means 110lo provides a stabilization zero for the circuit transfer function to ensure stability of the phase locked loop.
The decimation filter 102 comprises a down-sampling device 112 for down-sampling by a factor of ten the bitstream received from the frequency discrimin~tor 101.
Three integrators 113, 114 and 115 are connected in series between input of the decimation filter 102 and the input of the down-sampling device 112. Each delay is a ten bit delay. The first in the series, inlegl~tor 113, merely receives the one bit bitstream which con~titlltes the least significant bit. The other two integrators 114 and 115 function as ten bit adders. These are of course functioning at the bit rate of 10 MHz.
Three differentiators 125, 126 and 127 are connected in series after the sown-sampling device 112. The differentiators 125, 126 and 127 comprise first, second and third adders 128, 129 and 130 with delay means 131, 132 and 133, respectively, connected in feedforward loops between their "plus" and "minus" inputs. These adders 128, 129 and 130 and delays 131, 132 and 133 also are ten bit capacity, but in this case 2 5 operating at the lower bit rate of 1 MHz. For further information about the construction and operation of this kind of decimation filter 102 the reader is directed to the article "
"Decimation for Signa-Delta Modulation", IEEE Transactions on Communications, Vol.
34, No. 1, January 1986, pp 72 - 76, the entire contents of which are incorporated herein by reference.
3 o For details of a suitable phase-locked loop frequency discrimin~tor, the reader is directed to an article by R.D. Beards and M.A. Copeland entitled "An Oversampling Delta-Sigma Frequency Discrimin~tor" IEEE Transactions on Circuits and Systems Part II Analog and Digital Signal Processing, January 1994, Vol. 41, No. 1 pp 26 - 32, which is inco~ ted herein by reference. The frequency discrimin~tor disclosed byBeards and Copeland is particularly suitable because it ensures that the qu~nti7~tion error introduced by the di~çrimin~tQr is spectrally shaped SQ as to be acceptably low close to DC and multiples of the reference frequency F,l,f.
While the frequency discrimin~t~r 101 provides a single bit bitstream, it would be possible to use a frequency ~ rimin~tc~r which supplied a series of digital words comprising 2, 3 or 4 bits, to represent the frequency of the output signal, providing it did so with substantially zero static frequency error.
Although the ~ çri~"i~ Qr disclosed by Beards and Copeland is prerelled, other 10 discrimin~tors are viable. For example, the discrimin~t--r could provide a continuous time zero in the feedforward loop of the ~ çrimin~tQr. Altern~tively~ the discrimin~tor could be a third order ~i~çrimin~tor. It is also envisaged that the frequency discriminator might provide a transfer function of 2-Z-l in its feedback loop. Such a frequency discrimin~tor is shown in Figure 2 and referenced 101'.
Referring now to Figure 2, in the frequency discrimin~tor 101', the output signal Fo (from the VCO 107) is applied to a multimodulus divider 201 which divides it either N, N+l, N+2 or N+3 in dependence upon a division control signal from circuit 205which provided the transfer function 2-Z-l where Z-l is a delay operator. The divider 103 provides a divided frequency signal comprising a sequence of digital words at the 2 o 10 MHz bit rate and applies it to a phase/frequency detector 202. The phase frequency detector 202 coll~pares the 10 MHz divided frequency signal with the reference frequency signal Fl to provide a secondary signal which is applied to integrator 203, which is a discrete time integrator clocked by the reference signal Fref. Alternatively integrator 203 could be a continuous time integrator which need not be clocked. The integrated divided 2 5 signal from the output of integrator 203 is sampled by quantizer 204 to provide the 1 bit bitstream for supply to decimator 102 (Figure 1). The output of q~l~nti7~r 204 is also supplied to transfer function circuit 205 which also is clocked by the reference signal F,~f.
It should be appreciated that, although such a frequency discrimin~tc-r is 3 o especially suitable for use in the frequency synthesizer disclosed herein, it is not limited to such use, but rather could also be used in other applications.
Various mo~ific~tions are feasible within the scope of the present invention. For example the bit capacity of the differencing means 103 and the digital-to-analog converter _ 6 105 might be reduced for economy. Reductions to, for example eight bits, could introduce additional quantization noise due to the redundant bits, in which case further noise shaping could be added to ensure the stability and phase locking of the frequency synthesizer. Similar considerations apply to any other qu~nti7~tion noise in the5 decimation filter 102 or the filter 104.
Vulnerability of the analog portions of the circuit, i.e. digital-to-analog convertor 105, filter 106 and VCO 107 to extern~l sources of noise and intelrelt;llce can be mitig~ted by careful design and layout to ensure that power supplies are well decoupled and sensitive signals are chielded appropliately. An advantage of the present invention, lo however, is that the rem~ininp parts of the frequency synthesizer, being digital, are robust and substantially insen~itive to noise and intelrerence. Moreover, most or all, of the circuit can be fabricated in a single integrated circuit.
Although an embodiments of the invention has been described and illustrated in detail, it is to be clearly understood that the same is by way of illustration and example 15 only and is not to be taken by way of the limitation, the spirit and scope of the present invention being limited only by the appended claims.
FIELD OF THE INVENTION
The invention relates to frequency synth~si7~rs and especially fractional - N
frequency synthesi7~rs. The invention also relates to frequency discrimin~tors especially suitable for use in such frequency synthesizers.
BACKGROUND
Known fractional - N frequency synthesizers, such as that disclosed in US patentNo. 4,965,531, use an analog phase detector to detect the error signal or difference between the frequency being generated and a desired frequency. Such frequency synthesizers are not entirely satisfactory because analog phase detectors are not perfectly linear, can be a source of noise, and can themselves be sensitive to noise/intelrelence.
SUMMARY OF THE INVENTION:
According to one aspect of the invention, there is provided a fractional - N
synthesizer comprising a voltage controlled oscillator responsive to a control voltage (Vc) to generate an output signal (Fo) having a particular frequency, frequency discrimin~tor 20 means responsive to the output signal (Fo) and a reference signal (Fref) having a predetermined reference frequency to generate a digital signal (Do) representing the output signal frequency, differencing means responsive to the digital represent~tion signal and to a digital input signal representing a desired output frequency (Fd) to provide an error signal (e), and digital-to-analog conversion means for converting the error signal 2 5 to provide said control voltage (Vc).
With such a configuration, the combined transfer function of the frequency synthesizer can be such that q~l~nti7~tion noise generated by the frequency discrimin~tor is subst~nti~lly elimin~ted from the output signal (Fo)~
Preferably, filter means is provided between the digital-to-analog conversion 3 o means and the voltage controlled oscillator. This filter means may comprise an analog integl~tor and remove some quantization noise.
Filter means may be provided between the differencing means and the digital-to-analog conversion means. This filter means may comprise a digital integrator such that the output signal frequency (Fo) is phased locked to the reference signal frequency (FrCf).
Where both the first-mentioned filter means and the second-mentioned filter means are included, a stabilizing zero means will generally be included to ensure stability of the frequency synthesizer circuit. The stabilizing zero means may comprise a feed 5 forward loop and s--mming means provided in one of the filter means, conveniently that provided between hte differencing means and the digital-to-analog conversion means.
The frequency discrimin~tion means may comprise a frequency discrimin~tor connected to receive the output signal from the voltage-controlled oscillator and decimation filter provided between the output of the frequency discrimin~tor and the lo differencing means. The decimation filter will reduce the sampling rate of the frequency discrimin~tor, for example from 10 MHz to 1 MHz, contributing, in the process, to reduction of quantization noise.
The frequency discrimin~tion means may comprise a frequency discrimin~tor comprising a feedforward path including a multimodulus divider means for dividing the 15 frequency of the output signal (Fo) in dependence upon a division ratio control signal, comparison means for comI)~ring the divided signal with the reference signal to provide a second error signal, and a feedback path comprising a circuit having a transfer function of 2-Z-I and responsive to the error signal and the reference signal to provide the division ratio control signal.
According to a second aspect of the invention, there is provided a frequency discrimin~tor comprising a fee Irol~vard path including a multimodulus divider means for dividing the frequency of a digital signal (Fo) in dependence upon a division ratio control signal, co~p~ on means for comparing the divided frequency signal with a reference frequency signal to provide an error signal, and a feedback path comprising a transfer 25 function of 2-Z-I and responsive to the error signal and the reference frequency signal to provide the division ratio control signal.
The various objects, features, aspects and advantages of the present invention will become more appalellt from the following detailed description, in conjunction with the accompanying drawings in which:
Figure 1 illustrates a digital frequency syntht ci7er according to one aspect of the invention and Figure 2 illustrates a frequency ~is~rimin~t~r of the frequency synthesizer and in accordance with a second aspect of the invention.
. - 3 Referring first to Figure 1, a frequency synthpsi7~r comprises differencing means in the form of a subtracter 103, which receives at its "plus" input a ten bit digital word rt;plesenling a desired frequency (Fd) which the frequency synthPsi7Pr is required to generate. At its "minus" input, the differencing means 103 receives a digital signal Do 5 r~resellling the actual output frequency of the frequency synthesi7er in a manner which will become al)pa~ent later. This digital signal is a ten bit word The output of the differencing means 103 i.e. the error signal e, also ten bit word, is applied via a filter 104 to the input of a twelve bit digital-to-analog convertor 105. The digital to analog convertor 105 converts the twelve bit word to a corresponding control voltage Vc and 10 applies it via an analog filter 106 to the control input of a voltage controlled oscillator 107. A suitable voltage controlled oscillator 107 is available as part No. VCO-P-800 from Synergy Microwave Inc. The output of the voltage controlled oscillator 107 is the output signal (Fo) of the frequency synthesizer and has a frequency of about 800 MHz.
This output signal Fo also is applied to one input of frequency discrimination means 15 comprising a frequency (li.ccrimin~tor 101 and decimation means 102. A reference signal Fref which has a frequency of 10 MHz, is applied to a reference input of the frequency discrimin~tor 101. The drawing shows the reference signal Fref em~n~tin~ from a crystal oscillator 108. It will be appreciated, however, that any suitable source could be used and could be internal to the frequency discrimin~tor 101. The output of the frequency 20 ~ çrimin~tor 101 is a one bit bitstream the density of the ones representing the output frequency Fo~ This bitstream from the frequency discrimin~tor 101 is supplied todecimation filter 102 which converts it to ten bit words at a rate of 1 MHz, in the process contributing to qu~nti7~tion noise reduction. The output of the decimation filter 102 is applied to the "minus" input of dirre ~ncing means 103 as the digital signal Do 2 5 representing the measured output frequency.
The transfer functions of the various components are selected so that the complete 1QP con~titutes a phased locked loop with a bandwidth of around 50 KHz. The transfer functions of the three filters and the other components in the circuit all contribute towards a combined or overall circuit transfer function which reduces qu~nti7~tion noise 30 created by the frequency discrimin~tor 101 to an acceptable level in the control voltage Vc and hence in the phase error of the output signal Fo While various kinds of filter can be used, the specific embodiment uses an analog filter 106 is represented as an analog integrator ~0/S providing a one volt output to the VCO 107.
The filter 104 comprises an integrator formed by a summin~ device 140 and delay 5 109. The delay 109 is connected between the output of the summing device 108 and one of its inputs. The output of differencing means 103 is applied to the other input of ~u~ in~ device 108 and an input of a second sl-mming device 110, the other input of which receives the output of first summing device 108. The output of the second s~mming device 110 is the output of the filter 104. The second summing means 110lo provides a stabilization zero for the circuit transfer function to ensure stability of the phase locked loop.
The decimation filter 102 comprises a down-sampling device 112 for down-sampling by a factor of ten the bitstream received from the frequency discrimin~tor 101.
Three integrators 113, 114 and 115 are connected in series between input of the decimation filter 102 and the input of the down-sampling device 112. Each delay is a ten bit delay. The first in the series, inlegl~tor 113, merely receives the one bit bitstream which con~titlltes the least significant bit. The other two integrators 114 and 115 function as ten bit adders. These are of course functioning at the bit rate of 10 MHz.
Three differentiators 125, 126 and 127 are connected in series after the sown-sampling device 112. The differentiators 125, 126 and 127 comprise first, second and third adders 128, 129 and 130 with delay means 131, 132 and 133, respectively, connected in feedforward loops between their "plus" and "minus" inputs. These adders 128, 129 and 130 and delays 131, 132 and 133 also are ten bit capacity, but in this case 2 5 operating at the lower bit rate of 1 MHz. For further information about the construction and operation of this kind of decimation filter 102 the reader is directed to the article "
"Decimation for Signa-Delta Modulation", IEEE Transactions on Communications, Vol.
34, No. 1, January 1986, pp 72 - 76, the entire contents of which are incorporated herein by reference.
3 o For details of a suitable phase-locked loop frequency discrimin~tor, the reader is directed to an article by R.D. Beards and M.A. Copeland entitled "An Oversampling Delta-Sigma Frequency Discrimin~tor" IEEE Transactions on Circuits and Systems Part II Analog and Digital Signal Processing, January 1994, Vol. 41, No. 1 pp 26 - 32, which is inco~ ted herein by reference. The frequency discrimin~tor disclosed byBeards and Copeland is particularly suitable because it ensures that the qu~nti7~tion error introduced by the di~çrimin~tQr is spectrally shaped SQ as to be acceptably low close to DC and multiples of the reference frequency F,l,f.
While the frequency discrimin~t~r 101 provides a single bit bitstream, it would be possible to use a frequency ~ rimin~tc~r which supplied a series of digital words comprising 2, 3 or 4 bits, to represent the frequency of the output signal, providing it did so with substantially zero static frequency error.
Although the ~ çri~"i~ Qr disclosed by Beards and Copeland is prerelled, other 10 discrimin~tors are viable. For example, the discrimin~t--r could provide a continuous time zero in the feedforward loop of the ~ çrimin~tQr. Altern~tively~ the discrimin~tor could be a third order ~i~çrimin~tor. It is also envisaged that the frequency discriminator might provide a transfer function of 2-Z-l in its feedback loop. Such a frequency discrimin~tor is shown in Figure 2 and referenced 101'.
Referring now to Figure 2, in the frequency discrimin~tor 101', the output signal Fo (from the VCO 107) is applied to a multimodulus divider 201 which divides it either N, N+l, N+2 or N+3 in dependence upon a division control signal from circuit 205which provided the transfer function 2-Z-l where Z-l is a delay operator. The divider 103 provides a divided frequency signal comprising a sequence of digital words at the 2 o 10 MHz bit rate and applies it to a phase/frequency detector 202. The phase frequency detector 202 coll~pares the 10 MHz divided frequency signal with the reference frequency signal Fl to provide a secondary signal which is applied to integrator 203, which is a discrete time integrator clocked by the reference signal Fref. Alternatively integrator 203 could be a continuous time integrator which need not be clocked. The integrated divided 2 5 signal from the output of integrator 203 is sampled by quantizer 204 to provide the 1 bit bitstream for supply to decimator 102 (Figure 1). The output of q~l~nti7~r 204 is also supplied to transfer function circuit 205 which also is clocked by the reference signal F,~f.
It should be appreciated that, although such a frequency discrimin~tc-r is 3 o especially suitable for use in the frequency synthesizer disclosed herein, it is not limited to such use, but rather could also be used in other applications.
Various mo~ific~tions are feasible within the scope of the present invention. For example the bit capacity of the differencing means 103 and the digital-to-analog converter _ 6 105 might be reduced for economy. Reductions to, for example eight bits, could introduce additional quantization noise due to the redundant bits, in which case further noise shaping could be added to ensure the stability and phase locking of the frequency synthesizer. Similar considerations apply to any other qu~nti7~tion noise in the5 decimation filter 102 or the filter 104.
Vulnerability of the analog portions of the circuit, i.e. digital-to-analog convertor 105, filter 106 and VCO 107 to extern~l sources of noise and intelrelt;llce can be mitig~ted by careful design and layout to ensure that power supplies are well decoupled and sensitive signals are chielded appropliately. An advantage of the present invention, lo however, is that the rem~ininp parts of the frequency synthesizer, being digital, are robust and substantially insen~itive to noise and intelrerence. Moreover, most or all, of the circuit can be fabricated in a single integrated circuit.
Although an embodiments of the invention has been described and illustrated in detail, it is to be clearly understood that the same is by way of illustration and example 15 only and is not to be taken by way of the limitation, the spirit and scope of the present invention being limited only by the appended claims.
Claims (11)
1. A frequency synthesizer comprising a voltage controlled oscillator responsive to a control voltage to generate and output signal (F0) having a particular frequency, frequency discrimination means responsive to the output signal (F0) and a reference signal (Fref) having a predetermined reference frequency to generate a digital signal (D0) representing said particular frequency, differencing means responsive to the digital representation signal and to a digital input signal representing a desired output frequency (Fd) to provide an error signal (e) and means responsive to the error signal to provide said control voltage.
2. A frequency synthesizer as claimed in claim 1, wherein the means for providing the control voltage is a digital-to-analog convertor and filter means is provided between the digital-to-analog converter and the voltage controlled oscillator.
3. A frequency synthesizer as claimed in claim 2, wherein the filter means comprises an analog integrator.
4. A frequency synthesizer as claimed in claim 1, further comprising error signal filter means connected to the differencing means, the error signal filter means comprising an integrator, the arrangement being such that the output signal frequency (F0) is phased locked to the reference signal frequency (Fref).
5. A frequency synthesizer as claimed in claim 3, further comprising a second integrating filter differencing means and the digital-to-analog converter,and one of the first and second filter means comprising stabilizing zero means.
6. A frequency synthesizer as claimed in claim 5, wherein the second filter means comprises a first summing means and a feedback delay, and the stabilizing zero means comprises a second summing means having its inputs connected to the input and output, respectively, of the first summing means and its output connected to the digital-to-analog converter.
7. A frequency synthesizer as claimed in claim 1, wherein the frequency discrimination means comprises a delta-sigma frequency discriminator and decimation filter provided between an output of the delta-sigma frequency discriminator and the differencing means.
8. A frequency synthesizer as claimed in claim 1, wherein the frequency discrimination means comprises a frequency discriminator configured as a phase-locked loop.
9. A frequency synthesizer as claimed in claim 1, wherein the frequency discrimination means comprises a frequency discriminator comprising a feedforward path including a multimodulus divider means for dividing the frequency of the output signal (Fo) in dependence upon a division ratio control signal, comparison means for comparing the divided signal with the reference signal to provide a second error signal, and a feedback path comprising a circuit having a transfer function of 2-Z-1 and responsive to the error signal and the reference signal to provide the division ratio control signal.
10. A frequency synthesizer as claimed in claim 9, wherein the multimodulus divider divides the output signal F0 by a factor (N, N+1, ...) selected in dependence upon a division ratio signal, and applies the divided frequency signal to a phase/frequency detector clocked by the reference signal (Fref) to provide the second error signal, quantifier means for integrating the error signal, and quantizing means for sampling the integrated error signal to provide said bitstream.
11. A frequency discriminator comprising a feedforward path including a multimodulus divider means for dividing the frequency of a digital signal (Fo) in dependence upon a division ratio control signal, comparison means for comparing the divided frequency signal with a reference frequency signal to provide an error signal, and a feedback path comprising a transfer function of 2-Z-1 and responsive to the error signal and the reference frequency signal to provide the division ratio control signal.
Priority Applications (7)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA002123477A CA2123477A1 (en) | 1994-05-12 | 1994-05-12 | Delta-sigma fractional-n frequency synthesizer and frequency discriminator suitable for use therein |
| US08/737,370 US5781044A (en) | 1994-05-12 | 1995-05-12 | Delta-sigma fractional-N frequency synthesizer and frequency discriminator suitable for use therein |
| DE69506112T DE69506112T2 (en) | 1994-05-12 | 1995-05-12 | FREQUENCY SYNTHETIZER WITH BROKEN PART RATIO WITH DELTA-SIGMA FREQUENCY DISCRIMINATOR |
| EP95917858A EP0772913B1 (en) | 1994-05-12 | 1995-05-12 | Fractional-n frequency synthesizer with a delta-sigma frequency discriminator |
| CA002189950A CA2189950C (en) | 1994-05-12 | 1995-05-12 | Fractional-n frequency synthesizer with a delta-sigma frequency discriminator |
| JP52925495A JP3213754B2 (en) | 1994-05-12 | 1995-05-12 | Fractional-N frequency synthesizer with delta-sigma frequency discriminator |
| PCT/CA1995/000271 WO1995031861A1 (en) | 1994-05-12 | 1995-05-12 | Fractional-n frequency synthesizer with a delta-sigma frequency discriminator |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA002123477A CA2123477A1 (en) | 1994-05-12 | 1994-05-12 | Delta-sigma fractional-n frequency synthesizer and frequency discriminator suitable for use therein |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2123477A1 true CA2123477A1 (en) | 1995-11-13 |
Family
ID=4153584
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002123477A Abandoned CA2123477A1 (en) | 1994-05-12 | 1994-05-12 | Delta-sigma fractional-n frequency synthesizer and frequency discriminator suitable for use therein |
Country Status (6)
| Country | Link |
|---|---|
| US (1) | US5781044A (en) |
| EP (1) | EP0772913B1 (en) |
| JP (1) | JP3213754B2 (en) |
| CA (1) | CA2123477A1 (en) |
| DE (1) | DE69506112T2 (en) |
| WO (1) | WO1995031861A1 (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US6236703B1 (en) | 1998-03-31 | 2001-05-22 | Philsar Semiconductor Inc. | Fractional-N divider using a delta-sigma modulator |
Families Citing this family (47)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| DE19548539A1 (en) * | 1995-12-23 | 1997-06-26 | Bosch Gmbh Robert | Mixing oscillator with a phase locked loop for a radio receiver |
| JP4036950B2 (en) * | 1998-02-09 | 2008-01-23 | 沖電気工業株式会社 | Clock generation circuit |
| GB2335322B (en) | 1998-03-13 | 2002-04-24 | Ericsson Telefon Ab L M | Phase detector |
| US6356810B1 (en) * | 1998-10-29 | 2002-03-12 | Anritsu Company | Programmable frequency reference for a signal synthesizer |
| US6816100B1 (en) | 1999-03-12 | 2004-11-09 | The Regents Of The University Of California | Analog-to-digital converters with common-mode rejection dynamic element matching, including as used in delta-sigma modulators |
| US6107890A (en) * | 1999-05-05 | 2000-08-22 | Nortel Networks Corporation | Digital phase comparator and frequency synthesizer |
| US6518801B1 (en) * | 1999-08-05 | 2003-02-11 | Agere Systems Inc. | Alias suppression method for 1-bit precision direct digital synthesizer |
| DE10019487A1 (en) * | 2000-04-19 | 2001-11-08 | Siemens Ag | Frequency synthesizer |
| KR100346839B1 (en) * | 2000-10-10 | 2002-08-03 | 삼성전자 주식회사 | Fractional-N frequency synthesizer using sigma-delta modulator and method thereof |
| US6952138B2 (en) * | 2001-09-12 | 2005-10-04 | Telefonaktiebolaget Lm Ericsson (Publ) | Generation of a phase locked loop output signal having reduced spurious spectral components |
| US6873213B2 (en) * | 2001-10-02 | 2005-03-29 | Nec Compound Semiconductor Devices, Ltd. | Fractional N frequency synthesizer |
| US6710951B1 (en) | 2001-10-31 | 2004-03-23 | Western Digital Technologies, Inc. | Phase locked loop employing a fractional frequency synthesizer as a variable oscillator |
| JP4323425B2 (en) | 2002-07-03 | 2009-09-02 | テレフオンアクチーボラゲット エル エム エリクソン（パブル） | Phase-locked loop circuit, electronic device including phase-locked loop circuit, and method for generating periodic signal |
| US7436227B2 (en) * | 2003-05-02 | 2008-10-14 | Silicon Laboratories Inc. | Dual loop architecture useful for a programmable clock source and clock multiplier applications |
| US7295077B2 (en) * | 2003-05-02 | 2007-11-13 | Silicon Laboratories Inc. | Multi-frequency clock synthesizer |
| US7064617B2 (en) * | 2003-05-02 | 2006-06-20 | Silicon Laboratories Inc. | Method and apparatus for temperature compensation |
| US7288998B2 (en) * | 2003-05-02 | 2007-10-30 | Silicon Laboratories Inc. | Voltage controlled clock synthesizer |
| US7483569B2 (en) * | 2003-05-29 | 2009-01-27 | Carnegie Mellon University | Reduced complexity correlation filters |
| US7119630B1 (en) | 2003-09-25 | 2006-10-10 | Cypress Semiconductor Corp. | Frequency synthesizer having a more versatile and efficient fractional-N control circuit and method using vector values |
| US7126436B1 (en) | 2003-09-25 | 2006-10-24 | Cypress Semiconductor Corp. | Frequency synthesizer having a more versatile and efficient fractional-N control circuit and method |
| DE102004006995B4 (en) * | 2004-02-12 | 2007-05-31 | Infineon Technologies Ag | Digital phase-locked loop for sub-μ technologies |
| DE102004009116B3 (en) * | 2004-02-25 | 2005-04-28 | Infineon Technologies Ag | Delta-sigma frequency discriminator includes dither circuit for altering clock period of reference clock signal used for sampling register for suppression of modulation interference tones |
| US7068110B2 (en) * | 2004-06-28 | 2006-06-27 | Silicon Laboratories Inc. | Phase error cancellation |
| KR100910086B1 (en) * | 2004-12-29 | 2009-07-30 | 텔레폰악티에볼라겟 엘엠 에릭슨(펍) | Phase locked loop circuit, electronic device including a phase locked loop circuit and method for generating a periodic signal |
| US8072277B1 (en) | 2005-06-30 | 2011-12-06 | Cypress Semiconductor Corporation | Spread spectrum frequency synthesizer |
| US7912109B1 (en) | 2005-06-30 | 2011-03-22 | Cypress Semiconductor Corporation | Spread spectrum frequency synthesizer with first order accumulation for frequency profile generation |
| US7961059B1 (en) | 2005-06-30 | 2011-06-14 | Cypress Semiconductor Corporation | Phase lock loop control system and method with non-consecutive feedback divide values |
| US7932787B1 (en) | 2005-06-30 | 2011-04-26 | Cypress Semiconductor Corporation | Phase lock loop control system and method |
| US7741918B1 (en) | 2005-06-30 | 2010-06-22 | Cypress Semiconductor Corporation | System and method for an enhanced noise shaping for spread spectrum modulation |
| US7701297B1 (en) | 2005-06-30 | 2010-04-20 | Cypress Semiconductor Corporation | Spread spectrum frequency synthesizer with improved frequency shape by adjusting the length of a standard curve used for spread spectrum modulation |
| US7948327B1 (en) | 2005-06-30 | 2011-05-24 | Cypress Semiconductor Corporation | Simplified phase lock loop control model system and method |
| US8174326B1 (en) | 2005-06-30 | 2012-05-08 | Cypress Semiconductor Corporation | Phase lock loop control error selection system and method |
| US7813411B1 (en) | 2005-06-30 | 2010-10-12 | Cypress Semiconductor Corporation | Spread spectrum frequency synthesizer with high order accumulation for frequency profile generation |
| US7636386B2 (en) * | 2005-11-15 | 2009-12-22 | Panasonic Corporation | Method of continuously calibrating the gain for a multi-path angle modulator |
| US7856464B2 (en) * | 2006-02-16 | 2010-12-21 | Sigmatel, Inc. | Decimation filter |
| US7680227B2 (en) * | 2006-03-02 | 2010-03-16 | Broadcom Corporation | Method and system for filter calibration using fractional-N frequency synthesized signals |
| US7929929B2 (en) * | 2007-09-25 | 2011-04-19 | Motorola Solutions, Inc. | Method and apparatus for spur reduction in a frequency synthesizer |
| US7538706B2 (en) * | 2007-09-25 | 2009-05-26 | Mediatek Inc. | Mash modulator and frequency synthesizer using the same |
| KR101004791B1 (en) | 2008-12-31 | 2011-01-04 | 엘아이지넥스원 주식회사 | Apparatus and method for Processing control signal of Frequency Synthesizer |
| CN103392162B (en) * | 2011-02-25 | 2016-08-24 | 高通技术公司 | Capacitive touch sense architecture |
| US8860432B2 (en) | 2011-02-25 | 2014-10-14 | Maxim Integrated Products, Inc. | Background noise measurement and frequency selection in touch panel sensor systems |
| US9086439B2 (en) | 2011-02-25 | 2015-07-21 | Maxim Integrated Products, Inc. | Circuits, devices and methods having pipelined capacitance sensing |
| US8862648B2 (en) | 2011-05-24 | 2014-10-14 | Taiwan Semiconductor Manufacturing Company, Ltd. | Fast filter calibration apparatus |
| JP6794784B2 (en) * | 2015-11-13 | 2020-12-02 | セイコーエプソン株式会社 | Frequency synthesizer |
| US10291389B1 (en) * | 2018-03-16 | 2019-05-14 | Stmicroelectronics International N.V. | Two-point modulator with matching gain calibration |
| CN109150177B (en) * | 2018-06-26 | 2022-07-19 | 杭州雄迈集成电路技术股份有限公司 | Decimal frequency division implementation method with dithering mechanism |
| CN112953515A (en) * | 2021-01-26 | 2021-06-11 | 北京金迈捷科技有限公司 | Fractional phase-locked loop |
Family Cites Families (10)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| FR2514968A1 (en) * | 1981-10-16 | 1983-04-22 | Trt Telecom Radio Electr | FREQUENCY SYNTHESIZER WITH QUICK ACCORD |
| US5258724A (en) * | 1983-12-30 | 1993-11-02 | Itt Corporation | Frequency synthesizer |
| US4580107A (en) * | 1984-06-06 | 1986-04-01 | The United States Of America As Represented By The Secretary Of The Air Force | Phase lock acquisition system having FLL for coarse tuning and PLL for fine tuning |
| US4810974A (en) * | 1987-06-26 | 1989-03-07 | Texas Instruments Incorporated | Drift compensated digitally tuned voltage controlled oscillator |
| US4929918A (en) * | 1989-06-07 | 1990-05-29 | International Business Machines Corporation | Setting and dynamically adjusting VCO free-running frequency at system level |
| EP0840457A3 (en) * | 1990-10-22 | 1999-08-25 | NEC Corporation | PLL frequency synthesizer capable of changing an output frequency at a high speed |
| JP2861542B2 (en) * | 1991-10-25 | 1999-02-24 | 日本電気株式会社 | Phase locked loop synthesizer |
| US5563535A (en) * | 1994-11-29 | 1996-10-08 | Microunity Systems Engineering, Inc. | Direct digital frequency synthesizer using sigma-delta techniques |
| US5552750A (en) * | 1995-09-05 | 1996-09-03 | Motorola, Inc. | Method and apparatus for determining an instantaneous phase difference between two signals |
| US5630222A (en) * | 1995-12-04 | 1997-05-13 | Motorola Inc. | Method and apparatus for generating multiple signals at multiple frequencies |
-
1994
- 1994-05-12 CA CA002123477A patent/CA2123477A1/en not_active Abandoned
-
1995
- 1995-05-12 EP EP95917858A patent/EP0772913B1/en not_active Expired - Lifetime
- 1995-05-12 DE DE69506112T patent/DE69506112T2/en not_active Expired - Lifetime
- 1995-05-12 US US08/737,370 patent/US5781044A/en not_active Expired - Lifetime
- 1995-05-12 WO PCT/CA1995/000271 patent/WO1995031861A1/en active IP Right Grant
- 1995-05-12 JP JP52925495A patent/JP3213754B2/en not_active Expired - Fee Related
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US6236703B1 (en) | 1998-03-31 | 2001-05-22 | Philsar Semiconductor Inc. | Fractional-N divider using a delta-sigma modulator |
Also Published As
| Publication number | Publication date |
|---|---|
| US5781044A (en) | 1998-07-14 |
| EP0772913A1 (en) | 1997-05-14 |
| WO1995031861A1 (en) | 1995-11-23 |
| EP0772913B1 (en) | 1998-11-18 |
| DE69506112D1 (en) | 1998-12-24 |
| JP3213754B2 (en) | 2001-10-02 |
| DE69506112T2 (en) | 1999-07-22 |
| JPH10500264A (en) | 1998-01-06 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CA2123477A1 (en) | Delta-sigma fractional-n frequency synthesizer and frequency discriminator suitable for use therein | |
| US7573955B2 (en) | Digital phase locked loop for sub-μ technologies | |
| EP0429217B1 (en) | Frequency synthesiser | |
| EP1000462B1 (en) | Frequency synthesizer systems and methods for three-point modulation with a dc response | |
| Riley et al. | Delta-sigma modulation in fractional-N frequency synthesis | |
| CN1768479B (en) | Method and system of jitter compensation | |
| Miller et al. | A multiple modulator fractional divider | |
| US6236703B1 (en) | Fractional-N divider using a delta-sigma modulator | |
| US7250823B2 (en) | Direct digital synthesis (DDS) phase locked loop (PLL) frequency synthesizer and associated methods | |
| US7068110B2 (en) | Phase error cancellation | |
| US7391842B1 (en) | Direct synthesis clock generation circuits and methods | |
| US7557661B1 (en) | Direct digital synthesis (DDS) hybrid phase-lock loop for low-jitter synchronization | |
| US7271666B1 (en) | Method and apparatus for canceling jitter in a fractional-N phase-lock loop (PLL) | |
| US10348314B2 (en) | Spread spectrum clock generator | |
| JP2002509377A (en) | ΣΔ modulator controlled phase locked loop circuit and associated method | |
| US20030058055A1 (en) | Fractional-N type frequency synthesizer | |
| US6822593B2 (en) | Digital to digital Sigma-Delta modulator and digital frequency synthesizer incorporating the same | |
| US7453325B2 (en) | Single point modulator having a PLL circuit | |
| JPH04212522A (en) | Frequency synthesizer | |
| EP1427108A1 (en) | A third order sigma-delta modulator for noise shaping in a phase locked loop and method thereof | |
| Bax et al. | A/spl Sigma/-/spl Delta/frequency discriminator based synthesizer | |
| CA2189950C (en) | Fractional-n frequency synthesizer with a delta-sigma frequency discriminator | |
| EP1484674B1 (en) | Maximally digitized fractional-N frequency synthesizer and modulator with maximal fractional spurious removing | |
| CA2267496C (en) | A fractional-n divider using a delta-sigma modulator | |
| Fang et al. | A second order ΔΣ frequency discriminator with fractional-N divider and multi-bit quantizer |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| EEER | Examination request | ||
| FZDE | Discontinued |