CA2224261A1 - Low phase noise, high q, high gain amplifier in an integrated circuit - Google Patents
Low phase noise, high q, high gain amplifier in an integrated circuit Download PDFInfo
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- CA2224261A1 CA2224261A1 CA002224261A CA2224261A CA2224261A1 CA 2224261 A1 CA2224261 A1 CA 2224261A1 CA 002224261 A CA002224261 A CA 002224261A CA 2224261 A CA2224261 A CA 2224261A CA 2224261 A1 CA2224261 A1 CA 2224261A1
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03G—CONTROL OF AMPLIFICATION
- H03G1/00—Details of arrangements for controlling amplification
- H03G1/0005—Circuits characterised by the type of controlling devices operated by a controlling current or voltage signal
- H03G1/0035—Circuits characterised by the type of controlling devices operated by a controlling current or voltage signal using continuously variable impedance elements
- H03G1/007—Circuits characterised by the type of controlling devices operated by a controlling current or voltage signal using continuously variable impedance elements using FET type devices
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION, OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
- H03L7/00—Automatic control of frequency or phase; Synchronisation
- H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
- H03L7/08—Details of the phase-locked loop
- H03L7/081—Details of the phase-locked loop provided with an additional controlled phase shifter
- H03L7/0812—Details of the phase-locked loop provided with an additional controlled phase shifter and where no voltage or current controlled oscillator is used
- H03L7/0814—Details of the phase-locked loop provided with an additional controlled phase shifter and where no voltage or current controlled oscillator is used the phase shifting device being digitally controlled
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03G—CONTROL OF AMPLIFICATION
- H03G3/00—Gain control in amplifiers or frequency changers without distortion of the input signal
- H03G3/20—Automatic control
- H03G3/30—Automatic control in amplifiers having semiconductor devices
- H03G3/3036—Automatic control in amplifiers having semiconductor devices in high-frequency amplifiers or in frequency-changers
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03J—TUNING RESONANT CIRCUITS; SELECTING RESONANT CIRCUITS
- H03J3/00—Continuous tuning
- H03J3/02—Details
- H03J3/06—Arrangements for obtaining constant bandwidth or gain throughout tuning range or ranges
- H03J3/08—Arrangements for obtaining constant bandwidth or gain throughout tuning range or ranges by varying a second parameter simultaneously with the tuning, e.g. coupling bandpass filter
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03J—TUNING RESONANT CIRCUITS; SELECTING RESONANT CIRCUITS
- H03J3/00—Continuous tuning
- H03J3/02—Details
- H03J3/16—Tuning without displacement of reactive element, e.g. by varying permeability
- H03J3/18—Tuning without displacement of reactive element, e.g. by varying permeability by discharge tube or semiconductor device simulating variable reactance
- H03J3/185—Tuning without displacement of reactive element, e.g. by varying permeability by discharge tube or semiconductor device simulating variable reactance with varactors, i.e. voltage variable reactive diodes
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION, OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
- H03L7/00—Automatic control of frequency or phase; Synchronisation
- H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
- H03L7/08—Details of the phase-locked loop
- H03L7/081—Details of the phase-locked loop provided with an additional controlled phase shifter
- H03L7/0812—Details of the phase-locked loop provided with an additional controlled phase shifter and where no voltage or current controlled oscillator is used
- H03L7/0818—Details of the phase-locked loop provided with an additional controlled phase shifter and where no voltage or current controlled oscillator is used the controlled phase shifter comprising coarse and fine delay or phase-shifting means
Abstract
An amplitude feedback control loop to control the output power level of an amplifier. The amplifier is analyzed as a classical feedback control system. Some important additions to the amplitude and frequency feedback control are required to realize a practical form of the invention on an integrated circuit. In its most general sense, the input signal is the input referred thermal noise of the amplifier plus any external source of energy. In its realization as a source the external signal is not required.
Description
LOW PHASE NOISE, HIGH Q, HIGH GAIN AMPLIFIER IN AN INTEGRATED CIRCUIT
FIELD OF INVENTION
The present invention relates to: electronic oscillators which act as the source of an oscillating electronic signal;
narrow bandwidth bandpass filters as are used in communications systems; and the implementation of a low phase noise, high Q, high gain amplifier in an integrated circuit.
BACKGROUND OF THE INVENTION
El ectronic Osci11ators Many electronic circuits require an oscillating source of energy. Communications transmitters and receivers rely on the purity and frequency accuracy of the oscillator to carry their information or to receive information. If the signal frequency is in error, the receiver may receive a degraded signal or no signal at all. Similarly, if the purity of the oscillation is poor (which is also referred to as the phase noise of the oscillator), the desired signal may also degrade or be totally corrupted.
The need for a pure, low phase noise oscillator affects the cost of all communications systems to some extent, and in some cases actually is the limiting factor in the system performance. The cost of realizing a given low level of phase noise in an oscillator directly affects whether a communications system can be built for a reasonable cost.
To some extent, the difficulty in realizing a low phase noise oscillator on the semiconductor also makes it difficult to achieve low power consumption due to the need to amplify signals to higher power levels for transmission on and off of the integrated circuit (IC). More and more, communications systems have a portable component, such as a wireless telephone, pager, and the like. The power consumption of a portable wireless device is very important as it directly affects the life-span of the batteries, and the size and weight of the batteries required to operate the device.
Narrow Band Fil ters Communications systems transmit their signals over tightly restricted radio frequency channels or 'bands'. The bands have a certain 'bandwidth' which is directly related to the amount of information which can be passed through the information channel. Due to the relative scarcity of radio frequency bands and the limited ability of electronic instruments to use extremely high bands, most communications systems are forced to operate in close proximity to other bands. From the point of view of the communications user, all other signals can be regarded as interfering signals. Hence, communications systems attempt to use the band of interest while excluding interference from all other bands.
Communications systems all use some form of band limiting filter to allow the desired information to pass while excluding all other interference. The ability to build the band limiting filter such that it matches exactly the information sent usually affects the ultimate performance of the system, and the cost of its realization.
Band limiting is done with filters. While some filtering can be done digitally, due to the limitation of quantizers to digitize very large bands, and the high power consumption of such techniques, most communications systems rely on some form of analog filtering. An analog filter generally consumes no power, and loses very little in its implementation. The problem with analog filters is that they are limited in their ability to attain ideal filters by the quality or 'Q' of the components. This is the same limitation which applies to the quality of electronic oscillators discussed above. As a result, the Q of the filter components affects the ability to realize better communications systems.
In general, low loss passive filters are built with combinations of lumped elements known as inductors and capacitors, or with equivalent distributed resonant structures such as ceramics, crystals, resonant transmission lines, and the like, or with hybrid lumped element structures with partially resonant structures, such as shortened transmission lines with capacitors or inductors added.
At radio frequencies, high quality inductors are much more difficult to build than high quality capacitors. In general, inductors useful at radio frequencies are also much larger than capacitors. As a result the inductors tend to limit the radio filter design (or the low phase noise oscillator design).
Because the integrated circuit is very small, and necessarily made of semiconducting materials, it is very difficult to build a low loss inductor on an integrated circuit. As a result, filters requiring inductors or oscillators tend to be of a lower quality that those which can be achieved with components built external to the semiconductor (off-chip). This directly affects the cost of manufacturing a communications system.
Hence, the filter is limited for very similar reasons to the limitations on the low phase noise oscillator.
Historical Solutions Many techniques have been devised to make oscillators both accurate in frequency and having low phase noise. Most techniques require complex circuits and shielding which is not readily integrable into an integrated semiconductor solution.
As a result pure oscillators with low phase noise are not generally realizable in integrated circuit form without the aid of many external components, which are directly used to control the phase noise of the oscillator.
Traditionally, a separate external oscillator circuit is used to provide the low phase noise source. This solution works well but is more expensive than an integrated solution. More recently, a common approach is to implement a partial solution whereby a resonant circuit with a high quality factor (Q) is used external to the semiconductor and only the active transistors are on the semiconductor. This solution still suffers from the expense of the external resonator. Also, if the frequency is high, it can be very difficult to properly couple the internal circuit to the external resonator. This is especially true if low cost plastic packages are used with high pin counts.
Many recent attempts to build resonators directly on the semiconductor or immediately above the semiconductor using lithographic or other techniques have met with very limited success, and the resonators achieved are of comparatively poor quality, for example Q's of from 5 to 20.
Thermal amplifiers have been implemented in a very crude and uncontrolled form for many years in the form of a device known as a super regenerative receiver. This device, due to its lack of control, alternates itself from completely turned off, through the thermal amplifier state, into an oscillating state.
This type of device benefits from the momentary high Q achieved as it passes through the optimum gain state. The Q of the device varies over an extreme range as it passes from off to oscillating. As a result, the device must be pulsed on and off. The super regenerative receiver also benefits from the fact that it has very low power consumption. Super regenerative receivers have typically only been used in relatively simple applications, such as garage door openers, due to their crude control mechanisms. They can achieve very high sensitivity but are difficult to tune and are highly non-linear. This has made the super regenerative receiver a poorly understood device. As a result of these factors, super regenerative receivers have seen limited usefulness in more general high performance communications applications. Super regenerative receivers also require a stabilizing device such as a high Q filter or resonator. This increases their cost and limits the ability to integrate this type of device on an integrated circuit.
SU~ RY OF THE INrVENTION
This invention addresses a technique to produce a low phase noise, high Q, high gain amplifier which may be realized in a low power integrated circuit form without the need for an external high quality resonator. This type of application is sometimes called an active filter.
The high Q high gain amplifier may be used as the source of a low phase noise electronic signal, despite the fact that it is not technically an oscillator. It is more correctly an amplifier. For the purposes of discussion the invention will be referred to as the 'KT amplifier' due to the important role of thermal noise played in the invention.
The KT amplifier may have a signal as an input, in which case it is used as an active filter with very high Q and very high gain. The KT amplifier may be used with no external source, but may use its own thermal noise as its input to produce a high quality low phase noise sinusoidal signal. In this case, the device performs the identical function to an oscillator in a communications system.
The KT amplifier invention, in either of its forms as a source or as an active filter, can be realized on an integrated circuit at radio frequencies without the need for high quality lumped or distributed elements, but simply with a high quality low cost reference such as might be generated by a fundamental mode AT cut crystal based device.
As a result of this invention, it will be possible to build communications systems at lower cost, with high performance, and with very low power consumption. The Q of the oscillator or filter resonator is one measure of the quality of the device. Typical on-chip resonators today have Q's of from 5 to 20 at frequency ranges of use to radio devices. This invention presents a technique which allows Q's of from 10 to over 100 times higher than is common using on chip resonators.
Other advantages, objects and features of the present invention will be readily apparent to those skilled in the art from a review of the following detailed descriptions of a preferred embodiment in conjunction with the accompanying drawings and claims.
BRIEF DESCRIPTION OF THE DRAWINGS
The embodiments of the invention will now be described with reference to the accompanying drawings, in which:
Figure 1 is the KT amplifier modelled as a classic feedback control system;
Figure 2 is a simplified KT amplifier;
Figure 3 is a general form of an embodiment of the invention;
and Figure 4 is an example of a more detailed embodiment of the KT
amplifier invention.
DETAILED DESCRIPTION OF THE INVENTION
This invention makes use of an amplitude feedback control loop to control the output power level of the KT amplifier. The KT
amplifier is analyzed as a classical feedback control system, as shown in Figure 1. As will be shown later, some important additions to the amplitude and frequency feedback control are required to realize a practical form of the invention on an integrated circuit. In its most general sense, the input signal, Vin, is the input referred thermal noise of the amplifier plus any external source of energy. In its realization as a source the external signal, Vext, is not required.
Vin = KTFB + Fin where:
K is Boltzman's constant, T, is the absolute temperature in degrees Kelvin, B is the 3 dB bandwidth of the thermal noise in the steady state closed loop measured in Hertz (Hz), Vext is the external input signal, and F is the noise figure of the amplifier.
The output signal is the output of the KT amplifier.
Vo = VinG
where G is the closed loop gain, if the open loop gain is less than unity. If the open loop gain, AH, is greater than unity the device will oscillate and the output will increase until it is limited by the saturation of the amplifier.
For AH<1, the steady state closed loop gain of the system is:
Vo/Vin = A/(1-AH) For illustration purposes only, we can assume that the gain of H= 1 and is realized by a simple delay T. This is not necessary for the invention but makes the analysis simpler for illustration purposes. The KT amplifier may be redrawn as shown in Figure 2. In this case the steady state gain at frequency f is given by:
Vo/Vin = A/(1-Acos(2piTf)), where Pi = 3.14159.
This is a tuned amplifier with gain which approaches infinity for A = 1-E and E approaching 0. This high gain state occurs only at frequencies fo, where fo = n/2piT and n=integer.
In its preferred state the invention would reduce the open loop gain at all integer multiples except the preferred frequency.
The most easily realized preferred frequency is the case where n=1. In this case the undesired DC case would be limited for example by AC coupling and the undesired integers >1 could be limited by the amplifier bandwidth or by any other technique.
In this case, for Ec<l, the gain is given approximately by:
G(fo) = -l/E
where the negative sign simply indicates a phase inversion at the output relative to the input.
The Q of the KT amplifier is given by:
Q=fo/B
where B is the 3 dB bandwidth of the output signal. Hence, the gain at fo+/-B/2 is given by:
Gain(fo+/-B/2) = Gain(fo)/(2 .5) It can easily be shown for this example that for high gains, i.e. E<<l, that the Q may be expressed approximately as:
Q=3.45/EAo.5 Similarly, the 3 dB bandwidth of the output signal, B, can be shown to be approximately:
B=foE Ø5/3.45 As an example:
Let the open loop gain be AH = l-E where E=le-5, for n=l and AH far from unity for n not equal to 1, and assume the noise figure F is:
f=5 ds, Thermal noise KT = -174dBm/Hz, Input signal Vin = 0 T = 2nsec The output frequency fo would be fO 1/T = 500 MHz, the output gain would be G(fo) = -201Og(E) = 100 dB
The output Q would be:
Q = 3.45/(le-4)A.5 = 1091 The 3 dB bandwidth would be given by:
B = fo/Q = 500Mhz/1090 = 459 kHz The output power in a 50 ohm system would be:
Po = KTBFG(fo) = -174+101Og(459e3)+5+100 = -12 dBm (or 152mV
peak-to-peak (mVpp) in 50 Ohms) The noise floor of the source will simply be KTFA. (Note, normally H will have some losses, hence A will compensate by being slightly greater than 1.) The phase noise of the oscillator at any given frequency can then be easily calculated using Lessons model.
The same example device can be used to amplify signals inside of its 3 dB bandwidth. Within the limitations of the KT
amplifier output swing the signal will be amplified by 100 dB.
Hence, to achieve an output signal-to-noise ratio (SNR) of lOdB, it would be necessary to inject a signal of input power, Pin, where Pin = Po-G(fO) +SNR = -12 -100 +10= -102 dBm Hence, in this example, the input sensitivity of the KT
amplifier is -102dBm for an output SNR of +lOdB and the signal bandwidth could be up to 459kHz.
The output signal power of the amplifier, Po, would be just, Po = Pi + 100 = -102 +100 = -2 dBm (or 500 mVpp in 500hms) It should be noted that as the input signal is increased quickly relative to the bandwidth of the amplitude control loop, then the output signal increases linearly until the amplifier A starts to saturate. We can refer to the difference between the minimum useful input signal at a given SNR, to maximum signal before any significant change in closed loop gain, as the linear dynamic range. At end of the linear dynamic range the loop will have gain of less than unity and the Q will start to drop, i.e. the noise bandwidth will start to increase.
This will be compensated for, however, by the fact that the signal is growing, hence the SNR will be maintained at a useful level over a much wider range. It should be noted that this type of amplifier would not be useful to demodulate a small input signal in the presence of a larger input signal in the same band, as the larger input signal will reduce the Q and the gain of the amplifier. As a result, all practical amplifier applications must use some combination of traditional pre-filtering or feedback filtering to keep the loop from being captured by interfering signals. The extent of the traditional filtering required would depend upon the application. This limits the usefulness of the KT amplifier in some communications systems, but in no way limits its general usefulness as a signal source.
A practical embodiment of the invention is shown in Figure 3.
The classical feedback loop requires the addition of:
a) A gain controlled amplifier, possibly with both fine gain adjust and a crude or discrete gain adjust. The fine gain adjust is such that it has a very limited gain adjust over a large control voltage. This is very important to allow the amplifier to reach the tight levels of gain controls required for the invention. The discrete gain adjust will be used to place the amplifier at a gain within the control range of the fine gain adjust. The discrete gain adjusts will be noise immune as they are of a switched nature and do not rely on a low noise analog control. The need for the discrete gain adjust and the number of discrete levels required will depend upon the precision and the Q levels to be achieved and on the precision of the IC processes being used.
b) A method to measure the output signal strength of the amplifier, called here the received signal strength indicator circuit (RSSI).
c) A method to limit the bandwidth of the RSSI output. This filter will limit noise and upper order signal harmonics generated by the RSSI. The bandwidth of this filter will depend upon the application, as either a signal source (low noise oscillator) or as an active filter with communications signal as the input to the KT amplifier.
d) A method to generate a reference voltage, VREF.
e) A method to produce an amplitude error signal, Ea, by subtracting the output of the band-limited RSSI signal from the VREF signal.
f) A method to integrate the error signal Ea with a high gain integrator.
g) The output of the integrator is appropriately level shifted and scaled so as to control the fine gain adjust of the amplifier, A.
h) A method to measure the output in discrete steps such as an A/D converter, to allow the gain to be adjusted in discrete steps to within the fine adjust range where the integrator takes over.
The frequency of the signal being generated or filtered is controlled by the delay T. In an IC this delay cannot be achieved accurately without some form of feedback. The preferred method of the invention is to limit the range of delays through some physical means, which will limit the range of possible oscillation frequencies. The practical limits must take into account the tolerances achievable on a given IC
process. +/- 20~ of the desired delay is a commonly achievable accuracy with out special means.
Depending upon the application the fine frequency adjust required for the invention may be achieved in a number of different forms. If a low phase noise signal source is desired then a crystal oscillator reference should be used with a phase locked loop (PLL). This type of circuitry is readily achievable in IC form with only the crystal itself and portions of the PLL low pass loop filter required external to the IC.
In the simplest form the reference may be chosen as a fixed integer sub-multiple of the desired output frequency. In a more advanced variation of the invention the reference may be operated in a variable PLL which can achieve many sub-multiple division ratios, including rational numbers such as might be achieved with a fractional-N synthesizer based PLL for example.
For the purpose of simplicity, and without loss of generality, we will simply describe the integer sub-multiple example.
If the desired output frequency, fo, and the reference frequency fref, are related by the division ratio, n, as follows, fref = fo/N, Then within the bandwidth of the PLL the phase noise of the output source offset by some frequency, f, from its central spectral line, No(f), will not be that predicted by the Q of the free running KT amplifier, Nkt(f), but that of the reference, Nref(f), multiplied up to the output frequency, that is within the loop bandwidth:
No(f) = Nref(f) + 2010g(N) expressed in dBc/Hz offset from the center frequency by f Hz.
Outside of the bandwidth of the PLL the KT amplifier output phase noise will simply be that of the free running KT
amplifier. Due to the high quality of the free running phase noise spectrum of the KT amplifier, a given application can use a very narrow PLL loop bandwidth, when compared with the loop bandwidth of a traditional noisy oscillator. Due to the extreme problems which can be caused by high loop bandwidths, and high free running phase noise of the oscillator, in many applications the result is that the KT amplifier would be the only solution for an on chip low phase noise source.
If the in-close phase noise of the KT amplifier is not a requirement for the application as may be the case say for some less demanding applications then, the frequency control loop may be simplified to a simple frequency locked loop (FLL).
This type of loop simply counts the output frequency over a long period of time, and compares the result with a similar count of the reference frequency over the same period of time.
The comparison is used in its simplest form to decide to adjust the delay T up or down to achieve the correct output frequency.
The tolerance of the output frequency will be approximately the accuracy of the crystal in PPM, and the reciprocal of the count interval, which ever is greater. Typically a low noise fine frequency control loop can be built with a charge pump on an IC which pumps up or down to adjust the frequency. The fine phase adjust can be achieved in a number of ways. One method is to use a voltage variable capacitance, such as a bipolar transistor junction in back bias, as the delay adjust element.
Similar to the amplitude control loop the delay can be implemented with fine and crude adjustments, with the crude steps adjusted by discrete means and the fine adjust controlled by means of a large voltage to control a fine delay adjust.
This is the most noise immune approach. Again requirement for the crude adjust, and the number of levels of crude adjust will be determined by the precision of the application and the tolerances of the IC process used.
A more detailed embodiment of the invention is shown in Figure 4 with examples of realizable approaches on an IC
process.
Important aspects of the invention are as follows:
a) A method to adjust the gain of the closed loop to a high level of tolerance this requires as a minimum - a variable gain amplifier - a method to produce a signal, Vm, related to the signal strength of the output signal - method to filter the noise bandwidth of the signal Vm - a method to generate a reference voltage - a method to compare the reference voltage with the filtered Vm to produce an error signal Er - a method to integrate Er - a method to create an amplitude control signal, Ac, based upon the integrated value of Er - a method to adjust the gain of the amplifier, A, based upon the value of the control signal Ac.
b) A method to adjust the frequency or the phase of the KT
amplifier output to a high degree of tolerance the frequency control of the KT amplifier requires as a minimum - a method to divide the output signal to a lower frequency - a method to count the lower frequency over an interval of time, Tc - a method to compare the result of the count with the expected number of counts of the reference over the same period of time, - a method to limit the bandwidth of the comparison - a method to adjust the delay of the KT amplifier loop based upon the results of the band-limited comparison.
All of the above adjusts may need to be done in crude discrete steps and in fine vernier steps depending upon the application.
This does not change the generality of the invention.
Numerous modifications, variations and adaptations may be made to the particular embodiments of the invention described above without departing from the scope of the invention, which is defined in the claims.
FIELD OF INVENTION
The present invention relates to: electronic oscillators which act as the source of an oscillating electronic signal;
narrow bandwidth bandpass filters as are used in communications systems; and the implementation of a low phase noise, high Q, high gain amplifier in an integrated circuit.
BACKGROUND OF THE INVENTION
El ectronic Osci11ators Many electronic circuits require an oscillating source of energy. Communications transmitters and receivers rely on the purity and frequency accuracy of the oscillator to carry their information or to receive information. If the signal frequency is in error, the receiver may receive a degraded signal or no signal at all. Similarly, if the purity of the oscillation is poor (which is also referred to as the phase noise of the oscillator), the desired signal may also degrade or be totally corrupted.
The need for a pure, low phase noise oscillator affects the cost of all communications systems to some extent, and in some cases actually is the limiting factor in the system performance. The cost of realizing a given low level of phase noise in an oscillator directly affects whether a communications system can be built for a reasonable cost.
To some extent, the difficulty in realizing a low phase noise oscillator on the semiconductor also makes it difficult to achieve low power consumption due to the need to amplify signals to higher power levels for transmission on and off of the integrated circuit (IC). More and more, communications systems have a portable component, such as a wireless telephone, pager, and the like. The power consumption of a portable wireless device is very important as it directly affects the life-span of the batteries, and the size and weight of the batteries required to operate the device.
Narrow Band Fil ters Communications systems transmit their signals over tightly restricted radio frequency channels or 'bands'. The bands have a certain 'bandwidth' which is directly related to the amount of information which can be passed through the information channel. Due to the relative scarcity of radio frequency bands and the limited ability of electronic instruments to use extremely high bands, most communications systems are forced to operate in close proximity to other bands. From the point of view of the communications user, all other signals can be regarded as interfering signals. Hence, communications systems attempt to use the band of interest while excluding interference from all other bands.
Communications systems all use some form of band limiting filter to allow the desired information to pass while excluding all other interference. The ability to build the band limiting filter such that it matches exactly the information sent usually affects the ultimate performance of the system, and the cost of its realization.
Band limiting is done with filters. While some filtering can be done digitally, due to the limitation of quantizers to digitize very large bands, and the high power consumption of such techniques, most communications systems rely on some form of analog filtering. An analog filter generally consumes no power, and loses very little in its implementation. The problem with analog filters is that they are limited in their ability to attain ideal filters by the quality or 'Q' of the components. This is the same limitation which applies to the quality of electronic oscillators discussed above. As a result, the Q of the filter components affects the ability to realize better communications systems.
In general, low loss passive filters are built with combinations of lumped elements known as inductors and capacitors, or with equivalent distributed resonant structures such as ceramics, crystals, resonant transmission lines, and the like, or with hybrid lumped element structures with partially resonant structures, such as shortened transmission lines with capacitors or inductors added.
At radio frequencies, high quality inductors are much more difficult to build than high quality capacitors. In general, inductors useful at radio frequencies are also much larger than capacitors. As a result the inductors tend to limit the radio filter design (or the low phase noise oscillator design).
Because the integrated circuit is very small, and necessarily made of semiconducting materials, it is very difficult to build a low loss inductor on an integrated circuit. As a result, filters requiring inductors or oscillators tend to be of a lower quality that those which can be achieved with components built external to the semiconductor (off-chip). This directly affects the cost of manufacturing a communications system.
Hence, the filter is limited for very similar reasons to the limitations on the low phase noise oscillator.
Historical Solutions Many techniques have been devised to make oscillators both accurate in frequency and having low phase noise. Most techniques require complex circuits and shielding which is not readily integrable into an integrated semiconductor solution.
As a result pure oscillators with low phase noise are not generally realizable in integrated circuit form without the aid of many external components, which are directly used to control the phase noise of the oscillator.
Traditionally, a separate external oscillator circuit is used to provide the low phase noise source. This solution works well but is more expensive than an integrated solution. More recently, a common approach is to implement a partial solution whereby a resonant circuit with a high quality factor (Q) is used external to the semiconductor and only the active transistors are on the semiconductor. This solution still suffers from the expense of the external resonator. Also, if the frequency is high, it can be very difficult to properly couple the internal circuit to the external resonator. This is especially true if low cost plastic packages are used with high pin counts.
Many recent attempts to build resonators directly on the semiconductor or immediately above the semiconductor using lithographic or other techniques have met with very limited success, and the resonators achieved are of comparatively poor quality, for example Q's of from 5 to 20.
Thermal amplifiers have been implemented in a very crude and uncontrolled form for many years in the form of a device known as a super regenerative receiver. This device, due to its lack of control, alternates itself from completely turned off, through the thermal amplifier state, into an oscillating state.
This type of device benefits from the momentary high Q achieved as it passes through the optimum gain state. The Q of the device varies over an extreme range as it passes from off to oscillating. As a result, the device must be pulsed on and off. The super regenerative receiver also benefits from the fact that it has very low power consumption. Super regenerative receivers have typically only been used in relatively simple applications, such as garage door openers, due to their crude control mechanisms. They can achieve very high sensitivity but are difficult to tune and are highly non-linear. This has made the super regenerative receiver a poorly understood device. As a result of these factors, super regenerative receivers have seen limited usefulness in more general high performance communications applications. Super regenerative receivers also require a stabilizing device such as a high Q filter or resonator. This increases their cost and limits the ability to integrate this type of device on an integrated circuit.
SU~ RY OF THE INrVENTION
This invention addresses a technique to produce a low phase noise, high Q, high gain amplifier which may be realized in a low power integrated circuit form without the need for an external high quality resonator. This type of application is sometimes called an active filter.
The high Q high gain amplifier may be used as the source of a low phase noise electronic signal, despite the fact that it is not technically an oscillator. It is more correctly an amplifier. For the purposes of discussion the invention will be referred to as the 'KT amplifier' due to the important role of thermal noise played in the invention.
The KT amplifier may have a signal as an input, in which case it is used as an active filter with very high Q and very high gain. The KT amplifier may be used with no external source, but may use its own thermal noise as its input to produce a high quality low phase noise sinusoidal signal. In this case, the device performs the identical function to an oscillator in a communications system.
The KT amplifier invention, in either of its forms as a source or as an active filter, can be realized on an integrated circuit at radio frequencies without the need for high quality lumped or distributed elements, but simply with a high quality low cost reference such as might be generated by a fundamental mode AT cut crystal based device.
As a result of this invention, it will be possible to build communications systems at lower cost, with high performance, and with very low power consumption. The Q of the oscillator or filter resonator is one measure of the quality of the device. Typical on-chip resonators today have Q's of from 5 to 20 at frequency ranges of use to radio devices. This invention presents a technique which allows Q's of from 10 to over 100 times higher than is common using on chip resonators.
Other advantages, objects and features of the present invention will be readily apparent to those skilled in the art from a review of the following detailed descriptions of a preferred embodiment in conjunction with the accompanying drawings and claims.
BRIEF DESCRIPTION OF THE DRAWINGS
The embodiments of the invention will now be described with reference to the accompanying drawings, in which:
Figure 1 is the KT amplifier modelled as a classic feedback control system;
Figure 2 is a simplified KT amplifier;
Figure 3 is a general form of an embodiment of the invention;
and Figure 4 is an example of a more detailed embodiment of the KT
amplifier invention.
DETAILED DESCRIPTION OF THE INVENTION
This invention makes use of an amplitude feedback control loop to control the output power level of the KT amplifier. The KT
amplifier is analyzed as a classical feedback control system, as shown in Figure 1. As will be shown later, some important additions to the amplitude and frequency feedback control are required to realize a practical form of the invention on an integrated circuit. In its most general sense, the input signal, Vin, is the input referred thermal noise of the amplifier plus any external source of energy. In its realization as a source the external signal, Vext, is not required.
Vin = KTFB + Fin where:
K is Boltzman's constant, T, is the absolute temperature in degrees Kelvin, B is the 3 dB bandwidth of the thermal noise in the steady state closed loop measured in Hertz (Hz), Vext is the external input signal, and F is the noise figure of the amplifier.
The output signal is the output of the KT amplifier.
Vo = VinG
where G is the closed loop gain, if the open loop gain is less than unity. If the open loop gain, AH, is greater than unity the device will oscillate and the output will increase until it is limited by the saturation of the amplifier.
For AH<1, the steady state closed loop gain of the system is:
Vo/Vin = A/(1-AH) For illustration purposes only, we can assume that the gain of H= 1 and is realized by a simple delay T. This is not necessary for the invention but makes the analysis simpler for illustration purposes. The KT amplifier may be redrawn as shown in Figure 2. In this case the steady state gain at frequency f is given by:
Vo/Vin = A/(1-Acos(2piTf)), where Pi = 3.14159.
This is a tuned amplifier with gain which approaches infinity for A = 1-E and E approaching 0. This high gain state occurs only at frequencies fo, where fo = n/2piT and n=integer.
In its preferred state the invention would reduce the open loop gain at all integer multiples except the preferred frequency.
The most easily realized preferred frequency is the case where n=1. In this case the undesired DC case would be limited for example by AC coupling and the undesired integers >1 could be limited by the amplifier bandwidth or by any other technique.
In this case, for Ec<l, the gain is given approximately by:
G(fo) = -l/E
where the negative sign simply indicates a phase inversion at the output relative to the input.
The Q of the KT amplifier is given by:
Q=fo/B
where B is the 3 dB bandwidth of the output signal. Hence, the gain at fo+/-B/2 is given by:
Gain(fo+/-B/2) = Gain(fo)/(2 .5) It can easily be shown for this example that for high gains, i.e. E<<l, that the Q may be expressed approximately as:
Q=3.45/EAo.5 Similarly, the 3 dB bandwidth of the output signal, B, can be shown to be approximately:
B=foE Ø5/3.45 As an example:
Let the open loop gain be AH = l-E where E=le-5, for n=l and AH far from unity for n not equal to 1, and assume the noise figure F is:
f=5 ds, Thermal noise KT = -174dBm/Hz, Input signal Vin = 0 T = 2nsec The output frequency fo would be fO 1/T = 500 MHz, the output gain would be G(fo) = -201Og(E) = 100 dB
The output Q would be:
Q = 3.45/(le-4)A.5 = 1091 The 3 dB bandwidth would be given by:
B = fo/Q = 500Mhz/1090 = 459 kHz The output power in a 50 ohm system would be:
Po = KTBFG(fo) = -174+101Og(459e3)+5+100 = -12 dBm (or 152mV
peak-to-peak (mVpp) in 50 Ohms) The noise floor of the source will simply be KTFA. (Note, normally H will have some losses, hence A will compensate by being slightly greater than 1.) The phase noise of the oscillator at any given frequency can then be easily calculated using Lessons model.
The same example device can be used to amplify signals inside of its 3 dB bandwidth. Within the limitations of the KT
amplifier output swing the signal will be amplified by 100 dB.
Hence, to achieve an output signal-to-noise ratio (SNR) of lOdB, it would be necessary to inject a signal of input power, Pin, where Pin = Po-G(fO) +SNR = -12 -100 +10= -102 dBm Hence, in this example, the input sensitivity of the KT
amplifier is -102dBm for an output SNR of +lOdB and the signal bandwidth could be up to 459kHz.
The output signal power of the amplifier, Po, would be just, Po = Pi + 100 = -102 +100 = -2 dBm (or 500 mVpp in 500hms) It should be noted that as the input signal is increased quickly relative to the bandwidth of the amplitude control loop, then the output signal increases linearly until the amplifier A starts to saturate. We can refer to the difference between the minimum useful input signal at a given SNR, to maximum signal before any significant change in closed loop gain, as the linear dynamic range. At end of the linear dynamic range the loop will have gain of less than unity and the Q will start to drop, i.e. the noise bandwidth will start to increase.
This will be compensated for, however, by the fact that the signal is growing, hence the SNR will be maintained at a useful level over a much wider range. It should be noted that this type of amplifier would not be useful to demodulate a small input signal in the presence of a larger input signal in the same band, as the larger input signal will reduce the Q and the gain of the amplifier. As a result, all practical amplifier applications must use some combination of traditional pre-filtering or feedback filtering to keep the loop from being captured by interfering signals. The extent of the traditional filtering required would depend upon the application. This limits the usefulness of the KT amplifier in some communications systems, but in no way limits its general usefulness as a signal source.
A practical embodiment of the invention is shown in Figure 3.
The classical feedback loop requires the addition of:
a) A gain controlled amplifier, possibly with both fine gain adjust and a crude or discrete gain adjust. The fine gain adjust is such that it has a very limited gain adjust over a large control voltage. This is very important to allow the amplifier to reach the tight levels of gain controls required for the invention. The discrete gain adjust will be used to place the amplifier at a gain within the control range of the fine gain adjust. The discrete gain adjusts will be noise immune as they are of a switched nature and do not rely on a low noise analog control. The need for the discrete gain adjust and the number of discrete levels required will depend upon the precision and the Q levels to be achieved and on the precision of the IC processes being used.
b) A method to measure the output signal strength of the amplifier, called here the received signal strength indicator circuit (RSSI).
c) A method to limit the bandwidth of the RSSI output. This filter will limit noise and upper order signal harmonics generated by the RSSI. The bandwidth of this filter will depend upon the application, as either a signal source (low noise oscillator) or as an active filter with communications signal as the input to the KT amplifier.
d) A method to generate a reference voltage, VREF.
e) A method to produce an amplitude error signal, Ea, by subtracting the output of the band-limited RSSI signal from the VREF signal.
f) A method to integrate the error signal Ea with a high gain integrator.
g) The output of the integrator is appropriately level shifted and scaled so as to control the fine gain adjust of the amplifier, A.
h) A method to measure the output in discrete steps such as an A/D converter, to allow the gain to be adjusted in discrete steps to within the fine adjust range where the integrator takes over.
The frequency of the signal being generated or filtered is controlled by the delay T. In an IC this delay cannot be achieved accurately without some form of feedback. The preferred method of the invention is to limit the range of delays through some physical means, which will limit the range of possible oscillation frequencies. The practical limits must take into account the tolerances achievable on a given IC
process. +/- 20~ of the desired delay is a commonly achievable accuracy with out special means.
Depending upon the application the fine frequency adjust required for the invention may be achieved in a number of different forms. If a low phase noise signal source is desired then a crystal oscillator reference should be used with a phase locked loop (PLL). This type of circuitry is readily achievable in IC form with only the crystal itself and portions of the PLL low pass loop filter required external to the IC.
In the simplest form the reference may be chosen as a fixed integer sub-multiple of the desired output frequency. In a more advanced variation of the invention the reference may be operated in a variable PLL which can achieve many sub-multiple division ratios, including rational numbers such as might be achieved with a fractional-N synthesizer based PLL for example.
For the purpose of simplicity, and without loss of generality, we will simply describe the integer sub-multiple example.
If the desired output frequency, fo, and the reference frequency fref, are related by the division ratio, n, as follows, fref = fo/N, Then within the bandwidth of the PLL the phase noise of the output source offset by some frequency, f, from its central spectral line, No(f), will not be that predicted by the Q of the free running KT amplifier, Nkt(f), but that of the reference, Nref(f), multiplied up to the output frequency, that is within the loop bandwidth:
No(f) = Nref(f) + 2010g(N) expressed in dBc/Hz offset from the center frequency by f Hz.
Outside of the bandwidth of the PLL the KT amplifier output phase noise will simply be that of the free running KT
amplifier. Due to the high quality of the free running phase noise spectrum of the KT amplifier, a given application can use a very narrow PLL loop bandwidth, when compared with the loop bandwidth of a traditional noisy oscillator. Due to the extreme problems which can be caused by high loop bandwidths, and high free running phase noise of the oscillator, in many applications the result is that the KT amplifier would be the only solution for an on chip low phase noise source.
If the in-close phase noise of the KT amplifier is not a requirement for the application as may be the case say for some less demanding applications then, the frequency control loop may be simplified to a simple frequency locked loop (FLL).
This type of loop simply counts the output frequency over a long period of time, and compares the result with a similar count of the reference frequency over the same period of time.
The comparison is used in its simplest form to decide to adjust the delay T up or down to achieve the correct output frequency.
The tolerance of the output frequency will be approximately the accuracy of the crystal in PPM, and the reciprocal of the count interval, which ever is greater. Typically a low noise fine frequency control loop can be built with a charge pump on an IC which pumps up or down to adjust the frequency. The fine phase adjust can be achieved in a number of ways. One method is to use a voltage variable capacitance, such as a bipolar transistor junction in back bias, as the delay adjust element.
Similar to the amplitude control loop the delay can be implemented with fine and crude adjustments, with the crude steps adjusted by discrete means and the fine adjust controlled by means of a large voltage to control a fine delay adjust.
This is the most noise immune approach. Again requirement for the crude adjust, and the number of levels of crude adjust will be determined by the precision of the application and the tolerances of the IC process used.
A more detailed embodiment of the invention is shown in Figure 4 with examples of realizable approaches on an IC
process.
Important aspects of the invention are as follows:
a) A method to adjust the gain of the closed loop to a high level of tolerance this requires as a minimum - a variable gain amplifier - a method to produce a signal, Vm, related to the signal strength of the output signal - method to filter the noise bandwidth of the signal Vm - a method to generate a reference voltage - a method to compare the reference voltage with the filtered Vm to produce an error signal Er - a method to integrate Er - a method to create an amplitude control signal, Ac, based upon the integrated value of Er - a method to adjust the gain of the amplifier, A, based upon the value of the control signal Ac.
b) A method to adjust the frequency or the phase of the KT
amplifier output to a high degree of tolerance the frequency control of the KT amplifier requires as a minimum - a method to divide the output signal to a lower frequency - a method to count the lower frequency over an interval of time, Tc - a method to compare the result of the count with the expected number of counts of the reference over the same period of time, - a method to limit the bandwidth of the comparison - a method to adjust the delay of the KT amplifier loop based upon the results of the band-limited comparison.
All of the above adjusts may need to be done in crude discrete steps and in fine vernier steps depending upon the application.
This does not change the generality of the invention.
Numerous modifications, variations and adaptations may be made to the particular embodiments of the invention described above without departing from the scope of the invention, which is defined in the claims.
Claims (2)
1. An apparatus for adjusting the gain of a closed loop to a high level of tolerance, comprising:
a variable gain amplifier;
means for producing a signal, Vm, related to the signal strength of the output signal;
means for filtering the noise bandwidth of the signal Vm;
means for generating a reference voltage;
means for comparing the reference voltage with the filtered Vm to produce an error signal Er;
means for integrating Er;
means for creating an amplitude control signal, Ac, based upon the integrated value of Er; and a method to adjust the gain of the amplifier, A, based upon the value of the control signal Ac.
a variable gain amplifier;
means for producing a signal, Vm, related to the signal strength of the output signal;
means for filtering the noise bandwidth of the signal Vm;
means for generating a reference voltage;
means for comparing the reference voltage with the filtered Vm to produce an error signal Er;
means for integrating Er;
means for creating an amplitude control signal, Ac, based upon the integrated value of Er; and a method to adjust the gain of the amplifier, A, based upon the value of the control signal Ac.
2. A method for adjusting the frequency or the phase of a KT
amplifier output to a high degree of tolerance, comprising the steps of:
dividing the output signal to a lower frequency;
counting the lower frequency over an interval of time, Tc;
comparing the result of the count with the expected number of counts of the reference over the same period of time;
limiting the bandwidth of the comparison; and adjusting the delay of the KT amplifier loop based upon the results of the band-limited comparison.
amplifier output to a high degree of tolerance, comprising the steps of:
dividing the output signal to a lower frequency;
counting the lower frequency over an interval of time, Tc;
comparing the result of the count with the expected number of counts of the reference over the same period of time;
limiting the bandwidth of the comparison; and adjusting the delay of the KT amplifier loop based upon the results of the band-limited comparison.
Priority Applications (7)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA002224261A CA2224261A1 (en) | 1997-12-09 | 1997-12-09 | Low phase noise, high q, high gain amplifier in an integrated circuit |
| PCT/CA1998/001123 WO1999030416A1 (en) | 1997-12-09 | 1998-12-09 | An amplifier for continuous high gain, narrowband signal amplification |
| JP52955999A JP4299372B2 (en) | 1997-12-09 | 1998-12-09 | Amplifier for signal amplification in a narrow frequency band with continuous high gain |
| DE19882089T DE19882089B4 (en) | 1997-12-09 | 1998-12-09 | Amplifier for continuous narrow-band signal amplification with high amplification factor and amplification method |
| AU14777/99A AU1477799A (en) | 1997-12-09 | 1998-12-09 | An amplifier for continuous high gain, narrowband signal amplification |
| US09/209,051 US6057735A (en) | 1997-12-09 | 1998-12-09 | Amplifier for continuous high gain, narrowband signal amplification |
| CA002280878A CA2280878C (en) | 1997-12-09 | 1998-12-09 | An amplifier for continuous high gain, narrowband signal amplification |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA002224261A CA2224261A1 (en) | 1997-12-09 | 1997-12-09 | Low phase noise, high q, high gain amplifier in an integrated circuit |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2224261A1 true CA2224261A1 (en) | 1999-06-09 |
Family
ID=4161869
Family Applications (2)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002224261A Abandoned CA2224261A1 (en) | 1997-12-09 | 1997-12-09 | Low phase noise, high q, high gain amplifier in an integrated circuit |
| CA002280878A Expired - Fee Related CA2280878C (en) | 1997-12-09 | 1998-12-09 | An amplifier for continuous high gain, narrowband signal amplification |
Family Applications After (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002280878A Expired - Fee Related CA2280878C (en) | 1997-12-09 | 1998-12-09 | An amplifier for continuous high gain, narrowband signal amplification |
Country Status (6)
| Country | Link |
|---|---|
| US (1) | US6057735A (en) |
| JP (1) | JP4299372B2 (en) |
| AU (1) | AU1477799A (en) |
| CA (2) | CA2224261A1 (en) |
| DE (1) | DE19882089B4 (en) |
| WO (1) | WO1999030416A1 (en) |
Families Citing this family (22)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2000022450A (en) * | 1998-06-30 | 2000-01-21 | Omron Corp | Device and method for receiving signal, transmission/ reception device and network system |
| FI105609B (en) * | 1998-10-27 | 2000-09-15 | Nokia Mobile Phones Ltd | Method and arrangement for generating a transmission signal |
| CA2289345C (en) * | 1998-11-09 | 2002-10-29 | Mark Miles Cloutier | Inverted super regenerative receiver |
| US6278331B1 (en) * | 1999-03-30 | 2001-08-21 | Infineon Technologies Ag | System and method for compensating wafer parameters |
| KR100595839B1 (en) * | 1999-10-04 | 2006-07-05 | 에스케이 텔레콤주식회사 | Apparatus and method for compensating rf module gain using measured rf module noise |
| EP1229643A1 (en) * | 2001-02-02 | 2002-08-07 | Semiconductor Ideas to The Market (ItoM) BV | A receiver with AGC controlled resonant amplifier |
| WO2003052936A1 (en) * | 2001-12-19 | 2003-06-26 | Tait Electronics Limited | Improvements relating to frequency synthesis |
| US7400190B2 (en) * | 2002-09-05 | 2008-07-15 | Nxp B.V. | Self calibration of continuous-time filters and systems comprising such filters |
| US7429891B2 (en) * | 2003-02-14 | 2008-09-30 | Broadcom Corporation | Method and system for low noise amplifier (LNA) gain adjustment through narrowband received signal strength indicator (NRSSI) |
| JP4406378B2 (en) * | 2005-02-23 | 2010-01-27 | 株式会社ルネサステクノロジ | Transmitter and mobile communication terminal using the same |
| US7961060B1 (en) * | 2007-06-19 | 2011-06-14 | Cypress Semiconductor Corporation | Amplitude regulated resonant oscillator with sampled feedback |
| JP5524678B2 (en) * | 2010-03-31 | 2014-06-18 | 旭化成エレクトロニクス株式会社 | Recursive filter circuit |
| US8410844B2 (en) | 2010-03-31 | 2013-04-02 | Asahi Kasei Microdevices Corporation | Filter device and control method of the same |
| GB2494652A (en) * | 2011-09-14 | 2013-03-20 | Stephen Anthony Gerard Chandler | A transmitter amplifier with linearising RF feedback and automatic control of resonator Q, resonator frequency, loop phase, and loop gain |
| RU2738030C2 (en) | 2015-11-23 | 2020-12-07 | Анлотек Лимитед | Tunable filter |
| US9900197B1 (en) * | 2016-10-03 | 2018-02-20 | Keyssa Systems, Inc. | BPSK demodulation |
| US10439558B2 (en) * | 2017-04-28 | 2019-10-08 | Skyworks Solutions, Inc. | Apparatus and methods for power amplifiers with positive envelope feedback |
| EP3635865A1 (en) | 2017-05-24 | 2020-04-15 | Anlotek Limited | Apparatus and method for controlling a resonator |
| US11277110B2 (en) | 2019-09-03 | 2022-03-15 | Anlotek Limited | Fast frequency switching in a resonant high-Q analog filter |
| EP4070171A1 (en) | 2019-12-05 | 2022-10-12 | Anlotek Limited | Use of stable tunable active feedback analog filters in frequency synthesis |
| EP3926828A1 (en) | 2020-06-15 | 2021-12-22 | Anlotek Limited | Tunable bandpass filter with high stability and orthogonal tuning |
| US11955942B2 (en) | 2021-02-27 | 2024-04-09 | Anlotek Limited | Active multi-pole filter |
Family Cites Families (10)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US3588725A (en) * | 1969-05-27 | 1971-06-28 | Westinghouse Electric Corp | Q-invariant active resonator |
| JPS6058604B2 (en) * | 1977-04-18 | 1985-12-20 | 株式会社日立製作所 | Inclined variable equalizer |
| JPS5696507A (en) * | 1979-12-15 | 1981-08-04 | Matsushita Electric Works Ltd | Superregenerative receiver |
| NL8602894A (en) * | 1986-11-14 | 1988-06-01 | Philips Nv | FILTER SWITCHING. |
| CA2014968C (en) * | 1989-04-20 | 1999-12-14 | Nobukazu Hosoya | Active filter circuit |
| KR920010383B1 (en) * | 1990-10-23 | 1992-11-27 | 삼성전자 주식회사 | Homodyne tv receiver |
| JP3024297B2 (en) * | 1991-09-12 | 2000-03-21 | 日本電気株式会社 | Phase synchronous receiving circuit |
| JP2966226B2 (en) * | 1993-02-17 | 1999-10-25 | 三菱電機株式会社 | Automatic power amplifier control circuit |
| US5678213A (en) * | 1994-09-30 | 1997-10-14 | Lucent Technologies Inc. | Radio receiver for processing a multi-carrier signal with a large dynamic range |
| JPH09331222A (en) * | 1996-06-11 | 1997-12-22 | Nec Corp | Correcting device for gain control signal |
-
1997
- 1997-12-09 CA CA002224261A patent/CA2224261A1/en not_active Abandoned
-
1998
- 1998-12-09 US US09/209,051 patent/US6057735A/en not_active Expired - Lifetime
- 1998-12-09 AU AU14777/99A patent/AU1477799A/en not_active Abandoned
- 1998-12-09 DE DE19882089T patent/DE19882089B4/en not_active Expired - Lifetime
- 1998-12-09 CA CA002280878A patent/CA2280878C/en not_active Expired - Fee Related
- 1998-12-09 JP JP52955999A patent/JP4299372B2/en not_active Expired - Lifetime
- 1998-12-09 WO PCT/CA1998/001123 patent/WO1999030416A1/en active Application Filing
Also Published As
| Publication number | Publication date |
|---|---|
| CA2280878C (en) | 2002-02-05 |
| CA2280878A1 (en) | 1999-06-17 |
| JP4299372B2 (en) | 2009-07-22 |
| AU1477799A (en) | 1999-06-28 |
| JP2001512650A (en) | 2001-08-21 |
| US6057735A (en) | 2000-05-02 |
| DE19882089T1 (en) | 2000-03-16 |
| WO1999030416A1 (en) | 1999-06-17 |
| DE19882089B4 (en) | 2009-04-23 |
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