CA2415510A1 - Operating device for gas discharge lamps - Google Patents
Operating device for gas discharge lamps Download PDFInfo
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- CA2415510A1 CA2415510A1 CA002415510A CA2415510A CA2415510A1 CA 2415510 A1 CA2415510 A1 CA 2415510A1 CA 002415510 A CA002415510 A CA 002415510A CA 2415510 A CA2415510 A CA 2415510A CA 2415510 A1 CA2415510 A1 CA 2415510A1
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- Prior art keywords
- voltage
- operating device
- diode
- bridge
- current
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Classifications
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- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
- H05B41/14—Circuit arrangements
- H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
- H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
- H05B41/282—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices
- H05B41/2825—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices by means of a bridge converter in the final stage
-
- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
- H05B41/14—Circuit arrangements
- H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
- H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y10—TECHNICAL SUBJECTS COVERED BY FORMER USPC
- Y10S—TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y10S315/00—Electric lamp and discharge devices: systems
- Y10S315/07—Starting and control circuits for gas discharge lamp using transistors
Abstract
Free-running half-bridge inverter (HP) for operating gas discharge lamps having a current transformer as a feedback device. The half-bridge transistors (T1, T2) are essentially voltage controlled transistors (MOSFET). The drive circuits (1, 2) for the half-bridge transistors (T1, T2) contain a voltage threshold value switch (D2, R2) which, on reaching its voltage threshold, essentially carries a current which is proportional to the load current of the half-bridge inverter (HB).
Description
Patent-Treuhand-Gesellschaft fur elektrische Gluhlampen mbH., Munich Title Operating device for gas discharge lamps Technical Field The invention relates to an operating device for gas discharge lamps as claimed in the precharacterizing clause of claim 1. This relates ir_ particular to an improvement to the half-bridge inverter contained in the operating device, and tc> its drive. The invention furthermore relates to simplification of a switching-off device for the operating device, and to low-cost power factor correction for the current drawn from the mains.
Background art The document E° 0 093 469 (De Bi;il) describes an operating device for gas discharge lamps, which represents the prior art. This operating device contains a free-running half-bridge inverter, which uses a DC voltage to produce a high-frequency AC
voltage by switching an upper and a lower half-bridge transistor, which are connected in series, on and off alternatively. The DC voltage is generally produced by means of a bridge rectifier, comprising four rectifier diodes, from the mains voltage. In this cor_text, free-running means that the drive for the half-bridge transistors is obtair_ed from a load circuit, and that no independently osvillat__ng oscillator circuit is provided to producf~ said drive. Said drive is preferably oi~tained by means of a current transformer.
A primary winding of the current 'transformer is arranged in the 1 oad cv~rcu:it and a load current flows through it which i s essentially equivalent to the load current, which can essentially be equated to the current which is emitted fro::n the half-bridge inverter.
One secondary winding of the current transformer is arranged in each of two drive circuits, which each produce a signal which is supplied to the control electrodes of the half-bridge transistors. The load circuit is connected to the connection point of the half-bridge transistor. The main component oi= the load iC circuit is a lamp inductoz-, to which gas discharge lamps can be connected in series, via terminal connections. It is also possible to cor_nect a number of load circuits in parallel; the primary winding can then be arranged such that the ~~otal current from all the load circuits flows through :it.
Each of the drive cir_cui.ts produces a feedback signal, which is essentially proportional to the load current.
Ideally, the secondary windings must. be short-circuited for this purpose, but in practice they are terminated with a low impedanr_e. Otherwise, either saturation phenomena would occur in the current transistor or the primary winding wou'd have an. undesirably strong influence on the load circuit. According to the prior art, bipolar transist=ors are used for the half-bridge transistors, drawing their drive from the secondary windings. The base conr:ection of the bipolar transistors, which is used as a control electrode, naturally has a sufficiently low impedance to avoid the abovementioned effects.
The voltage drop across the secondary windings in the abovemer:tioned condit:~.ors represents a measure of the load current and, in the prior art, forms feedback signals. These are v~ each case supplied to a timer which, in the simplest case, comprises a timing capacitor and a timing resistor connected in series. If the respective timir:g capacitor is charged to an integration value wr_ich is cuff icient to drive a switching-off transistor, the respective half-bridge transistor is switched off.
A resonance capacitor, which together with the lamp inductor forms a resonance circuit, is effectively connected in parall.e~_ with a gas discharge lamp and in series with the lamp inductor, in particular in order to start gas discharge lamp's. This resonance circuit is operated close to its resonance point for starting, 0 thus resulting in a voltage which is sufficiently high to start a gas d:~scharge lamp being formed across the resonant capacitor.
A high current is accordingly formed in the lamp inductor and thus in the half-bridge transistors. In order to avoid components being overloaded, the amplitude of the load current is limited in the prior art. This is done via in each case one first voltage threshold value switch, which is connected in parallel with the respec~ive~ timing resistor. If the load current rises above a predetermined level, then the respective feedback signal reaches a value which can break through the respective first voltage threshold value switch, thus leading t.o the respective half-bridge transistor being switched off immediately.
Desclosure of the Invention The object or the present invention is to provide an operating device for gas discharge lamps as claimed in the precharacterizing clause of claim l, which makes the topology described in the prior art feasible not only for half bridges with bipolar transistors, which require a drive current of course, but also allows voltage controlled semiconductor switches such as MOF
field-effect transisvors MOSFET;~ to be used. The object on ;~rh,_ch this problem i~; based essentially includes the provision of a drive signal for the semiconductor switches which is proportional to the load current.
This object i.s achieved by an operating device for gas discharge lamps raving the features of the precharacterizing clause of claim 1 and by mE~ans of the features of the characterizing part of claim 1.
Particularly advantageous refinements can be found in the dependent claims.
Bipolar transistors are increasingly being replaced by voltage controlled semiconductor switches such as MOSFETs and IGBTs, mainly for cost reasons.
If one of the secondary windings described above is used to drive a voltage controlled semiconductor switch rather than a bipolar transistor, then the termination of the secondary winding no longer has a low impedance but a high impedance, and the disadvantages mentioned in the section relating to the prior art occur.
According to the invention, the drive circuits are each equipped with a second voltage threshold value switch, which has a second. voltage threshold and is connected in parallel with the secondary winding. In the simplest case, the second voltagE= threshold value switch comprises a zener diode and a current measurement resistor conr_ected in series, with the zener diode having a zener voltage which corresponds to the second voltage threshold. If the voltage across the secondary winding rises, starting from zero, then the second voltage threshold value switch initially has no effect.
On reaching the second voltage threshold, the zener diode starts to conduct, and the secondary winding is terminated with a low impedance, as desired. The value of the second voltage threshold must be lower than a threshold voltage which the voltage controlled semiconductor switch requires, as a minimum, as a drive. The size of th~~ current measurement resistor has to satisfy two cond;.~dons . F ins tly, the value of the current measurement resistor mint be small enough to ensure a low-impedance termination on the secondary winding. Secondly, the value of the current measurement resistor must be hich enough to allow the voltage across the secondary winding to rise further as far as the first voltage threshold.
Since a current whicr: is essentially proportional to the load current flows in the current measurement resistor according to the invention, the voltage across the current measuremer:t resistor is, of course, also a measure of the load current. The voltage across the current measurement resistor may thus be used, according to the invention, in order to detect a fault '~5 situation. For this purpose, it is supplied to a switching-off device. In order to suppress interference, the time average of the voltage across the current measurement resistor is formed in the switching-off device. If this exceeds a given limit value, the switching-off device prevents further oscillation of the ha~'yf-bridge :inverter. This is done in particular by suppressing the drive signal for one of the two half-bridge tr_ans.istors.
The operating devices under discussion generally have two mains voltage terminals which can be connected to a mains voltage, thus allowincx a mair_s current. to flow.
Relevant standards (for example: IE:C 1000-3-2) specify maximum amplitudes for the harmonics in the mains current. In order to comply with these Standards, operating devices hare so-called PFC circuits (Power Factor Correction) . One lc;w-cost imp'~ementation for these PFC circuits is represented by so-called pumping circuits, as are described, for example, in EP 253 224 ~5 (Zuchtriegel) or EP 1 028 606 (Rudolph). If a pumping circuit is combined with a free-running half-bridge inverter according to she prior art, this leads to problems in producing the necessary starting voltage for the gas discarge lamps, and problems due to the high power losses during sw;~tching of the half-bridge transistors. Said problems c;ccur in particular in the case of high-power gas discharge lamps. One reason for this, inter alia, is the storage times, which are typical for bipolar transistors and do not allow the switching-off time to be defined exactly. The present invention allows the use of voltage controlled semiconductor switches such as MOSFETs, which have no storage times and therefore allow ~~aid problems to be avoided. This means that the half-bridge inverter according to the invention in conjunction with a pumping circuit can be used advantageously even for a load which consumes a power of more than 100 W.
A further effect which occurs in the case of the half-bridge inverter according to the invention with a pumping circuit is the heavy modulation of the operating frequency by the mains voltage, which is subject to the oscillation of the half-bridge inverter.
Depending on the instantaneous value of the mains voltage, said operating frequency is within a frequency band which has a bandwidth of more than 10 kHz. The electromagnetic interference caused by an operating device according to ~:~e invention is thus distributed over a wide frequency band. The amount of energy reaching an appliar_ce that is subject to interference is thus advantageously low. Furthermore, the complexity for suppression of an operating device according to the invention can be kept low.
A further advantageous application cf the current measurement resistor according to t;he invention is in the starting circuit for the free-running ~~alf-bridge inverter. ~n order t~~ start the half-bridge inverter, the normal process is t.o charge a starting capacitor and, when a trigger voltage is reac_~ed across the charge-storage capacit~r, to discharge a portion of the charge stored in the charcfe-storage capacitor via a trigger element ~o the control electrode of a half-bridge capacitor. In this case, one problem that can occur is that the charge pulse produced in this way at the relevant control electrode is too short and too small, and continued oscillation of the half-bridge inverter is no~ tri_ggHred. According to the invention, a portion of the stored charge in the charging capacitor is supplied via a diode to th.e current measurement resistor accor_d:ing to the invention. This makes it possible to ensure that the half-bridge inverter starts to oscillate reliably.
Brief Description of the Drawings The invention will be explained in more detail in the ,.~5 following text with reference to exc=_rnplary embodiments.
In the figures:
Figure 1 shows the basic circuit of the operating device according to the invention, Figure 2 shows an exemplary embodiment of a drive circuit acccrding to the invention, Figure 3 shows an exemplary embodiment of an operating device according to the invention having a pumping circuit, and Figure 4 shows an exemplary embodiment of a switching-off device according to the invention.
In the following text, resistors are denoted by the letter R, transistors by the letter T, diodes by the letter D, capacitors by tf~e letter C and connecting terminals by the letter J, in each case followed by a number.
Best Made for carrying out the invention Figure 1 shows the basic circuit of an operating device according to the invention. The operating device can be connected to a mains voltage via the connecting terminals Jl, J2. The mains voltage is supplied to a block FR, which cc>nt:ains generally known filter and rectifier devices. The filter devices have the task of suppressing interference. The rectifier device 1G generally comprises a bridge rectifier having four diodes. The rectifier_ device is u:>ed to supply a DC
voltage to a half-bridge inverter ~iB. The half-bridge inverter essentially contains an upper semiconductor switch T1 and a lower semiconductor switch T2, which i5 are connected in series and, according to the invention, are voltage-controlled. The exemplary embodiment in Figure 1 uses N-channel MOSFETs. However, it is also possible to use, for example, IGBTs or P-channel MOSFETs. With the N-channel MOSFET used in 2C Figure l, the positive output of the rectifier device must be supplied via a node 3 to the upper transistor Tl, whil.e the negative output of the rectifier device is connected to the ground potential M. The same polarity is used for commercially available IGBTs, but 25 the opposite polarity must be used for P-channel MOSFETs.
An energy-storage capacitor C1 is connected between the node 3 and the ground potential M and temporarily 30 stores energy from the mains voltage, before it is emitted to a lamp LP.
In order to drive the half-bridge transistors T1, T2, the half-bridge inverter H3 contains a drive circuit l, 35 2 for each half-bridge transistor T1, T2. The drive circuits l, 2 are each connected via a connection A to the respective gate connection and via a connection B
to the respective sc"rce connection, of the relevant half-bridge trar:sistor. The drive circuit ? for the _ C. _ lower half-bridge transistor T2 has a third connection S, to which a switchir_g-off device can be connected.
The connection point c,f the half-b ridge transistors T1, T2 forms a node 4, to which <~ load circuit is conr_ected. A second connection of the load circuit in Figure 1 is connect:ed to 'he ground potential M. In an equivalent manner, tr:e second c:onr:ection of: the load circuit may alternatively be. connected to the node 3.
The load c;.~rcuit esser~.tially comprises a series circuit formed by a primary winding L2 of a current transformer, a lamp inductor Ll, a resonance capacitor C2 and a coupling capacitor C3. One or more series-connected lamp:: LP can be connected vi.a the lamp terminals J3, J4 in parallel with the resonance capacitor C2. In the exemplary embodiment, no provision is made for preheating the lamp filaments. However, generally known devices for fi:Lament heating are available to those skilled in the art, and c:an be used with the operating device according to the invention.
It is also possible to operate a number of load circuits connected in parallel. The function of the individual elements of the :Load circuit can be found in the prior art.
Figure 2 shows one preferred exemplary embodiment of a drive circuit according to the invention. A secondary winding L3 of the current transformer is connected between. a node 20 and the connection B, which is known from Figure 1. The anode of a diode D1 is connected to the node 20, and its cathode is connected to a node 21.
The node 21 is conr:ected via a resistor R3 to the connection A, whicr: is known from Figure 1. P.n integraticn element ~s connected in parallel with the ~5 secondary winding L3 and is in tie form of a timing resistor Rl and a aiming capacitor C4 connected in series, and has an integration constant which corresponds to the product of the values of R1 and C4.
The connect;.~on point or Rl and C4 forms a node 22. Pn integration value is tappecz off in parallel with C4, and is supplied to the control electrode of a semiconductor switch T3. 'rhe switching path of the semiconductor sw;~t:ch T3 is connected between the connections A and B . As in the exe~r.plary emx>odiment, a resistor R4 may be connected in parallel with this, in order to improve she switching reliability. The sem;_conductor switch f3 is preferably in the form of a small signal bipolar transistor.
A first voltage threshold value :;witch with a first voltage threshold is connected between the node 21 and the node 22, and i~~ in t:he form of a zener diode D3. If the vel~~age which is fed ;~ni=o the drive circuit from L3 exceeds a value which leads to the zener voltage of D3 being exceeded, then ~he timing capacitor C4 is charged not only via the ti_mlng resistor Rl but also via D3, so that the integration constani~ of the integration element is reduced.
According to the invention, a second voltage threshold value switch with a second voltage threshold is connected between the rode 21 and the connection B.
This is preferably formed by a zener diode D2 and a current measu=ement resistor R2 connected in series. If the vo~'ytage at L3 rises, the associated half-bridge transistor is first of all driven via the connection A.
After the voltage at R2 rises further, the zener voltage of D2 is, according to the invention, exceeded.
A current flow trerefcre occurs via the current measurement resistor R2, which is essentially proportional to ~he load current ._n the load circuit.
This prevents the current transformer from being saturated, and the integration ele_nent is charged in proportion to the load current. If the current in the load circuit becomes so great that the zener voltage of D3 is exceeded, then this leads to the associated half-bridge transistor being switched off quickly.
- 1.L -One connection S i.s connected to the connection point between D2 and the current measurement resistor R2. A
voltage which is proportional to true load current can be tapped off between the connection S and the connection B and can be supplied to a switching-off device, as described below. Since the voltages in the switching-off device are in general related to the ground potential M, cnly the drive circuit associated with the lower half-bridge transistor has a connection i0 S.
The following tab'~e summarizes the preferred sizes of the components illustrated in Figure 2.
Component Value D2 5.6V
Rl 1 . 8kS2 R4 2.2kSZ
C4 lOnF
In Figure 3, the half-bridge converter HB according to the invention is prov-~ded in an operating device with a pumping circuit, as l.; described in Figures I and 2. In contrast to Figure l, the positive output of the rectifier device in the block FR is not connected directly to the node '?, but via two parallel-connected series circui ts, each havin~~ twc> diodes . A first diode series circuit with ~ f;~rst= diode connection point is formed by the diodes D5 and D6. A second diode series circuit with a second diode connect=ion point is formed by the diodes D4 and D7. I>ifferenis nodes of the load circuit whic_~ is known from Figure ~. are connected to the diode connection points via reactive two-pole networks.
The lamp terminal J3 is conr_ected to the first diode connection po;~nt via a pumping capacitor C6. The lamp Ltd -terminal J3 is distinguished. from the lamp terminal J4 in that the value of th.e amplitude of its AC voltage component with respect to the ground potential is higher. The resonance capacitor C2 from Figure 1 is omitted. Its function is carried out by the pumping capacitor C6.
The connection point of the primary winding L2 and of the lamp inductor Ll is connected ~_o the second diode 1C connection point via a pumping =inductor L4 and a capacitor C7 connected in series. However, the pumping inductor L4 may a~.~so be connected d,~rectly to the node 4, which is known from Figure 1 and represents the connection point of t2~e hal==-bridge transistors T1 and T2. The capacitor C7 is es~;entially used for blocking any DC component in the current t=hrough the pumping inductor L4.
The node 4, which is known from Figure i, is connected to the first diode connection point via a second pumping capacitor C5.
Figure 3 shows a pumping circuit structure having three so-called pumping branches: one pumping branch is represented by the pumping capacitc;r C6, a further by the second pumping capacitor C5, <~nd a third by the pumping inductor L4 . Each pumping branch intrinsically already acts as a PFC
circuit, so that i_t is not always necessary for al's three pumping branches to be provided. In fact, any desired combination of the pumping branches z.s possible.
A further variation option relates 1.o the diodes D5 and D7. These diodes may also carry oat functions which are associated with the rectifier device in the block FR.
Corresponding diodes in the rectifier device can then be omitted.
Figure 4 shows how the current measurement resistor R2 according to the invention and the connection S
connected to it from Figure 2 ca:~ advantageously be used for a switching-off device and a starting device for the operating device.
The switching-off device contains a generally known thyristor simulation c~ompri:~ing the resistors R42, R43, R44 and R45 and tha transistors T41. and T42. The tr.yristor simulation is connected to the rode 3 from Figure 1 via a resistor F?41. The other end of the thyristor simulation is connected to ground potential M.
A voltage wr.ich is proportional to the load current is fed via the connection S into a voltage divider comprising the resi:~tors R46 and R47. The voltage divider divides the voltage that is fed in to a value which normally does not cause t:he operating device to be switched off. The time average of the load current is formed by a capacitor C40, wh-_ch is fed from the voltage divider, and is provided in the form of a voltage related to ground potential. This voltage is supplied to the control e=_ectrode of a semiconductor switch, which is in the form of a bipolar transistor T43. If the mean value of the load current exceeds a predetermined level ~.n the event of a fault., then the thyristor simulation is triggered via the collector connection of T43. A connection G2, which is connected to the control electrode of the lower half-bridge transistor, i.s in conseauer~ce connected via a diode D42 to ground potential M.. This prevents further oscillation of the half-bridge inverter.
The half-bridge inve-ter starts to oscillate with the aid of a generally known starting capacitor C91, which is charged from the r<.ains voltage via the resistor R41.
C41 is connected to a trigger diode D40 (~IAC). When the voltage c:n C4~ reaches the trigger voltage of the trigger diode D4G, the control electrode of the lower half-bridge transistor has a starting pulse applied to it via a diode D41 and the connect_ on G2 . In practice, the starting pulse may turn out to be short s~o that the half-bridge inverter_ does not reliably start to oscillate. The ccrnec:t~ion S is therefore advantageously used: according to the invention, the connection S is connected to the ~rigcxer diode D4G via a diode D43. The starting pulse passes not only via the diode D41 but, according to 'he invention, also vz.a the diode D43 and then via the diode D2 and the resistor R3 from Figure 2. The starting pulse is thus lengthened and enlarged, thus ~.~eading to the calf-bridge inverter starting to oscillate reliably.
Background art The document E° 0 093 469 (De Bi;il) describes an operating device for gas discharge lamps, which represents the prior art. This operating device contains a free-running half-bridge inverter, which uses a DC voltage to produce a high-frequency AC
voltage by switching an upper and a lower half-bridge transistor, which are connected in series, on and off alternatively. The DC voltage is generally produced by means of a bridge rectifier, comprising four rectifier diodes, from the mains voltage. In this cor_text, free-running means that the drive for the half-bridge transistors is obtair_ed from a load circuit, and that no independently osvillat__ng oscillator circuit is provided to producf~ said drive. Said drive is preferably oi~tained by means of a current transformer.
A primary winding of the current 'transformer is arranged in the 1 oad cv~rcu:it and a load current flows through it which i s essentially equivalent to the load current, which can essentially be equated to the current which is emitted fro::n the half-bridge inverter.
One secondary winding of the current transformer is arranged in each of two drive circuits, which each produce a signal which is supplied to the control electrodes of the half-bridge transistors. The load circuit is connected to the connection point of the half-bridge transistor. The main component oi= the load iC circuit is a lamp inductoz-, to which gas discharge lamps can be connected in series, via terminal connections. It is also possible to cor_nect a number of load circuits in parallel; the primary winding can then be arranged such that the ~~otal current from all the load circuits flows through :it.
Each of the drive cir_cui.ts produces a feedback signal, which is essentially proportional to the load current.
Ideally, the secondary windings must. be short-circuited for this purpose, but in practice they are terminated with a low impedanr_e. Otherwise, either saturation phenomena would occur in the current transistor or the primary winding wou'd have an. undesirably strong influence on the load circuit. According to the prior art, bipolar transist=ors are used for the half-bridge transistors, drawing their drive from the secondary windings. The base conr:ection of the bipolar transistors, which is used as a control electrode, naturally has a sufficiently low impedance to avoid the abovementioned effects.
The voltage drop across the secondary windings in the abovemer:tioned condit:~.ors represents a measure of the load current and, in the prior art, forms feedback signals. These are v~ each case supplied to a timer which, in the simplest case, comprises a timing capacitor and a timing resistor connected in series. If the respective timir:g capacitor is charged to an integration value wr_ich is cuff icient to drive a switching-off transistor, the respective half-bridge transistor is switched off.
A resonance capacitor, which together with the lamp inductor forms a resonance circuit, is effectively connected in parall.e~_ with a gas discharge lamp and in series with the lamp inductor, in particular in order to start gas discharge lamp's. This resonance circuit is operated close to its resonance point for starting, 0 thus resulting in a voltage which is sufficiently high to start a gas d:~scharge lamp being formed across the resonant capacitor.
A high current is accordingly formed in the lamp inductor and thus in the half-bridge transistors. In order to avoid components being overloaded, the amplitude of the load current is limited in the prior art. This is done via in each case one first voltage threshold value switch, which is connected in parallel with the respec~ive~ timing resistor. If the load current rises above a predetermined level, then the respective feedback signal reaches a value which can break through the respective first voltage threshold value switch, thus leading t.o the respective half-bridge transistor being switched off immediately.
Desclosure of the Invention The object or the present invention is to provide an operating device for gas discharge lamps as claimed in the precharacterizing clause of claim l, which makes the topology described in the prior art feasible not only for half bridges with bipolar transistors, which require a drive current of course, but also allows voltage controlled semiconductor switches such as MOF
field-effect transisvors MOSFET;~ to be used. The object on ;~rh,_ch this problem i~; based essentially includes the provision of a drive signal for the semiconductor switches which is proportional to the load current.
This object i.s achieved by an operating device for gas discharge lamps raving the features of the precharacterizing clause of claim 1 and by mE~ans of the features of the characterizing part of claim 1.
Particularly advantageous refinements can be found in the dependent claims.
Bipolar transistors are increasingly being replaced by voltage controlled semiconductor switches such as MOSFETs and IGBTs, mainly for cost reasons.
If one of the secondary windings described above is used to drive a voltage controlled semiconductor switch rather than a bipolar transistor, then the termination of the secondary winding no longer has a low impedance but a high impedance, and the disadvantages mentioned in the section relating to the prior art occur.
According to the invention, the drive circuits are each equipped with a second voltage threshold value switch, which has a second. voltage threshold and is connected in parallel with the secondary winding. In the simplest case, the second voltagE= threshold value switch comprises a zener diode and a current measurement resistor conr_ected in series, with the zener diode having a zener voltage which corresponds to the second voltage threshold. If the voltage across the secondary winding rises, starting from zero, then the second voltage threshold value switch initially has no effect.
On reaching the second voltage threshold, the zener diode starts to conduct, and the secondary winding is terminated with a low impedance, as desired. The value of the second voltage threshold must be lower than a threshold voltage which the voltage controlled semiconductor switch requires, as a minimum, as a drive. The size of th~~ current measurement resistor has to satisfy two cond;.~dons . F ins tly, the value of the current measurement resistor mint be small enough to ensure a low-impedance termination on the secondary winding. Secondly, the value of the current measurement resistor must be hich enough to allow the voltage across the secondary winding to rise further as far as the first voltage threshold.
Since a current whicr: is essentially proportional to the load current flows in the current measurement resistor according to the invention, the voltage across the current measuremer:t resistor is, of course, also a measure of the load current. The voltage across the current measurement resistor may thus be used, according to the invention, in order to detect a fault '~5 situation. For this purpose, it is supplied to a switching-off device. In order to suppress interference, the time average of the voltage across the current measurement resistor is formed in the switching-off device. If this exceeds a given limit value, the switching-off device prevents further oscillation of the ha~'yf-bridge :inverter. This is done in particular by suppressing the drive signal for one of the two half-bridge tr_ans.istors.
The operating devices under discussion generally have two mains voltage terminals which can be connected to a mains voltage, thus allowincx a mair_s current. to flow.
Relevant standards (for example: IE:C 1000-3-2) specify maximum amplitudes for the harmonics in the mains current. In order to comply with these Standards, operating devices hare so-called PFC circuits (Power Factor Correction) . One lc;w-cost imp'~ementation for these PFC circuits is represented by so-called pumping circuits, as are described, for example, in EP 253 224 ~5 (Zuchtriegel) or EP 1 028 606 (Rudolph). If a pumping circuit is combined with a free-running half-bridge inverter according to she prior art, this leads to problems in producing the necessary starting voltage for the gas discarge lamps, and problems due to the high power losses during sw;~tching of the half-bridge transistors. Said problems c;ccur in particular in the case of high-power gas discharge lamps. One reason for this, inter alia, is the storage times, which are typical for bipolar transistors and do not allow the switching-off time to be defined exactly. The present invention allows the use of voltage controlled semiconductor switches such as MOSFETs, which have no storage times and therefore allow ~~aid problems to be avoided. This means that the half-bridge inverter according to the invention in conjunction with a pumping circuit can be used advantageously even for a load which consumes a power of more than 100 W.
A further effect which occurs in the case of the half-bridge inverter according to the invention with a pumping circuit is the heavy modulation of the operating frequency by the mains voltage, which is subject to the oscillation of the half-bridge inverter.
Depending on the instantaneous value of the mains voltage, said operating frequency is within a frequency band which has a bandwidth of more than 10 kHz. The electromagnetic interference caused by an operating device according to ~:~e invention is thus distributed over a wide frequency band. The amount of energy reaching an appliar_ce that is subject to interference is thus advantageously low. Furthermore, the complexity for suppression of an operating device according to the invention can be kept low.
A further advantageous application cf the current measurement resistor according to t;he invention is in the starting circuit for the free-running ~~alf-bridge inverter. ~n order t~~ start the half-bridge inverter, the normal process is t.o charge a starting capacitor and, when a trigger voltage is reac_~ed across the charge-storage capacit~r, to discharge a portion of the charge stored in the charcfe-storage capacitor via a trigger element ~o the control electrode of a half-bridge capacitor. In this case, one problem that can occur is that the charge pulse produced in this way at the relevant control electrode is too short and too small, and continued oscillation of the half-bridge inverter is no~ tri_ggHred. According to the invention, a portion of the stored charge in the charging capacitor is supplied via a diode to th.e current measurement resistor accor_d:ing to the invention. This makes it possible to ensure that the half-bridge inverter starts to oscillate reliably.
Brief Description of the Drawings The invention will be explained in more detail in the ,.~5 following text with reference to exc=_rnplary embodiments.
In the figures:
Figure 1 shows the basic circuit of the operating device according to the invention, Figure 2 shows an exemplary embodiment of a drive circuit acccrding to the invention, Figure 3 shows an exemplary embodiment of an operating device according to the invention having a pumping circuit, and Figure 4 shows an exemplary embodiment of a switching-off device according to the invention.
In the following text, resistors are denoted by the letter R, transistors by the letter T, diodes by the letter D, capacitors by tf~e letter C and connecting terminals by the letter J, in each case followed by a number.
Best Made for carrying out the invention Figure 1 shows the basic circuit of an operating device according to the invention. The operating device can be connected to a mains voltage via the connecting terminals Jl, J2. The mains voltage is supplied to a block FR, which cc>nt:ains generally known filter and rectifier devices. The filter devices have the task of suppressing interference. The rectifier device 1G generally comprises a bridge rectifier having four diodes. The rectifier_ device is u:>ed to supply a DC
voltage to a half-bridge inverter ~iB. The half-bridge inverter essentially contains an upper semiconductor switch T1 and a lower semiconductor switch T2, which i5 are connected in series and, according to the invention, are voltage-controlled. The exemplary embodiment in Figure 1 uses N-channel MOSFETs. However, it is also possible to use, for example, IGBTs or P-channel MOSFETs. With the N-channel MOSFET used in 2C Figure l, the positive output of the rectifier device must be supplied via a node 3 to the upper transistor Tl, whil.e the negative output of the rectifier device is connected to the ground potential M. The same polarity is used for commercially available IGBTs, but 25 the opposite polarity must be used for P-channel MOSFETs.
An energy-storage capacitor C1 is connected between the node 3 and the ground potential M and temporarily 30 stores energy from the mains voltage, before it is emitted to a lamp LP.
In order to drive the half-bridge transistors T1, T2, the half-bridge inverter H3 contains a drive circuit l, 35 2 for each half-bridge transistor T1, T2. The drive circuits l, 2 are each connected via a connection A to the respective gate connection and via a connection B
to the respective sc"rce connection, of the relevant half-bridge trar:sistor. The drive circuit ? for the _ C. _ lower half-bridge transistor T2 has a third connection S, to which a switchir_g-off device can be connected.
The connection point c,f the half-b ridge transistors T1, T2 forms a node 4, to which <~ load circuit is conr_ected. A second connection of the load circuit in Figure 1 is connect:ed to 'he ground potential M. In an equivalent manner, tr:e second c:onr:ection of: the load circuit may alternatively be. connected to the node 3.
The load c;.~rcuit esser~.tially comprises a series circuit formed by a primary winding L2 of a current transformer, a lamp inductor Ll, a resonance capacitor C2 and a coupling capacitor C3. One or more series-connected lamp:: LP can be connected vi.a the lamp terminals J3, J4 in parallel with the resonance capacitor C2. In the exemplary embodiment, no provision is made for preheating the lamp filaments. However, generally known devices for fi:Lament heating are available to those skilled in the art, and c:an be used with the operating device according to the invention.
It is also possible to operate a number of load circuits connected in parallel. The function of the individual elements of the :Load circuit can be found in the prior art.
Figure 2 shows one preferred exemplary embodiment of a drive circuit according to the invention. A secondary winding L3 of the current transformer is connected between. a node 20 and the connection B, which is known from Figure 1. The anode of a diode D1 is connected to the node 20, and its cathode is connected to a node 21.
The node 21 is conr:ected via a resistor R3 to the connection A, whicr: is known from Figure 1. P.n integraticn element ~s connected in parallel with the ~5 secondary winding L3 and is in tie form of a timing resistor Rl and a aiming capacitor C4 connected in series, and has an integration constant which corresponds to the product of the values of R1 and C4.
The connect;.~on point or Rl and C4 forms a node 22. Pn integration value is tappecz off in parallel with C4, and is supplied to the control electrode of a semiconductor switch T3. 'rhe switching path of the semiconductor sw;~t:ch T3 is connected between the connections A and B . As in the exe~r.plary emx>odiment, a resistor R4 may be connected in parallel with this, in order to improve she switching reliability. The sem;_conductor switch f3 is preferably in the form of a small signal bipolar transistor.
A first voltage threshold value :;witch with a first voltage threshold is connected between the node 21 and the node 22, and i~~ in t:he form of a zener diode D3. If the vel~~age which is fed ;~ni=o the drive circuit from L3 exceeds a value which leads to the zener voltage of D3 being exceeded, then ~he timing capacitor C4 is charged not only via the ti_mlng resistor Rl but also via D3, so that the integration constani~ of the integration element is reduced.
According to the invention, a second voltage threshold value switch with a second voltage threshold is connected between the rode 21 and the connection B.
This is preferably formed by a zener diode D2 and a current measu=ement resistor R2 connected in series. If the vo~'ytage at L3 rises, the associated half-bridge transistor is first of all driven via the connection A.
After the voltage at R2 rises further, the zener voltage of D2 is, according to the invention, exceeded.
A current flow trerefcre occurs via the current measurement resistor R2, which is essentially proportional to ~he load current ._n the load circuit.
This prevents the current transformer from being saturated, and the integration ele_nent is charged in proportion to the load current. If the current in the load circuit becomes so great that the zener voltage of D3 is exceeded, then this leads to the associated half-bridge transistor being switched off quickly.
- 1.L -One connection S i.s connected to the connection point between D2 and the current measurement resistor R2. A
voltage which is proportional to true load current can be tapped off between the connection S and the connection B and can be supplied to a switching-off device, as described below. Since the voltages in the switching-off device are in general related to the ground potential M, cnly the drive circuit associated with the lower half-bridge transistor has a connection i0 S.
The following tab'~e summarizes the preferred sizes of the components illustrated in Figure 2.
Component Value D2 5.6V
Rl 1 . 8kS2 R4 2.2kSZ
C4 lOnF
In Figure 3, the half-bridge converter HB according to the invention is prov-~ded in an operating device with a pumping circuit, as l.; described in Figures I and 2. In contrast to Figure l, the positive output of the rectifier device in the block FR is not connected directly to the node '?, but via two parallel-connected series circui ts, each havin~~ twc> diodes . A first diode series circuit with ~ f;~rst= diode connection point is formed by the diodes D5 and D6. A second diode series circuit with a second diode connect=ion point is formed by the diodes D4 and D7. I>ifferenis nodes of the load circuit whic_~ is known from Figure ~. are connected to the diode connection points via reactive two-pole networks.
The lamp terminal J3 is conr_ected to the first diode connection po;~nt via a pumping capacitor C6. The lamp Ltd -terminal J3 is distinguished. from the lamp terminal J4 in that the value of th.e amplitude of its AC voltage component with respect to the ground potential is higher. The resonance capacitor C2 from Figure 1 is omitted. Its function is carried out by the pumping capacitor C6.
The connection point of the primary winding L2 and of the lamp inductor Ll is connected ~_o the second diode 1C connection point via a pumping =inductor L4 and a capacitor C7 connected in series. However, the pumping inductor L4 may a~.~so be connected d,~rectly to the node 4, which is known from Figure 1 and represents the connection point of t2~e hal==-bridge transistors T1 and T2. The capacitor C7 is es~;entially used for blocking any DC component in the current t=hrough the pumping inductor L4.
The node 4, which is known from Figure i, is connected to the first diode connection point via a second pumping capacitor C5.
Figure 3 shows a pumping circuit structure having three so-called pumping branches: one pumping branch is represented by the pumping capacitc;r C6, a further by the second pumping capacitor C5, <~nd a third by the pumping inductor L4 . Each pumping branch intrinsically already acts as a PFC
circuit, so that i_t is not always necessary for al's three pumping branches to be provided. In fact, any desired combination of the pumping branches z.s possible.
A further variation option relates 1.o the diodes D5 and D7. These diodes may also carry oat functions which are associated with the rectifier device in the block FR.
Corresponding diodes in the rectifier device can then be omitted.
Figure 4 shows how the current measurement resistor R2 according to the invention and the connection S
connected to it from Figure 2 ca:~ advantageously be used for a switching-off device and a starting device for the operating device.
The switching-off device contains a generally known thyristor simulation c~ompri:~ing the resistors R42, R43, R44 and R45 and tha transistors T41. and T42. The tr.yristor simulation is connected to the rode 3 from Figure 1 via a resistor F?41. The other end of the thyristor simulation is connected to ground potential M.
A voltage wr.ich is proportional to the load current is fed via the connection S into a voltage divider comprising the resi:~tors R46 and R47. The voltage divider divides the voltage that is fed in to a value which normally does not cause t:he operating device to be switched off. The time average of the load current is formed by a capacitor C40, wh-_ch is fed from the voltage divider, and is provided in the form of a voltage related to ground potential. This voltage is supplied to the control e=_ectrode of a semiconductor switch, which is in the form of a bipolar transistor T43. If the mean value of the load current exceeds a predetermined level ~.n the event of a fault., then the thyristor simulation is triggered via the collector connection of T43. A connection G2, which is connected to the control electrode of the lower half-bridge transistor, i.s in conseauer~ce connected via a diode D42 to ground potential M.. This prevents further oscillation of the half-bridge inverter.
The half-bridge inve-ter starts to oscillate with the aid of a generally known starting capacitor C91, which is charged from the r<.ains voltage via the resistor R41.
C41 is connected to a trigger diode D40 (~IAC). When the voltage c:n C4~ reaches the trigger voltage of the trigger diode D4G, the control electrode of the lower half-bridge transistor has a starting pulse applied to it via a diode D41 and the connect_ on G2 . In practice, the starting pulse may turn out to be short s~o that the half-bridge inverter_ does not reliably start to oscillate. The ccrnec:t~ion S is therefore advantageously used: according to the invention, the connection S is connected to the ~rigcxer diode D4G via a diode D43. The starting pulse passes not only via the diode D41 but, according to 'he invention, also vz.a the diode D43 and then via the diode D2 and the resistor R3 from Figure 2. The starting pulse is thus lengthened and enlarged, thus ~.~eading to the calf-bridge inverter starting to oscillate reliably.
Claims (8)
1. An operating device for operating gas discharge lamps, having the following features:
~ a free-running half-bridge inverter (HB) which contains two half-bridge transistors (T1, T2) connected in series, ~ a load circuit which is connected to the connection point between the half-bridge transistors (4) and which contains a primary winding (L2) of a current transformer through which a load current flows which is drawn from the half-bridge inverter (HB), ~ in each case one drive circuit (1, 2) for each half-bridge transistor (T1, T2), which in each case contains the following components:
- a secondary winding (L3) of the current transformer, - an integration element (R1, C4) which essentially integrates the voltage across the secondary winding (L3) of the current transformer and switches off the relevant half-bridge transistor on reaching a predetermined integration value, - a first voltage threshold value switch (D3), which reduces the integration constant of the integration element on reaching a given first voltage threshold, characterized in that ~ the half-bridge transistors (T1, T2) are essentially voltage controlled transistors, and ~ at least one drive circuit (1, 2) has a second voltage threshold value switch (D2, R2) with a second voltage threshold which is lower than the first voltage threshold, with the second voltage threshold value switch (D2, R2) being connected in parallel with the secondary winding (L3).
~ a free-running half-bridge inverter (HB) which contains two half-bridge transistors (T1, T2) connected in series, ~ a load circuit which is connected to the connection point between the half-bridge transistors (4) and which contains a primary winding (L2) of a current transformer through which a load current flows which is drawn from the half-bridge inverter (HB), ~ in each case one drive circuit (1, 2) for each half-bridge transistor (T1, T2), which in each case contains the following components:
- a secondary winding (L3) of the current transformer, - an integration element (R1, C4) which essentially integrates the voltage across the secondary winding (L3) of the current transformer and switches off the relevant half-bridge transistor on reaching a predetermined integration value, - a first voltage threshold value switch (D3), which reduces the integration constant of the integration element on reaching a given first voltage threshold, characterized in that ~ the half-bridge transistors (T1, T2) are essentially voltage controlled transistors, and ~ at least one drive circuit (1, 2) has a second voltage threshold value switch (D2, R2) with a second voltage threshold which is lower than the first voltage threshold, with the second voltage threshold value switch (D2, R2) being connected in parallel with the secondary winding (L3).
2. The operating device as claimed in claim 1, characterized in that the second voltage threshold value switch contains a zener diode (D2) and a current measurement resistor (R2) connected in series.
3. The operating device as claimed in claim 2, characterized in that the voltage across the current measurement resistor (R2) is supplied to a switching-off device, which evaluates the time mean value or the instantaneous value of this voltage and, if a given limit value is exceeded, prevent; further oscillation of the half-bridge inverter (HB).
4. The operating device as claimed in claim 1, characterized in that the operating device has two mains voltage terminals (J1, J2), which can be connected to a mains voltage, and power factor correction for a mains current flowing via the mains voltage terminals (J1, J2) is achieved by means of a pumping circuit.
5. The operating device as claimed in claim 4, characterized in that the pumping circuit has the following features:
~ a portion of the mains current flows via a first pumping diode (D5) which, with a second pumping diode (D6), forms a first diode series circuit having a first diode connection point, with the diodes being connected such that they allow current to flow from the mains terminals to the half-bridge inverter (HB), ~ the operating device has at least two lamp terminals (J3, J4), which can be connected to lamp connections, with one lamp terminal (J3) being connected to the first diode connection point via a pumping capacitor (C6).
~ a portion of the mains current flows via a first pumping diode (D5) which, with a second pumping diode (D6), forms a first diode series circuit having a first diode connection point, with the diodes being connected such that they allow current to flow from the mains terminals to the half-bridge inverter (HB), ~ the operating device has at least two lamp terminals (J3, J4), which can be connected to lamp connections, with one lamp terminal (J3) being connected to the first diode connection point via a pumping capacitor (C6).
6. The operating device as claimed in claim 5, characterized in that the pumping capacitor (C6) is connected to that lamp terminal (J3) which, with respect to a reference ground potential (M), is at a voltage which has the maximum value for the AC voltage component in comparison to the voltage at :he other lamp terminals (J4).
7. The operating device as claimed in claim 5, characterized by the following features:
~ a second diode series circuit formed by two diodes (D4, D7) is connected in parallel with the first diode series circuit, thus forming a second diode connection point, with the diodes (D4, D7) being connected such that they allow current to flow from the mains to the half-bridge inverter (HB), ~ the second diode connection point is connected at least via a pumping inductor (L4) to the connection point (4) of the half-bridge transistors (T1, T2).
~ a second diode series circuit formed by two diodes (D4, D7) is connected in parallel with the first diode series circuit, thus forming a second diode connection point, with the diodes (D4, D7) being connected such that they allow current to flow from the mains to the half-bridge inverter (HB), ~ the second diode connection point is connected at least via a pumping inductor (L4) to the connection point (4) of the half-bridge transistors (T1, T2).
8. The operating device as claimed in claim 2, characterized in teat the operating device contains a starting capacitor (C41), which is connected to the current measurement resistor (R2) via a trigger diode (D40) and a diode (D43) connected in series.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| DE10200049.2 | 2002-01-02 | ||
| DE10200049A DE10200049A1 (en) | 2002-01-02 | 2002-01-02 | Control gear for gas discharge lamps |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2415510A1 true CA2415510A1 (en) | 2003-07-02 |
Family
ID=7711453
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002415510A Abandoned CA2415510A1 (en) | 2002-01-02 | 2002-12-30 | Operating device for gas discharge lamps |
Country Status (6)
| Country | Link |
|---|---|
| US (1) | US6677716B2 (en) |
| EP (1) | EP1326484B1 (en) |
| CN (1) | CN100438715C (en) |
| AT (1) | ATE336156T1 (en) |
| CA (1) | CA2415510A1 (en) |
| DE (2) | DE10200049A1 (en) |
Families Citing this family (9)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| DE10241327A1 (en) * | 2002-09-04 | 2004-03-18 | Patent-Treuhand-Gesellschaft für elektrische Glühlampen mbH | Circuit arrangement for operating discharge lamps |
| DE10351621B4 (en) * | 2003-11-05 | 2013-05-16 | Osram Gmbh | Electronic ballast and method with weiterzuvertet in case of failure of the light-emitting device converter |
| DE102004001618A1 (en) * | 2004-01-09 | 2005-08-11 | Patent-Treuhand-Gesellschaft für elektrische Glühlampen mbH | Circuit arrangement for operating light sources |
| DE602005013083D1 (en) * | 2004-06-21 | 2009-04-16 | Koninkl Philips Electronics Nv | CONTROL METHOD FOR A GAS DISCHARGE LAMP |
| JP2009539218A (en) * | 2006-05-31 | 2009-11-12 | コーニンクレッカ フィリップス エレクトロニクス エヌ ヴィ | Method and system for operating a gas discharge lamp |
| EP2145380B1 (en) * | 2007-04-27 | 2016-01-06 | Koninklijke Philips N.V. | Self-oscillating switch circuit for use in a switching dc-dc-converter |
| JP4816634B2 (en) * | 2007-12-28 | 2011-11-16 | ウシオ電機株式会社 | Substrate heating apparatus and substrate heating method |
| US20090200953A1 (en) * | 2008-02-08 | 2009-08-13 | Ray James King | Methods and apparatus for a high power factor ballast having high efficiency during normal operation and during dimming |
| US8258712B1 (en) * | 2008-07-25 | 2012-09-04 | Universal Lighting Technologies, Inc. | Ballast circuit for reducing lamp striations |
Family Cites Families (11)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| NL8201631A (en) | 1982-04-20 | 1983-11-16 | Philips Nv | DC AC CONVERTER FOR IGNITION AND AC POWERING A GAS AND / OR VAPOR DISCHARGE LAMP. |
| US5173643A (en) * | 1990-06-25 | 1992-12-22 | Lutron Electronics Co., Inc. | Circuit for dimming compact fluorescent lamps |
| US5545955A (en) * | 1994-03-04 | 1996-08-13 | International Rectifier Corporation | MOS gate driver for ballast circuits |
| JP3484863B2 (en) * | 1995-03-29 | 2004-01-06 | 東芝ライテック株式会社 | Power supply device, discharge lamp lighting device and lighting device |
| EP0757512B1 (en) * | 1995-07-31 | 2001-11-14 | STMicroelectronics S.r.l. | Driving circuit, MOS transistor using the same and corresponding applications |
| DE19548506A1 (en) * | 1995-12-22 | 1997-06-26 | Patent Treuhand Ges Fuer Elektrische Gluehlampen Mbh | Circuit arrangement for operating a lamp |
| US5619106A (en) * | 1996-06-24 | 1997-04-08 | General Electric Company | Diodeless start circiut for gas discharge lamp having a voltage divider connected across the switching element of the inverter |
| EP0855132B1 (en) * | 1996-08-14 | 2002-06-19 | Koninklijke Philips Electronics N.V. | Circuit arrangement |
| US6078143A (en) * | 1998-11-16 | 2000-06-20 | General Electric Company | Gas discharge lamp ballast with output voltage clamping circuit |
| US6541925B1 (en) * | 1998-11-18 | 2003-04-01 | Koninklijke Philips Electronics N.V. | Resonant converter circuit with suppression of transients during changes in operating condition |
| ITMI991131A1 (en) * | 1999-05-21 | 2000-11-21 | St Microelectronics Srl | HALF-BRIDGE SEMI-BRIDGE PILOTING ARCHITECTURE AT VARIABLE FREQUENCY, IN PARTICULAR FOR ELECTRIC LOADS |
-
2002
- 2002-01-02 DE DE10200049A patent/DE10200049A1/en not_active Withdrawn
- 2002-12-04 AT AT02027137T patent/ATE336156T1/en active
- 2002-12-04 EP EP02027137A patent/EP1326484B1/en not_active Expired - Lifetime
- 2002-12-04 DE DE50207779T patent/DE50207779D1/en not_active Expired - Lifetime
- 2002-12-20 US US10/323,748 patent/US6677716B2/en not_active Expired - Fee Related
- 2002-12-30 CA CA002415510A patent/CA2415510A1/en not_active Abandoned
-
2003
- 2003-01-02 CN CNB031009034A patent/CN100438715C/en not_active Expired - Fee Related
Also Published As
| Publication number | Publication date |
|---|---|
| EP1326484A3 (en) | 2005-01-05 |
| DE50207779D1 (en) | 2006-09-21 |
| CN100438715C (en) | 2008-11-26 |
| US20030122504A1 (en) | 2003-07-03 |
| EP1326484A2 (en) | 2003-07-09 |
| ATE336156T1 (en) | 2006-09-15 |
| CN1430460A (en) | 2003-07-16 |
| DE10200049A1 (en) | 2003-07-17 |
| EP1326484B1 (en) | 2006-08-09 |
| US6677716B2 (en) | 2004-01-13 |
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