CN101743683B - 功率因数校正（pfc）控制器及使用有限状态机调节pwm控制信号工作周期的方法 - Google Patents

功率因数校正（pfc）控制器及使用有限状态机调节pwm控制信号工作周期的方法 Download PDF

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CN101743683B
CN101743683B CN2008800230275A CN200880023027A CN101743683B CN 101743683 B CN101743683 B CN 101743683B CN 2008800230275 A CN2008800230275 A CN 2008800230275A CN 200880023027 A CN200880023027 A CN 200880023027A CN 101743683 B CN101743683 B CN 101743683B
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约翰·L·梅安森
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Cirrus Logic Inc
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4225—Arrangements for improving power factor of AC input using a non-isolated boost converter
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M3/00—Conversion of analogue values to or from differential modulation
    • H03M3/30—Delta-sigma modulation
    • H03M3/458—Analogue/digital converters using delta-sigma modulation as an intermediate step
    • H03M3/476—Non-linear conversion systems
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/10—Efficient use of energy, e.g. using compressed air or pressurized fluid as energy carrier

Abstract

本发明公开一种功率因数校正(power factor correction，PFC)控制器以及一种使用有限状态机(304)调节脉宽调制(pulse width modulation，PWM)开关控制信号(CS0)工作周期的方法。所述功率因数校正(PFC)控制器具有目标电流发生器(300)，该目标电流发生器(300)接收链路输出电压(Vcic(t))，并产生与整流线路输入电压(Vxic(t))成比例的目标电流(V(itarget))。所述功率因数校正(PFC)控制器还包括比较器(303)，该比较器(303)输出两电平电流比较结果信号。对两电平电流比较结果信号作出响应的有限状态机(304)产生具有工作周期的开关控制信号(CS0)，可调节该工作周期，以控制所述开关，从而使感测电流与整流线路输入电压近似成比例，由此进行功率因数校正。

Description

功率因数校正(PFC)控制器及使用有限状态机调节PWM控制信号工作周期的方法
技术领域
本发明大体上涉及信号处理领域，特别涉及一种功率因数校正(powerfactor correction，PFC)控制器以及一种使用有限状态机调节脉宽调制(pulse width modulation，PWM)开关控制信号工作周期的方法。 
背景技术
功率因数校正器经常利用开关模式升压级，将交流电(alternatingcurrent，AC)电压(例如线路/电源电压)转换为直流电(direct current，DC)电压或DC-至-DC，其中输入电流与输入电压成比例。功率因数校正器向许多利用调整输出电压的设备提供功率因数校正(，PFC)和调整后的输出电压。 
图1为功率因数校正器100的一典型实施例，包括开关模式升压级102。电压源101向全波二极管桥式整流器103提供交流电(AC)输入电压Vin(t)。电压源101(例如电压Vin(t))例如可以是公用事业用电，诸如在美国为60Hz/120V线路(电源)电压，或在欧洲为50Hz/230V线路(电源)电压。与输入电压Vin(t)相关的输入速率是电压源101的频率(例如在美国为60Hz，在欧洲为50Hz)。整流器103对输入电压Vin(t)进行整流，并向开关模式升压级102提供整流时变线路输入电压Vx(t)。在任何时刻t时的实际电压被称为瞬时输入电压。除非另有说 明，术语“线路速率”在下文中是指且被定义为与整流线路电压Vx(t)相关的整流输入频率。线路速率也等于与输入电压Vin(t)相关的输入频率的两倍。可根据诸如Vrms的均方根(Root Mean Square，RMS)电压，测量和提供整流线路输入电压。 
开关模式升压级102包括对其控制的开关108(例如场效应晶体管(Field Effect Transistor，FET))，并根据开关108的受控方式提供功率因数校正(PFC)。开关模式升压级102也由开关108控制，且通过二极管111调整能量从整流线路输入电压Vx(t)经过感应器110向电容器106的传输。当开关108接通时，即处于“ON”状态，感应器电流iL斜升。当开关108未接通时，即处于“OFF”状态，感应器电流iL斜降，并向再充电电容器106提供电流iL。感应器电流iL发生斜降的时间段通常被称为“感应器回描时间”。可将感应电阻109与感应器110有效地串联起来使用。 
功率因数校正器100的功率因数校正(PFC)控制器114控制开关108，并由此控制功率因数校正，且调节开关模式升压级102的输出功率。功率因数校正技术的目的是使开关模式升压级102对电压源101显示出电阻性。因此，功率因数校正(PFC)控制器114试图控制感应器电流iL，以使平均感应器电流iL与整流线路输入电压Vx(t)成比例。在以下两文献中描述了功率因数校正(PFC)控制器114的实施例：1.Unitrode产品数据表“UCC2817，UCC2818，UCC3817，UCC3818BiCMOS Power Factor Preregulator”(SLUS3951)，由Texas InstrumentsIncorporated于2000年2月发表，并于2006年2月修订，版权 2006-2007(在下文中被称为“Unitrode数据表”)；以及2.国际整流器数据表“Datasheet No.PD60230rev CIR1150(S)(PbF)andIR11501(S)(PbF)”，由International Rectifier于2007年2月5日发表。功率因数校正(PFC)114提供脉宽调制(PWM)控制信号CS0，以控制开关108的导电率。 
存在以下两种开关级操作模式：不连续传导模式(“DiscontinuousConduction Mode，DCM”)和连续传导模式(“Continuous ConductionMode，CCM”)。在DCM中，当感应器电流iL等于零时，功率因数校正(PFC)控制器114(或升压转换器)的开关108被接通(即处于“ON”)。在CCM中，当感应器电流不等于零时，功率因数校正(PFC)控制器114(或升压转换器)的开关108被接通，即处于“ON”，且能量传输感应器110中的电流从未在开关周期内达到零。在CCM中，电流摆动比在DCM中少，因而使感应器电流iL的I2R功率损耗和波纹电流减小，从而降低感应器芯的损耗。较低的电压摆动也可减少电磁干扰(Electro Magnetic Interference，EM1)，由此可使用更小的输入滤波器。由于当感应器电流iL不等于零时开关108被断开，即处于“OFF”，二极管111的反向恢复需很快，以最大程度减小损耗。 
开关108的开关速率典型地在20kHz至100kHz范围内。应避免较慢的开关频率，以避免人类听觉的音频范围，并避免增加感应器110的尺寸。应典型地避免较快的开关频率，这是因为较快的开关频率会增加开关损耗，且在使用上更难以满足无线电频率干扰(Radio FrequencyInterference，RFI)标准的要求。 
电容器106向负载112提供蓄能。电容器106需充分大，以通过线路速率周期保持大体上稳定的链路输出电压Vc(t)。链路输出电压Vc(t)在稳定负载条件下大体上保持稳定。然而，当负载条件变化时，链路输出电压Vc(t)也会变化。功率因数校(PFC)控制器114对链路输出电压Vc(t)的变化作出响应，并调节控制信号CS0，以尽快恢复为大体上稳定的输出电压。功率因数校正(PFC)控制器114包括小电容器115，以阻止任何高频开关信号与线路(电源)输入电压Vin(t)耦合。 
功率因数校正(PFC)控制器114通过宽带宽电流回路116和较慢电压回路118接收两个反馈信号，即整流线路输入电Vx(t)和链路输出电压Vc(t)。从二极管整流器103与感应器110之间的节点120检测整流线路输入电压Vx(t)。从二极管111与负载112之间的节点122检测链路输出电压Vc(t)。电流回路116工作于频率fc，该频率足以使PFC控制器114对整流线路输入电压Vx(t)的变化作出响应，并使感应器电流iL跟踪整流线路输入电压Vx(t)，以提供功率因数校正。由电流回路116控制的感应器电流iL具有5kHz至10kHz的控制带宽。电压回路118工作于慢得多的频率控制带宽下，约5Hz至20Hz。通过工作于5Hz至20Hz，电压回路118起到低通滤波器的作用，对链路输出电压Vc(t)的谐波波纹分量进行滤波。 
仍需若干在PFC控制器114外的外部元件，用于各自的功率因数校正器100。例如，功率因数校正器100可典型地由两个比例积分(Proportional-Integral，PI)控制器组成，一个PI控制器与电流回路116相关，另一个PI控制器与电压回路118相关。消除与电流回路116相 关的PI控制器既是所期望的，又可简化功率因数校正器100的电路。此外，典型的功率因数校正器100并不主要由数字电路组成。因此，需要且期望获得主要由数字电路组成的功率因数校正器，并最大程度减少在PFC控制器外用于功率因数校正器的外部元件数量。 
发明内容
本发明提供一种功率因数校正(PFC)控制器以及一种使用有限状态机调节脉宽调制(PWM)开关控制信号工作周期的方法。 
在功率因数校正器和对应方法的一实施例中，功率因数校正器(PFC)包括开关模式升压级、目标电流发生器、比较器和有限状态机。开关模式升压级具有开关和与该开关耦合的感应器。开关模式升压级接收整流线路输入电压，并提供链路输出电压。可从开关模式升压级观测感测电流。目标电流发生器接收链路输出电压，并产生与整流线路输入电压成比例的目标电流。比较器接收代表目标电流的目标电流值和代表感测电流的感测电流值。比较器输出两电平电流比较结果信号。对两电平电流比较结果信号作出响应的有限状态机可产生具有工作周期的开关控制信号，该工作周期可被调节，以控制开关，从而使感测电流与整流线路输入电压近似成比例，由此进行功率因数校正。 
在功率因数校正(PFC)和对应方法的另一实施例中，功率因数校正器(PFC)包括开关模式升压级、目标电流发生器、比较器和有限状态机。开关模式升压级具有开关和与该开关耦合的感应器。开关模式升压级接收整流线路输入电压，并提供链路输出电压。可从开关模式升压级观测感测电流。目标电流发生器接收链路输出电压，并产生与整流线路输入电压成比例的目标电流。波纹电流测定仪产生可估计峰-峰感应器波纹电流I的波纹电流。对目标电流、波纹电流和感测电流作出响应的比较器输出两电平电流比较结果信号。对两电平电流比较结果信号作出响应的有限状态机可产生具有工作周期的开关控制信号，该工作周期可被调节，以控制开关，从而使感测电流与整流线路输入电压近似成比例，由此进行功率因数校正。
在每个实施例中，可将目标电流发生器、比较器和有限状态机结合起来，组成功率因数校正(PFC)控制器。可将PFC控制器安装在单一的集成电路上，并按比例相应确定整流线路输入电压和链路输出电压，以用于集成电路。 
附图说明
通过参考附图，本领域技术人员可更好地理解本发明，并更清楚地认识本发明的众多物体、特征和优点。在一些图中使用相同的参考编号，以指定一种相同或相似的元件。 
图1(标记为现有技术)显示一种功率因数校正器。 
图2A显示功率因数校正器的一优选实施例，该功率因数校正器具有开关模式升压级和有限状态机，该有限状态机利用感测电流和目标电流，以调节PWM控制信号的工作周期，该PWM控制信号控制开关模式升压级的开关。 
图2B显示功率因数校正器的另一优选实施例，该功率因数校正器具有开关模式升压级和有限状态机，该有限状态机利用感测电流和目标电流，以调节PWM控制信号的工作周期，该PWM控制信号控制开关模式升压级的开关。 
图3显示图2A和图2B中PFC控制器实施例的细节。 
图4显示图3中一有限状态机实施例，且图3显示时基控制器的一数字式实施例的细节。 
图5显示图4中时基控制器一数字式实施例工作的高级逻辑流程图。 
图6A显示图2A和图2B中功率因数校正器的整流线路输入电流(例如平均升压感应器电流iL)和目标电流(例如itarget)的电流波形实施例，且时间标度为10毫秒。 
图6B为图2A和图2B中功率因数校正器的整流线路输入电流(例如平均升压感应器电流iL)和目标电流(例如itarget)的电流波形实施例，且时间标度为10微秒。 
图6C显示被馈送进代表性电压Vx函数产生器内的参数实施例，该函数产生器用于产生断开时间，该断开时间为线路输入电压Vx(t)的函数。 
图7A显示图6C中代表性电压Vx函数产生器的传递函数，图6C显示相对于线路输入电压(Vx)值被映射的断开时间-最短断开时间比(toff/toffmin)。 
图7B显示当输入电压Vin(t)工作于120伏特(RMS)输入电压源时，线路输入电压Vx(t)的10毫秒周期。 
图7C显示图7B中的10毫秒周期，给出当将图7A中的传递函数作用于图7B中的线路输入电压Vx(t)的10毫秒周期上时，得到的断开时间-最短断开时间比。 
图8A显示当输入电压Vin(t)工作于240伏特(RMS)输入电压源时，线路输入电压Vx(t)的10毫秒周期。 
图8B显示图8A中的10毫秒周期，给出当将图7A中的传递函数作用于图8A 中的线路输入电压Vx(t)的10毫秒周期上时，得到的断开时间-最短断开时间比。 
图9A显示图2A中功率因数校正器一优选开关控制算法的电流波形实施例，其中开关电流在其接通期间有50％高于目标电流、50％低于目标电流。 
图9B显示图2B中功率因数校正器另一优选开关控制算法的电流波形实施例，其中升压感应器电流有50％高于目标电流、50％低于目标电流。 
图10显示通过图3和图4中的有限状态机实施以下两种优选开关控制算法得到的状态图：具有图9A中电流波形实施例特点的优选开关控制算法，或具有图9B中电流波形实施例特点的另一优选开关控制算法。 
图11显示图2A中功率因数校正器又一优选开关控制算法的电流波形实施例，其中当开关电流达到由目标电流和波纹电流确定的峰值电流时，可确定开关电流的接通时间。 
图12显示通过图3和图4中有限状态机实施以下算法得到的状态图：具有图11中电流波形实施例特点的优选开关控制算法。 
图13显示图2A中与存储器耦合的功率因数校正器。 
图14A至14J为图9A、图9B和图10所示开关控制算法函数实施例的Mathematica代码。 
图15A至15J为图11和图12所示开关控制技术算法函数实施例的Mathematica代码。 
具体实施方式
功率因数校正器包括开关模式升压级和功率因数校正(PFC)控制器，该功率因数校正控制器包括有限状态机，该有限状态机调节脉宽调制(PWM)开关控制信号的工作周期，用于根据本发明原理控制开关模式升压级的控制开关。在本发明的实施例中(例如如图2A和图2B所示)，有必要观测感应器(感应器210)中的电流(例如iL)。在下文中将详细讨论两个可能的电流观测实施例。“isense”在这里被用作对观测电流的通称。当开关208接通时，在图2A中的功率因数校正器200A实施例仅观测感测电流isense(例如开关电流isenseA)。图2B 中的功率因数校正器200B实施例观测所有时间(例如无论开关208接通或断开与否)的感测电流isense(例如升压感应器电流isenseB)。 
图2显示功率因数校正器200A，且功率因数校正器200A包括开关模式升压级202A和PFC控制器2114。PFC控制器214根据本发明的原理，使用有限状态机调节PWM开关控制信号CS0的工作周期。功率因数校正器202A包括全波二极管桥式整流器203、电容器215、开关模式升压级202A和PFC控制器214。开关模式升压级202A还包括感应器210、二极管211、开关208和电容器206。功率因数校正器200A实施高带宽电流回路216和较慢电压回路218。线路(电源)电压源201可与功率因数校正器200A的输入端耦合，且负载212可与功率因数校正器200A的输出端耦合。功率因数校正器200A以与功率因数100相似的方式工作，该方式除开关208的受控方式外如上文所述。 
在功率因数校正器200A的实施例中，对开关208的开关进行计算和运行，以使作为线路输入电流的升压感应器电流iL的平均电流随整流线路输入电压Vx(t)成比例变化，其中以某种方式选择比例，以调节电容器链路电压/输出电压Vc(t)。通过将感测电流电阻209A与开关208串联耦合(例如在开关208与线路输入电压Vx(t)的负节点之间耦合)，可进一步实现使功率因数校正器200A对电压源201呈现电阻性的目的。因此，感测电流isense在这种情况下为开关电流isenseA。利用开关电流isenseA提供开关模式升压级202A的PFC控制。稍后当更详细地讨论图9A时，将对该PFC控制进行讨论。PFC控制器214也具有有限状态机，用于调节PWM开关控制信号的工作周期。 
PFC控制器214及其工作和函数可在单一的集成电路上实施。由电阻R1和R2组成的分压器与PFC控制器214的输入端耦合，且线路输入电压Vx(t)被馈送到PFC控制器214的输入端内，另一由电阻R3和R4组成的分压器与PFC控制器214的输入端耦合，且线路输出电压Vc(t)被馈送到PFC控制器214的输入端内。选择电阻R1、R2、R3和R4的电阻值，以使分压器按比例降低线路输入电压Vx(t)和链路输出电压Vc(t)，使二者分别成为可用于集成电路的比例线路输入电压VxIC(t)和比例链路输出电压VcIC(t)。在实施功率因数校正器200A时，只有当开关208接通时才能观测感应器电流iL。 
图2B显示功率因数校正器200B的另一优选实施例。功率因数校正器200B除未使用开关电流电阻209A外，与功率因数校正器200A相同；其中开关电流isenseA用于提供开关模式升压级202A的PFC控制，且可将升压感应器电流感应电阻209B与感应器210串联起来有效利用。这一可选实施例中的升压感应器电流isenseB用于提供开关模式升压级202B的PFC控制。稍后当更详细地讨论图9B时，将对该PFC控制进行讨论。PFC控制器214仍根据本发明的原理，使用有限状态机(有限状态机304)调节PWM开关控制信号CS0的工作周期。功率因数校正器200B中的PFC控制器214及其工作和函数也可在单一的集成电路上实施。可使用功率因数校正器200A的由相同电阻R1、R2、R3和R4组成的分压器，按比例降低功率因数校正器200B中的线路输入电压Vx(t)和链路输出电压Vc(t)，使二者分别成为可用于集成电路的比例线路输入电压Vx1C(t)和比例链路输出电压Vc1C(t)。在实施功率因数校正器 200B时，可在所有时间观测感应器电流iL(例如无论开关208接通或断开与否)。 
感应电阻209A或209B的典型值为0.05至0.5欧姆。感测电流的方式当然不局限于使用图2A和图2B中公开的感应电阻或功率因数校正器200A和200B。其他电流检测技术可可选地包括使用电流感应变压器或霍尔(Hall)效应设备。 
图3显示PFC控制器214一实施例的细节，该控制器214实施例用于图2A和2B中分别显示的功率因数校正器200A和200B中。PFC控制器214实施例包括目标电流发生器300、比较器303和有限状态机304，它们以图3中所示方式耦合在一起。目标电流发生器300包括模-数转换器(ADC)320、乘法器301、数-模转换器(DAC)307、乘法器312、数字式比例积分(PI)电压控制器302和ADC 322。数字式PI控制器302可以是具有比例线路路径和积分线路路径的典型PI控制器。较慢电压回路218的集成电路(IC)比例链路输出电压VcIC(t)被馈送到ADC 322内，且ADC 322的相应数字输出信号323被馈送到数字PI电压控制器302内。PI控制器302优选采用无限脉冲响应(Infinite Impulse Response，IIR)数字滤波器形式。IIR滤波器的输出(例如来自PI控制器302的PI输出信号315)与负载212的估计功率需求成比例，且该负载212将被传递到开关模式升压级202A或202B。PI输出信号315被馈送到乘法器312内。等价于1/Vrms 2的换算因数信号317也被馈送到乘法器312内。乘法器312与来自换算因数信号317和PI输出信号315的值相乘，产生比例数字输出电压信号309。 
ADC 320接收IC比例线路输入电压VxIC(t)，并提供相应的数字输出信号319，该数字输出信号319为IC比例整流线路输入电压VxIC(t)的数字表示。数字输出信号319被馈送到乘法器301内。乘法器301将比例数字输出电压信号309的值与数字输出信号319的值相乘，得到数字输出乘积值350。数字输出乘积值350被馈送到DAC 307内，由DAC 307将数字输出乘积值350转换成代表目标电流值(itarget)的参考电压V(itarget)。感测电流电压V(isense)(例如感测电流值)为代表感测电流信号isense的电压，该感测电流信号isense可以是开关模式升压级202A内的开关电流isenseA，或开关模式升压级202B内的升压感应器电流isenseB。比较器303将参考电压V(itarget)与感测电流电压V(isense)作比较。参考电压V(itarget)与目标电流itarget对应，且与之成比例，该目标电流itarget为线路输入电流iL要求的平均感应器电流。与输入电压Vx(t)成比例的目标电流itarget可确保输入电流iL具有高功率因数和低失真。目标电流itarget具有如图6A和图6B所示的全波整流正弦电流波形。 
比较器303输出电压比较结果信号314，该电压比较结果信号314在这一实施例中为一种两电平电流比较结果信号(例如“两电平”意味着诸如高于要求阈值的一比较状态，以及诸如低于要求阈值的另一比较状态)，且被馈送到有限状态机(finite state machine，FSM)304内。根据本发明的原理，有限状态机304处理FSM算法(例如开关控制算法)，该FSM算法基于接收到的电压比较结果信号314，调节PWM开关控制信号CS0的工作周期，以控制开关208。本发明中FSM算法的一实施例是控制或调节PWM开关控制信号CS0的开关定时，以使升压感应 器电流iL在百分之五十(50％)的时间内高于目标电流阈值，在其余百分之五十(50％)的时间内低于目标电流阈值。FSM算法的这一实施例将在稍后说明图9A、图9B和图10时更详细地讨论。本发明中FSM算法的另一实施例是控制或调节PWM开关控制信号CS0的开关定时，以使控制开关208被启动，直到升压感应器电流iL达到目标电流itarget，该itarget等于由itarget+iRIPPLE/2确定的峰值电流值ipeak，其中itarget为目标电流，iRIPPLE为感应器210中的峰-峰波纹电流。FSM算法的这一其它实施例以及这些电流值和等式将在稍后说明图11和图12时更详细地讨论。 
图3中显示的PFC控制器214实施例是举例性的，还会存在许多可能的变化。乘法器可以是模拟式的而不是数字式的，从而消除相关的模-数(A/D)和数-模(D/A)转换步骤。模拟输入信号可以是电流而不是电压。输入正弦波可以由函数产生器的表格查寻功能合成，并同步化到外部线路中。比较器可以是数字式的，且电流输入可以是数字化的。在所有情形下，可产生电流参考信号，且可将其与开关模式电流作比较。该比较被有限状态机(FSM)用于控制开关控制工作周期。 
现参见图4，说明有限状态机(FSM)304的一实施例。FSM 304包括时基控制器410。时基控制器实施例在约翰·梅安森(John Melanson)于2007年9月28日递交并转让给共同受让人思睿逻辑有限公司，奥斯丁，特克萨斯州(Cirrus Logic，Inc.，Austin，Texas)的名为“具有综合反应的系统的时基控制”(“Time-Based Control of a System HavingIntegration Response”)(下文称为“Time-Based Control patent case”，即 “时基控制专利案”)的未决美国专利申请编号11/864,366中公开。时基控制专利权案在此被引用为本说明书的参考资料。 
如其名称所示，时基控制器410实施一种时基控制方法，而不是上述传统的基于大小的控制方法之一。时间计算逻辑422接收电压比较结果信号314，并对此作出响应，确定何时应改变开关208的状态，以将感测电流信号isense的平均值保持在目标平均值(例如目标电流itarget)。时基控制器410包括脉宽调制器(PWM)424，该脉宽调制器424对控制信号430断言或解除断言，以在时间计算逻辑422所指示的时间改变开关208的状态。此外，振荡器432被馈送到FSM 304和时间计算逻辑块422内。可利用振荡器432实现期望的FSM算法。 
现参见图5，说明图4中时基控制器410的数字式实施例工作的高级逻辑流程图。正如本技术领域所已知，被说明的过程可由专用集成电路(application specific integrated circuit，ASIC)、通用数字硬件执行程序代码或其它数字电路实施，且该通用数字硬件执行程序代码来自可引导被说明操作的有形数据存储介质。进而，可利用被说明的过程实施任何时基恒定周期、时基恒定断开时间、时基可变周期或时基可变断开时间控制方法。 
图5中所示过程从框501开始，然后继续到框502，框502描述时基控制器410，时基控制器410断言控制信号CS0(例如将控制信号CS0置于第一个状态)，以接通开关208，并由此开始时间间隔A(例如图7A和图9中的时间tonA)。然后，过程在框504重复，直到在正转换中比较器418通过发信号，表明在这种情况下感测电流信号isense值已与目标电流itarget值交叉，从而指示时间间隔已结束。由此，电压比 较结果信号314与感测电流信号isense在相对大小方面的变化引起比较器303输出的极性变化，表明(在本实施例中)感测电流信号isense至少与目标电流itarget一样大(或在其它实施例中，感测电流信号isense等于或小于目标电流itarget)。在正转换中，比较器303指示感测电流信号isense已与目标电流itarget交叉；作为对此作出的响应，时间计算逻辑422基于数字计数器或计时器的值，记录时间间隔A(框506)的持续时间。然后，时间计算逻辑422例如利用等式之一(例如用于计算接通时间tonB的等式B)，计算时间间隔B的持续时间(例如图7A和图9中的时间tonB)。稍后将根据图10(框508)讨论所述等式。 
然后，如框510和框512所示，脉宽调制器424检测(例如利用数字计数器或计时器)时间间隔B的计算持续时间何时已从时间比较器303经过，且时间比较器303可指示时间间隔B的结束。如果确定时间间隔B的计算持续时间已经过，作为对此作出的响应，脉宽调制器424对控制信号CS0解除断言(例如将控制信号CS0置于第二个状态)，以断开开关208。脉宽调制器424随后根据选择控制方法(框514)，等待固定或可变的断开时间(图7A和图9中的时间间隔toff)，并再次断言控制信号CS0，以接通开关208，且开始下一工作周期的时间间隔A(例如时间tonA)，如框502所示。随后，过程如已述继续进行。(54)现参见图6A，该图以10毫秒为时间标度，由曲线600(单位为安培)显示图2A和图2B中功率因数校正器的线路输入电流(例如平均升压感应器电流iL)和目标电流itarget的电流波形实施例。从100Hz整流线路(电源)速率实施例方面，查看波形实施例。从图6A中的这一 线路(电源)速率看，目标电流itarget被视为全波整流正弦波形的一半，且发生变化或移动。即使代表感应器电流iL电流波形的粗黑线显示为实线，它们实际上不是实线，且表示感应器电流iL快速移动，因而相对于目标电流itarget的电流波形显示为实线。现参见图6B，该图以10微秒为时间标度，由曲线602(单位为安培)显示图2A和图2B中功率因数校正器的线路输入电流(例如平均升压感应器电流iL)和目标电流(例如itarget)的电流波形实施例。从100kHz开关速率(例如开关208的开关速率)实施例方面，查看波形实施例。从图6B 中所示的这方面，目标电流itarget被视为伪稳定。 
因此，在本说明书中，术语“开关速率固定”被定义和理解为信号在线路(电源)速率标度上变化，但在开关速率标度上(从开关速率看)被视为稳定或恒定，特别是为了实施本发明所述的FSM算法(例如开关控制算法)。由于开关速率远高于线路输入速率(高出100至1000倍数量级)，将信号视为稳定是可能的。如果例如断开时间缓慢变化，则术语“开关速率固定”及其定义和含意仍适用。也可将缓慢改变断开时间toff的功率因数校正器200A和200B描述为“由快速电流反馈(例如电流回路216)控制的接通时间，由慢电压前馈(例如Vx)控制的断开时间”。慢电压前馈不对快速电流反馈的工作作出实质性的干扰。 
在本说明书中，当讨论工作模式或平均值时，除非另有说明，认为以开关速率作为时间标度。因此，当开关模式升压级202A或202B给出与线路(电源)电压成比例的电流时，意味着该电流在几个开关速率周期上取平均。此外，“恒定断开时间”模式将意味着断开时间在开关速率下 的几个开关周期内相对相同，但在线路(电源)速率上可变化。在线路速率处的变化对效率和无线电频率干扰(Radio Frequency Interference，RFI)可以是有利的。 
功率因数校正器200A和200B的实施例以及任何其它适当的功率因数校正器实施例能够通常以至少三种不同的方法，改变PWM控制信号CS0的工作周期。作为优选实施例的第一种方法是在感测电流isense的周期循环上，改变开关208的接通时间ton(例如图7A、图7B和图9中所示的tonA和tonB)，并将开关208的断开时间toff(例如图7A、7B和图9所示的toff)保持为常数。改变工作周期的第一种方法的优点是实现这类算法所需的数学计算相对简单，噪音被分布在宽谱上，且对PFC控制器214的控制是稳定的。例如，计算优选实施例之一中开关208的接通时间tonB仅涉及递归等式TonB’＝(TonB +TonA)/2。将在稍后描述图10时对该等式进行更详细的讨论。第二种方法是在感测电流isense的周期循环上，将开关208的接通时间ton(例如图7A、图7B和图9中所示的tonA和tonB)保持为常数，并改变开关208的断开时间toff(例如图7A、图7B和图9中所示的toff)。第三种方法是使感测电流isense的周期保持恒定，并相应改变开关208的接通时间ton和断开时间toff。第三种方法需要进行更多的数学计算，但在需要完全控制开关208的控制信号CS0的开关频率速率的某些情形下是优选的。在感测电流isense的周期被保持恒定的第三种方法中，在比较器303的输出350中可能会存在噪音，所以可增加更多的滤波处理。一实施例将是进行滤波，该滤波 具有下列关于接通时间ton的数学关系：tonB′＝3/4tonB+1/4tonA′。其中tonA′用于标识从感测电流isense的这一周期得到的新值。 
从线路(电源)速率方面改变断开时间toff可使开关模式升压级202A和202B具有更高功率因数和更高效率。期望断开时间toff在输入正弦周期的半周期上变化。当电压输入电源很高时(例如240V)，感应器回描时间(断开时间toff)可以相当长。根据图7所示，断开时间toff增加。改变断开时间toff的目的是在使效率最高的同时，将失真减到最小。从线路速率标度方面(例如与100Hz线路速率相关的10毫秒)看，会认为断开时间toff是“恒定的”(例如不快速改变)。然而，可在线路速率标度下改变断开时间toff，以改善性能。 
持续时间产生器可产生用于FSM 304的时间间隔(例如开关208的断开时间toff或开关周期)。在图6C中显示作为周期函数产生器652的一持续时间产生器实施例。周期函数产生器652接收IC比例整流线路输入电压VxIC(t)。参数实施例被馈送到周期函数产生器652内，以产生时间间隔，该时间间隔为IC比例整流线路输入电压VxIC(t)的函数。参数实施例用于确定周期函数产生器652的特性。被馈送到周期函数产生器652内的参数实施例可至少包括感应器210的感应器值654、时间间隔参数656、功率电平658和RMS电压660，其中功率电平658是与从电容器206测得的Vc相关的功率电平，RMS电压660是由RMS电压检测器测得的在电容器215两极间的RMS电压电平Vx。 
图7A显示图6C 中代表性电压Vx函数产生器的传递函数，图6C显示相对于线路输入电压(Vx)值被映射的断开时间-最短断开时间比 (toff/toffmin)。图7A通常显示断开时间toff如何作为瞬时线路电压Vx的函数(例如分段线性函数实施例)而变化。可选择该函数使效率最高，且该函数可可选地为数学函数或查看数据表函数。对于给定输入电压源(例如120伏特或240伏特)，该关系用于确定线路电压Vx如何在给定时间周期上变化，以及断开时间与最小断开时间比toff/toffmin如何在相同的时间周期上变化。在图14C的33至38行中显示断开时间的数学代码实施例作为瞬时线路电压Vx的分段线性函数发生变化。 
图7B显示当输入电压Vin(t)工作于120伏特(RMS)输入电压源时，线路输入电压Vx(t)的10毫秒周期。图7C显示与图7B中相同的10毫秒周期，且图7B显示当将图7A中的传递函数作用于图7B中的线路输入电压Vx(t)的10毫秒周期上时，得到的断开时间-最短断开时间比。在图7B和图7C所示的实施例中，链路输出电压Vc(t)的值是400伏特(例如在图2A和图2B中链路电容器206两极间的电压)。线路电压Vx的最高峰值等于120伏特(RMS)乘以√2(2的平方根)，约为170伏特。在这种情况下，断开时间与最小断开时间比toff/toffmin在1至1.3之间变化。因此，如果最小断开时间toffmin被设定成8微秒，则在输入正弦波峰处(例如在图7B中，在5毫秒时为170伏特)的断开时间toff将为10.4微秒。 
如图7B所示，曲线图702说明线路电压在10ms周期的最初5ms时，从0V增至170V，然后在10ms周期的第二个5ms时，从170V降至0V。在相同的10ms周期上，图7C显示曲线图704，曲线图704描述比值toff/toffmin在最初的几毫秒时保持在1，在5ms时升 至1.3，在之后的几毫秒时降至1，并在经过10ms时保持在1。从线路输入速率标度上查看图7B和图7C，所以在该时间标度，断开时间toff看起来在10ms检测周期实施例上变化。然而，从开关速率标度上(例如与100kHz相关的10微秒)看，将认为断开时间toff不快速变化。 
图8A显示当输入电压Vin(t)工作于240伏特(RMS)输入电压源时，线路输入电压Vx(t)的10毫秒周期。图8B显示与图8A中相同的10毫秒周期，且图8A显示当将图7A中的传递函数作用于图8A中的整流线路输入电压Vx(t)的10毫秒周期上时，得到的断开时间与最短断开时间比(toff/toffmin)。在该实施例中，链路输出电压Vc(t)的值是400伏特(例如在图2A和图2B 中链路电容器206两极间的电压)。线路电压Vx的最高峰值等于240伏特(RMS)乘以√2(2的平方根)，约为330伏特。此外，在该实施例中，最小断开时间toffmin仍为8微秒，所以最大断开时间toff为24微秒。因此，断开时间与最小断开时间比toff/toffmin可在1至3之间变化。 
在线路输入速率标度查看图8A和图8B，所以在该时间标度，断开时间toff看起来在10ms周期实施例上变化。然而，从开关速率标度方面(例如与100kHz线路速率相关的10微秒)看，将认为断开时间toff不快速变化。 
现参见图9A，对于控制开关208的一控制技术优选实施例，在开关速率标度显示开关电流isenseA和目标电流target的波形实施例。目标电流itarget被认为和视为相对于开关电流isenseA呈开关速率固定。开关电流isenseA在开关电流的接通时间期间显示出有百分之五十(50％)高于(例如 上方时间期间)和有百分之五十(50％)低于(例如下方时间期间)目标电流itarget。现参见图9B，在开关速率标度显示升压感应器电流isenseB和目标电流itarget的波形实施例。目标电流itarget被认为和视为相对于升压感应器电流isenseB呈开关速率固定。升压感应器电流isenseB在开关电流的接通时间显示出百分之五十(50％)高于和百分之五十(50％)低于在开关周期期间的目标电流itarget(例如tonA+tonB+toff)。在图9A和图9B中的波形有助于说明采用FSM算法对FSM 304实施的控制技术实施例。 
如上文所示，在开关模式升压级202中的开关电流isenseA和在开关功率转换器202B中的升压感应器电流isenseB通常可被称为感测电流信号isense。时基控制器410用于调节开关定时，以使感测电流isense(例如开关电流isenseA或升压感应器电流isenseB)高于目标电流(itarget)阈值达开关208接通时间ton的百分之五十(50％)，并低于目标电流(itarget)阈值达接通时间ion的其余百分之五十(50％)，且FSM算法(例如开关控制算法)将目标电流itarget视为相对于感测电流isense呈开关速率固定。 
升压感应器电流iL包括近似线性斜坡，且这些线性斜坡可确保升压感应器电流iL的平均电流(例如线路输入电流)跟踪目标电流itarget。在该控制技术实施例中，改变工作周期的第一种方法是当从开关速率标度看时，开关208的接通时间ton(例如在图7A、图7B和图9中所示的tonA和tonB)被改变，且认为断开时间toff在感测电流isense的周期循环上保持为常数。因此，该控制技术只需当开关208被启动或处于“ON”时检测感测电流isense，从而使其与图2A中的感应电阻209A和图2B 中的感应电阻209B相容。由于时基控制器410能够如上所述，实现对开关208的开关定时控制，因而不要求或不需要与电流回路216相关的PI控制器，由此可降低功率因数校正器200A和200B的电路复杂性。 
现参见图10，在图3和图4中显示FSM 304的状态图1000。状态图1000显示如何对功率因数校正器202A和202B的实施例实施FSM算法(例如开关控制算法)。现共同讨论图9A、图9B和图10。计时器与振荡器432和数字计数器一同实施，其中来自振荡器432的时钟驱动计数器的时钟输入。具有预先设定和向上/向下计数的数字计数器在本技术领域中广为人知。 
状态图1000显示对于该优选控制技术实施例，FSM算法移至状态S0，其中数字计数器被重新设定。在状态S0，开关208接通，且计数值TonA被起动。计数值TonA用于确定和跟踪开关208的接通时间tonA的经过时间。由比较器303比较感测电流isense与目标电流itarget，该比较器303比较感测电流电压V(isense)与参考电压V(iL)，该参考电压V(iL)对应于目标电流itarget。当感测电流isense继续小于目标电流itarget时，新计数值TonA’通过使当前值TonA增加一(1)，用于计算TonA的更新计数值，即 
TonA′＝TonA+1    等式A 
然后，当比较器303改变状态时，即当感测电流isense变得大于或等于目标电流itarget时，计数值TonA在锁存器中被俘获。计数值TonB用于确定和跟踪开关208的接通时间tonB的经过时间。通过使旧计数值 TonB加上被捕获的计数值TonA，并将所得总数除以二(2)，得出新计数值TonB’，以提供更新的接通时间tonB，即 
TonB′＝(TonB+TonA)/2    等式B 
然后重新设定数字计数器。 
FSM算法移至状态S1，在该状态S1，开关208仍处于接通状态，计时器则跟踪确定是否已达到当前计数值TonB，该当前计数值TonB反映了已经过的更新接通时间tonB。当已达到计数值TonB时，FSM算法随后移至状态S2，在该状态S2开关208被断开。然后再次重新设定数字计数器。计数值Toff用于确定和跟踪开关208的断开时间toff的经过时间，且计数值Toff被确定为线路输入电压Vx(t)的函数。当数字计数器达到计数值Toff时，FSM算法向后循环至状态S0，并随后重复。与根据状态图1000实施FSM算法并行，电压回路218在线路速率(例如100Hz或120Hz)被更新。 
现参见图11，对于控制开关208的另一优选控制技术实施例，图11按开关速率标度显示作为图2A中开关电流的开关电流isenseA和峰值电流iPEAK的波形实施例。如在前实施例中所讨论的，按与线路输入电压Vx(t)成比例计算目标电流itarget，且峰值电流iPEAK被认为和视为相对于开关电流isenseA呈开关速率固定。通过接通开关208，开关电流isenseA显示出斜升，直至开关电流isenseA达到峰值电流iPEAK。当开关电流isenseA达到峰值电流iPEAK时，开关208随后被断开达断开时间期间toff。 
波纹电流iRIPPLE被计算为在一个周期上感应器电流iL的最小值与最大值之间的差值。峰-峰感应器波纹电流iRIPPLE被计算为断开时间toff乘以比例链路输出电压Vc(t)与线路输入电压Vx(t)之间的差值，再除以感应器210的感应系数值L，即 
iRIPPLE＝toff*(Vc(t)-Vx(t))/L    等式C 
感应器中的预期峰值电流为： 
iPEAK＝itarget+iRIPPLE/2    等式D 
因此，如果感应器值L已知且具有合理的精确度，则可控制峰值电流iPEAK。所以，感应器值L可由运行时间测量确定。例如，如果从目标电流itarget至iPEAK测量开关电流isenseA的时间，则感应系数值可被确定为VX(t)*(delta时间/delta电流)。 
现参见图12，状态图1200显示对于这一优选控制技术实施例，FSM算法移至状态S0，在该状态S0开关208处于接通状态。然后，状态图1200显示FSM算法等待，直到在某条件下，峰值电流iPEAK大于目标电流itarget与波纹电流iRIPPLE的总和再除以二(2)，即 
iPEAK＞itarget+iRIPPLE/2    等式E 
当满足上述条件时，则在状态图1200中显示的FSM算法移至状态S1，在该状态开关208被接通。FSM算法然后等待“恒定的”断开时间(如之前所讨论)，然后返回状态S0，并随后重复。 
在图7A、7B、7C、8A、8B、9A、9B、10、11和12中的波形实施例确定了用于控制开关208的控制技术算法的函数和工作。这些函数和工作的特点确定了功率因数校正器200A和200B的工作特性和工作频率。同样地，期望对于不同的应用，允许对工作参数进行配置。在优选实施例中，参数被储存在可从外部存储器载入的数字寄存器中。例如只读存储器(read-only memory，ROM)、电可擦除可编程只读存储器(Electrically Erasable Programmable Read-Only Memory，EEPROM)、闪存、一次性可编程(one-time programmable，OTP)存储器可用于储存被选择值，且这些值在功率因数校正器200A或200B加电时被载入到数字寄存器中。可设定至少一些寄存器的缺省值。对于可编程功率因数校正器，存储器的编程允许对不同应用进行配置。 
在约翰·梅安森(John Melanson)于2007年12月31日递交并转让给共同受让人思睿逻辑有限公司(Cirrus Logic，Inc.，Austin，Texas)的名为可编程功率控制系统(“Programmable Power Control System”)(下文称为“Programmable Power Control Systems patent case”’，即“可编程功率控制系统专利案”)的未决美国专利中请编号11/967,275中讨论和公开了可编程功率因数校正器实施例。可编程功率控制系统专利案在此被引用为参考资料。 
图13显示功率因数校正器200A的一实施例，该功率因数校正器200A与存储器1300合并，该存储器1300与PFC控制器214耦合。存储器1300储存工作参数，例如参数实施例(例如感应器值654)以及用于确定电压Vx函数产生器652的传递函数的时间间隔(例如断开时间toff或感测电流周期参数656、功率电平658、RMS电平660)。在至少一个实施例中，存储器1300是非挥发性存储介质。存储器1300例如可以是读写存储器、一次性可编程存储器、用于表格查询的存储器或可加载寄存器。如何和何时将PFC工作参数载入存储器1300内、以及从存储器1300中存取是一种设计选择。 
图14A至14J显示用于如图9A、图9B和图10显示的一控制技术算法函数实施例的Mathematica代码(即Mathematica是一种由沃尔夫勒姆研究公司“Wolfram Research in Champaign，Illinois”开发的工程电脑程序)。图15A至15J显示用于如图11和图12显示的一控制技术算法函数实施例的Mathematica代码。 
虽然已对本发明进行了详细说明，但应理解的是，可在不偏离本专利申请的权利要求书所确定的本发明的精神和保护范围条件下，对本发明作出各种改变、替换和变更。 

Claims (35)

1.一种功率因数校正器(power factor corrector，PFC)，其特征在于包括： 
具有开关和与该开关耦合的感应器的开关模式升压级，所述开关模式升压级接收整流线路输入电压，并提供链路输出电压，且从所述开关模式升压级观测感测电流； 
目标电流发生器，用于接收所述链路输出电压，并产生与所述整流线路输入电压成比例的目标电流； 
比较器，用于接收代表所述目标电流的目标电流值和代表所述感测电流的感测电流值，并输出两电平电流比较结果信号；以及 
对所述两电平电流比较结果信号作出响应的有限状态机，用于产生具有工作周期的开关控制信号，调节该工作周期，以控制所述开关，从而使所述感测电流与所述整流线路输入电压成比例，由此进行功率因数校正。 
2.根据权利要求1所述的功率因数校正器，其特征在于所述感测电流为所述开关的开关电流。 
3.根据权利要求2所述的功率因数校正器，其特征在于所述有限状态机产生所述开关控制信号，以使所述开关电流在所述开关的百分之五十的接通时间期间大于所述目标电流，并在所述开关的其余百分之五十的接通时间期间小于所述目标电流。 
4.根据权利要求1所述的功率因数校正器，其特征在于所述感测电流为所述感应器内的升压感应器电流。 
5.根据权利要求4所述的功率因数校正器，其特征在于所述有限状态机产生所述开关控制信号，以使所述升压感应器电流在所述开关的百分之五十的接通时间期间大于所述目标电流，并在所述开关的其余百分之五十的接通时间期间小于所述目标电流。 
6.根据权利要求1所述的功率因数校正器，其特征在于还包括计时器，该计时器由所述有限状态机利用，用于测量所述感测电流小于所述目标电流时的下方时间期间，并用于根据所述下方时间期间，调节所述开关控制信号的所述工作周期。 
7.根据权利要求6所述的功率因数校正器，其特征在于还包括时间期间产生器，用于根据所述下方时间期间，接收代表所述整流线路输入电压的信号，并调节所述开关的断开时间期间。 
8.根据权利要求7所述的功率因数校正器，其特征在于还包括用于检测所述线路输入电压的RMS电压的检测器，且所述时间期间产生器接收和使用所述RMS电压，以响应地确定所述断开时间期间。 
9.根据权利要求7所述的功率因数校正器，其特征在于从所述链路输出电压检测功率电平，且所述时间期间产生器接收和使用所述功率电平，以响应地确定所述断开时间期间。 
10.根据权利要求6所述的功率因数校正器，其特征在于还包括时间期间产生器，用于根据所述下方时间期间，接收代表所述整流线路输入电压的信号，并调节所述开关的开关周期。 
11.根据权利要求10所述的功率因数校正器，其特征在于还包括用于检测所述线路输入电压的RMS电压的检测器，且所述时间期间产生器接收和使用所述RMS电压，以响应地确定所述开关周期。 
12.根据权利要求10所述的功率因数校正器，其特征在于从所述链路输出电压检测功率电平，且所述时间期间产生器接收和使用所述功率电平，以响应地确定所述开关周期。 
13.根据权利要求6所述的功率因数校正器，其特征在于所述有限状态机基于对开关的前接通时间期间和所述下方时间期间作出响应的递推关系，计算接通时间期间。 
14.根据权利要求6所述的功率因数校正器，其特征在于所述计时器还包括： 
振荡器；以及 
对所述振荡器和所述两电平电流比较结果信号响应地计数的数字计数器； 
其中所述开关控制信号控制对所述数字计数器作出响应的所述开关。 
15.根据权利要求1所述的功率因数校正器，其特征在于所述目标电流发生器、所述比较器和所述有限状态机被整合在单一的集成电路中。 
16.根据权利要求15所述的功率因数校正器，其特征在于所述集成电路还包括计时器，该计时器包括振荡器和计数器。 
17.一种功率因数校正(power factor correction，PFC)方法，其特征在于包括： 
将开关与感应器耦合在一起，形成开关模式升压级，该开关模式升压级接收整流线路输入电压，并提供链路输出电压； 
从所述开关模式升压级观测感测电流； 
接收链路输出电压； 
产生与所述整流线路输入电压成比例的目标电流； 
比较代表所述目标电流的目标电流值和代表所述感测电流的感测电流值； 
作为对所述比较作出的响应，输出两电平电流比较结果信号；以及 
作为对所述两电平电流比较结果信号作出的响应，产生具有工作周期的开关控制信号，调节该工作周期，以控制所述开关，从而使所述感测电流与所述整流线路输入电压成比例，由此进行所述功率因数校正。 
18.根据权利要求17所述的方法，其特征在于从所述开关模式升压级观测感测电流还包括： 
感测所述开关的开关电流。 
19.根据权利要求18所述的方法，其特征在于产生开关控制信号还包括： 
产生所述开关控制信号，以使所述开关电流在所述开关的百分之五十的接通时间期间大于所述目标电流，并在所述开关的其余百分之五十的接通时间期间小于所述目标电流。 
20.根据权利要求17所述的方法，其特征在于从所述开关模式升压级观测感测电流还包括： 
感测在所述感应器内的升压感应器电流。 
21.根据权利要求20所述的方法，其特征在于产生开关控制信号还包括： 
产生所述开关控制信号，以使所述升压感应器电流在所述开关的百分之五十的接通时间期间大于所述目标电流，并在所述开关的其余百分之五十的接通时间期间小于所述目标电流。 
22.根据权利要求17所述的方法，其特征在于还包括： 
测量所述感测电流小于所述目标电流时的下方时间期间；以及 
根据所述下方时间期间，调节所述开关的开关周期的接通时间期间。 
23.根据权利要求22所述的方法，其特征在于还包括： 
根据所述下方时间期间，调节所述开关的断开时间期间。 
24.根据权利要求22所述的方法，其特征在于还包括： 
根据所述下方时间期间，调节所述开关的开关周期。 
25.根据权利要求22所述的方法，其特征在于还包括： 
基于对前接通时间期间和所述下方时间期间作出响应的递推关系，计算所述开关的接通时间期间。 
26.一种合并有功率因数校正控制器的集成电路，其特征在于该功率因数校正控制器包括目标电流发生器、比较器和有限状态机，所述集成电路被配置如下： 
开关模式升压级接收整流线路输入电压，并提供链路输出电压； 
接收代表整流线路输入电压的信号； 
从开关模式升压级观测感测电流； 
接收代表链路输出电压的其他信号； 
产生与所述整流线路输入电压成比例的目标电流； 
比较代表所述目标电流的目标电流值和代表所述感测电流的感测电流值； 
开关模式升压级具有开关和与该开关耦合的感应器， 
作为对所述比较作出的响应，输出两电平电流比较结果信号；以及 
作为对所述两电平电流比较结果信号作出的响应，产生具有工作周期的开关控制信号，调节该工作周期，以控制所述开关，从而使所述感测电流与所述整流线路输入电压成比例，由此进行功率因数校正。 
27.一种功率因数校正器(power factor corrector，PFC)，其特征在于包括： 
具有开关和与该开关耦合的感应器的开关模式升压级，该开关模式升压级接收整流线路输入电压，并提供链路输出电压，且从所述开关模式升压级观测感测电流；以及 
目标电流发生器，用于接收所述链路输出电压，并产生与所述整流线路输入电压成比例的目标电流； 
波纹电流测定仪，用于产生估计峰-峰感应器波纹电流I的波纹电流； 
比较器，对所述目标电流、所述波纹电流和所述感测电流作出响应，用于输出两电平电流比较结果信号；以及 
对所述两电平电流比较结果信号作出响应的有限状态机，该有限状态机产生具有工作周期的开关控制信号，调节该工作周期，以控制所述开关，从而使所述感测电流与所述整流线路输入电压成比例，由此进行功率因数校正。 
28.根据权利要求27所述的功率因数校正器，其特征在于当所述感测电流小于所述目标电流与所述波纹电流一半的总和时，所述有限状态机接通所述开关达接通时间期间；当所述感测电流大于或等于所述目标电流与所述波纹电流一半的总和时，所述有限状态机断开所述开关达断开时间期间。 
29.根据权利要求28所述的功率因数校正器，其特征在于根据所述波纹电流调节所述断开时间期间。 
30.根据权利要求27所述的功率因数校正器，其特征在于还包括计时器，该计时器包括振荡器和计数器，且由所述有限状态机利用所述计时器，测量所述开关的接通时间期间和断开时间期间。 
31.根据权利要求27所述的功率因数校正器，其特征在于所述目标电流发生器、所述比较器和所述有限状态机被整合在单一的集成电路中。 
32.一种功率因数校正方法，其特征在于包括： 
将开关与感应器耦合在一起，形成开关模式升压级，该开关模式升压级接收整流线路输入电压，并提供链路输出电压；从所述开关模式升压级观测感测电流； 
接收链路输出电压； 
产生与所述整流线路输入电压成比例的目标电流； 
产生估计峰-峰感应器波纹电流I的波纹电流； 
作为对所述目标电流、所述波纹电流和所述感测电流作出的响应，产生两电平电流比较结果信号；以及 
作为对所述两电平电流比较结果信号作出的响应，产生具有工作周期的开关控制信号，调节该工作周期，以控制开关，从而使感测电流与整流线路输入电压成比例，由此进行功率因数校正。 
33.根据权利要求32所述的方法，其特征在于产生开关控制信号还包括： 
当所述感测电流小于所述目标电流与所述波纹电流一半的总和时，接通所述开关达接通时间期间；以及 
当所述感测电流大于或等于所述目标电流与所述波纹电流一半的总和时，断开所述开关达断开时间期间。 
34.如权利要求33所述的方法，其特征在于还包括： 
根据所述波纹电流，调节所述断开时间期间。 
35.一种合并有功率因数校正控制器的集成电路，其特征在于该功率因数校正控制器包括目标电流发生器、比较器和有限状态机，所述集成电路被配置如下： 
开关模式升压级接收整流线路输入电压，并提供链路输出电压；
接收代表整流线路输入电压的信号； 
从开关模式升压级观测感测电流； 
接收代表链路输出电压的另一信号； 
产生与所述整流线路输入电压成比例的目标电流； 
产生估计峰-峰感应器波纹电流I的波纹电流； 
开关模式升压级具有开关和与该开关耦合的感应器， 
作为对所述目标电流、所述波纹电流和所述感测电流作出的响应，产生两电平电流比较结果信号；以及 
作为对所述两电平电流比较结果信号作出的响应，产生具有工作周期的开关控制信号，调节该工作周期，以控制所述开关，从而使所述感测电流与所述整流线路输入电压成比例，由此进行功率因数校正。 
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