CN102424117A - Method for compensating phase lag of magnetic bearing of magnetic suspension control moment gyro - Google Patents

Abstract

The invention relates to a method for compensating phase lag of a magnetic bearing of a magnetic suspension control moment gyro, in particular to a method for compensating phase lag of a switching power amplifier of the magnetic bearing of the magnetic suspension control moment gyro. The method provided by the invention comprises the following steps of: calculating a rated nutation frequency according to rated revolution speed of a magnetic suspension control moment gyro rotor and system parameters, testing phase frequency characteristic of a power amplifier system and determining a phase angle required to be compensated at the rated nutation frequency, thus obtaining an asymmetry factor of a double-parallel asymmetric sampling resistance network on the basis of measuring equivalent inductance and resistance of a winding of the magnetic bearing, and finally configuring the double-parallel asymmetric sampling resistance network according to the asymmetry factor. The method provided by the invention belongs to the technical field of control on spacecraft inertia actuating mechanism and can be applied to high-stability control on the magnetic suspension control moment gyro at high rotating speed.

Description

A kind of method that compensates magnetic suspension control torque gyroscope magnetic bearing phase delay

Technical field

The present invention relates to a kind of method that compensates magnetic suspension control torque gyroscope magnetic bearing phase delay, be applicable to the high stable control under the high rotating speed of magnetic suspension control torque gyroscope, belong to spacecraft inertia actuating mechanism controls technical field.

Background technology

Magnetic suspension control torque gyroscope (MSCMG) does not have friction because of having, hangs down vibration, is easy to realize that outstanding advantage such as high precision and long life becomes the important development direction of spacecraft attitudes control inertia actuating units such as space station, space maneuver platform and quick maneuvering satellite.

In the MSCMG magnetic bearings control system, the effect of switch power amplifier is to provide corresponding electric current to produce needed electromagnetic force to the electromagnet bearing coil.The electromagnetic bearing switch power amplifier mainly contains pulse duration modulation (PWM) type, sampling maintenance, the chain rate that stagnates than four kinds of forms such as type and minimum pulse width types.Wherein, the advantage of PWM close power amplifier is that switching frequency is fixed, and can limit minimum conducting and the width that turn-offs pulse, and the output wave shape quality is good, stable state accuracy is high, reliability is high, in electromagnet bearing, has obtained widespread use.Electromagnet bearing PWM close power amplifier mainly comprises controller, PWM generator, full-bridge power circuit and current sensor.

Electromagnet bearing PWM close power amplifier is actual to be a current tracking control circuit, and it is control variable with the electric current, and its target is the actual current of output can be tried one's best follow the tracks of input control signal undistortedly.But the phase delay outwardness of electromagnetic bearing switch power amplifier, it mainly comprises digital control time-delay and controlled object time-delay.

The MSCMG rotor-support-foundation system has strong gyro effect, and that gyro effect is mainly reflected in nutating stability for the influence of system stability is stable with precession, and wherein nutating stability is the principal element that influences system stability.The basic reason of nutating unstability is the magnetic bearing control system phase delay.The magnetic bearing control system phase delay mainly comprises the phase delay that magnetic bearing power amplification system phase delay and various LPF link cause, and the phase delay of magnetic bearing power amplification system is the main aspect of whole magnetic bearing control system phase delay.Therefore compensate magnetic bearing power amplification system phase delay and become emphasis and the difficult point that improves magnetic suspension rotor system stability.

At present, the method for compensation electromagnetic bearing power amplification system phase delay mainly comprises methods such as various Prediction Control (like the Smith Prediction Control), anticipatory control control and the control of boosting.The biggest advantage of Prediction Control is that big latency issue is converted into the design problem that does not have big time-delay, yet Prediction Control needs known object math modeling accurately, and is difficult in the reality obtain.So the robustness of Prediction Control is relatively poor, when model is inaccurate, can cause poor system performance, even unstable.Phase angle anticipatory control method need seal in the anticipatory control link in passage, must increase the hardware circuit or the computed in software amount of system.Control method can increase the bandwidth of power amplification system to a certain extent though boost, and it has increased power consumption and hardware circuit inevitably, and can't fundamentally compensate the controlled object time-delay.In addition; Improve the proportionality coefficient of power amplification system controller and the phase delay that the current feedback coefficient also can reduce system to a certain extent; But, make its effect of phase compensation not remarkable at high frequency treatment because of its restriction that receives power amplifier magnification factor and power amplifier voltage and switching frequency etc.Therefore; The method of existing compensation electromagnetic bearing power amplifier phase delay or reduced the reliability of system because of the hardware circuit that has increased system; Because of the software complexity that has increased power amplification system has brought more calculating time-delay, and calculate the phase delay that the increase of delaying time has aggravated power amplification system conversely.

Summary of the invention

The objective of the invention is: overcoming existing compensation electromagnetic bearing close power amplifier phase delay method need increase the calculating time-delay or increase a large amount of hardware circuits and the deficiency of method complicacy; On the basis that only increases the little hardware circuit; Through two parallel asymmetric sampling resistor networks, realize effective compensation to MSCMG magnetic bearing power amplification system phase delay.

Technical solution of the present invention is: rated speed of rotation and system parameter according to the magnetic suspension control torque gyroscope rotor are calculated specified nutation frequency; The phase angle of testing the open loop phase-frequency characteristic of power amplification system and confirming to compensate at specified nutation frequency place; And on the basis of test magnetic bearing winding equivalent inductance and resistance, draw the dissymmetry factor of two parallel asymmetric sampling resistor networks, at last according to dissymmetry factor configuration pair parallel asymmetric sampling resistor networks.Specifically may further comprise the steps:

(1) according to the rated speed of rotation Ω of magnetic suspension control torque gyroscope rotor _nSpecified nutation frequency ω with system parameter calculating magnetic suspension rotor _n:

${\mathrm{\ω}}_n=\frac{1}{2}[\left(\frac{J_a}{J_r}{\mathrm{\Ω}}_n\right)+\sqrt{{\left(\frac{J_a}{J_r}{\mathrm{\Ω}}_n\right)}^2+\frac{{8l}_m(l_s{k}_i{\mathrm{\λ}}_k-l_m{k}_x)}{J_r}}]$

Wherein, J _a, J _rBe respectively the polar moment of inertia and the equator rotor inertia of rotor; l _m, l _sIt is respectively the magnetic suspension rotor center reaches the radial transducer center to the radial direction magnetic bearing center distance; k _i, k _xBe respectively the current stiffness and the displacement rigidity of magnetic bearing; λ _kIt is the amplitude of control current amplitude versus frequency characte i (j ω);

(2) phase-frequency characteristic of test magnetic bearing power amplification system obtains the phase delay θ of magnetic bearing power amplification system at specified nutation frequency place _n

(3) confirm the phase theta that the magnetic bearing switch power amplification system need compensate at specified nutation frequency place _c=θ _n-θ ₀, θ wherein ₀Be the limit value of close power amplifier system at specified nutation frequency place phase delay;

(4) adopt multi-meter and secohmmeter can measure the equivalent resistance R of bearing coil winding respectively _sAnd inductance L _s

(5) confirm the big or small R of current sampling resistor _m, current sample is with the ratio k of divider resistance _AbAnd the op amp of current sampling circuit gain k;

(6) dissymmetry factor

of definite two parallel asymmetric sampling resistor networks

(7) take into account the power consumption and the signal to noise ratio of sampling resistor network, according to the divider resistance R of the two parallel asymmetric sampling resistor networks of dissymmetry factor δ configuration _a, R _b, R _cAnd R _d, R wherein _a=R _d,

Principle of the present invention is: utilize the current detecting principle, in conjunction with this concrete object of electromagnetic bearing switch power amplifier, proposed the phase compensating method based on two parallel asymmetric sampling resistor networks.This phase compensating method has been realized the compensation to the close power amplifier phase place through the design of sampling resistor network degree of asymmetry when realizing current detecting.

Below the principle of the two parallel asymmetric sampling resistor network that the present invention adopted is explained.

According to the critical speed stability criterion of MSCMG, can get the specified nutation frequency ω of rotor _nWith rated speed of rotation Ω _nAnd satisfy between the system parameter following the relation:

${\mathrm{\ω}}_n=\frac{1}{2}[\left(\frac{J_a}{J_r}{\mathrm{\Ω}}_n\right)+\sqrt{{\left(\frac{J_a}{J_r}{\mathrm{\Ω}}_n\right)}^2+\frac{{8l}_m(l_s{k}_i{\mathrm{\λ}}_k-l_m{k}_x)}{J_r}}]$

J wherein _a, J _rBe respectively the polar moment of inertia and the equator rotor inertia of rotor; l _m, l _sBe respectively the magnetic suspension rotor center to the radial direction magnetic bearing center and to the radial transducer center apart from k _i, k _xBe respectively the current stiffness and the displacement rigidity of magnetic bearing; λ _kBe control current amplitude versus frequency characte i (j ω _n) amplitude.

Existing MSCMG magnetic bearing control system adopts two closed loop controlling structures of position ring and electric current loop usually, and wherein position ring is an outer shroud, adopts Decentralized PID to add the intersection controlling schemes usually.The phase place that PID control can compensate is less, mainly realize the leading in phase of system through intersection control, and the phase place that intersection control can compensate is usually in 90 °.Therefore, for the maintenance system has certain stability margin at specified nutation frequency place, the phase delay of close power amplifier system should limit within the specific limits, and the inventive method is used for compensating that part of phase delay that the magnetic bearing power amplification system surpasses limit value just.The limit value of phase delay is designated as θ at specified nutation frequency place ₀So can get the phase angle that the MSCMG power amplification system need compensate at specified nutation frequency place is θ _c=θ _n-θ ₀

Fig. 2 is the bipolarity PWM close power amplifier schematic diagram based on two parallel asymmetric sampling resistor networks; Its control signal and electric current loop feedback signal are done to export controlling quantity through controller after the difference; Carry out the PWM modulation then and generate the PWM ripple; The PWM ripple drives bipolarity H full-bridge inverter and comes the track reference electric current to produce corresponding control current, and last current detection circuit is realized the detection to coil current.R among the figure _M1And R _M2Be sampling resistor, R _A1, R _B1, R _C1, R _D1And R _M1Constitute asymmetric sampling resistor network I; R _A2, R _B2, R _C2, R _D2And R _M2Constitute asymmetric sampling resistor network II, and the satisfied R that concerns _A1=R _A2=R _a, R _B1=R _B2=R _b, R _C1=R _C2=R _c, R _D1=R _D2=R _dAnd R _M1=R _M2=R _m, i.e. symmetry fully between two sampling resistor networks.Asymmetric sampling resistor network I constitutes two parallel asymmetric sampling resistor networks with II; K is the op amp gain of current sampling circuit; i _CoIt is the electric current loop feedback factor; k _AmpIt is the proportionality coefficient of power amplification system controller (employing proportional regulator); k _aIt is the ratio magnification factor of PWM MOD; R _sAnd L _sBe respectively the resistance and the inductance of the equivalence of magnetic bearing winding; S ₁, S ₂, S ₃, S ₄Constitute four power switch pipes, wherein S of H full-bridge ₁And S ₃The last brachium pontis of forming the H full-bridge, S ₂And S ₄Form the following brachium pontis of H full-bridge, upper and lower bridge arm is according to pwm signal conducting successively and shutoff; S is digital select switch, works as S ₂And S ₃The sampled value of gating sampling resistor network I is worked as S during conducting ₁And S ₄The sampled value of gating sampling resistor network II during conducting.

S when power switch pipe ₂, S ₃Conducting, S ₁, S ₄The principle of equivalence of two parallel asymmetric sampling resistor networks is as shown in Figure 3 during shutoff, wherein U ₁And U ₂Represent winding terminal input section point voltage and the terminal voltage signal behind current detection circuit respectively, U _N1And U _N2Expression sampling resistor R _M1The node voltage at two ends, U ₃And U ₄Represent R respectively _A1And R _B1And R _C1And R _D1Between terminal voltage.Can get according to the nodal method of analysis:

$\left[\begin{array}{c}{\mathrm{<figure-callout\; id="1"\; label="U\; n"\; filenames="HDA0000105859200000012.png"\; state="\{\{state\}\}">U</figure-callout>}}_n=U_1\\ -{\mathrm{<figure-callout\; id="1"\; label="U\; n"\; filenames="HDA0000105859200000012.png"\; state="\{\{state\}\}">U</figure-callout>}}_n\&CenterDot;\frac{1}{R_m}+U_{n2}(\frac{1}{R_m}+\frac{1}{R_c+R_d}+\frac{1}{\frac{R_m(R_c+R_d)}{R_m+R_c+R_d}+{\mathrm{sL}}_s})\end{array},=0---\left(1\right)\right]$

Therefore,

${\mathrm{<figure-callout\; id="2"\; label="U\; n"\; filenames="HDA0000105859200000012.png"\; state="\{\{state\}\}">U</figure-callout>}}_n=\frac{U_{n1}}{R_m}\&CenterDot;\frac{1}{\frac{1}{R_m}+\frac{1}{R_d+R_c}+\frac{1}{R_s+\frac{R_m(R_c+R_d)}{R_m+R_c+R_d}+{\mathrm{sL}}_s}}---\left(2\right)$

In the formula, s representes the variable of the math modeling-transfer function of control system in complex domain.Because R _c+ R _d＞＞R _m, therefore,

${\mathrm{<figure-callout\; id="2"\; label="U\; n"\; filenames="HDA0000105859200000012.png"\; state="\{\{state\}\}">U</figure-callout>}}_n=\frac{U_{n1}}{R_m}\&CenterDot;\frac{1}{\frac{1}{R_m}+\frac{1}{R_d+R_c}+\frac{1}{R_s+R_m+{\mathrm{sL}}_s}}.$

Can get according to the nodal method of analysis,

$U_3=\frac{R_b}{R_a+R_b}{\mathrm{<figure-callout\; id="1"\; label="U\; n"\; filenames="HDA0000105859200000012.png"\; state="\{\{state\}\}">U</figure-callout>}}_n,$ ${\mathrm{<figure-callout\; id="4"\; label="U"\; filenames="HDA0000105859200000021.png"\; state="\{\{state\}\}">U</figure-callout>}}_4=\frac{R_c}{R_c+R_d}{U}_{n2}---\left(3\right)$

U in the formula ₃And U ₄Represent R respectively _A1And R _B1Between and R _C1And R _D1Between node voltage.Can get by (3),

${\mathrm{<figure-callout\; id="2"\; label="U"\; filenames="HDA0000105859200000012.png"\; state="\{\{state\}\}">U</figure-callout>}}_2=k(U_3-U_4)=\frac{{\mathrm{kR}}_b}{R_a+R_b}{U}_{n1}-\frac{{\mathrm{kR}}_c}{R_c+R_d}\&CenterDot;\frac{U_1}{R_m}\&CenterDot;\frac{1}{\frac{1}{R_m}+\frac{1}{R_c+R_d}+\frac{1}{R_s+R_m+{\mathrm{sL}}_s}}---\left(4\right)$

Therefore with U ₁Be input, U ₃Ssystem transfer function G for output ₁(s) can be expressed as:

$G_1\left(s\right)=\frac{U_2}{{\mathrm{<figure-callout\; id="1"\; label="U"\; filenames="HDA0000105859200000012.png"\; state="\{\{state\}\}">U</figure-callout>}}_1}=\frac{{\mathrm{kR}}_b}{R_a+R_b}-\frac{{\mathrm{kR}}_c}{R_c+R_d}\&CenterDot;\frac{1}{1+\frac{R_m}{R_c+R_d}+\frac{R_m}{R_s+R_m+{\mathrm{sL}}_s}}$

$=\frac{{\mathrm{kR}}_b}{R_a+R_b}-\frac{{\mathrm{kR}}_c(R_s+R_m+{\mathrm{sL}}_s)}{(R_d+R_c)(R_m+R_s+R_m)+R_m(R_s+R_m)+{\mathrm{sL}}_s(R_c+R_d+R_m)}---\left(5\right)$

Consider (R _c+ R _d) (R _m+ R _SmThe R of)＞＞ _m(R _s+ R _m) and R _c+ R _d＞＞R _m, definition

Following formula can further be reduced to:

$G_1\left(s\right)=\frac{U_2}{{\mathrm{<figure-callout\; id="1"\; label="U"\; filenames="HDA0000105859200000012.png"\; state="\{\{state\}\}">U</figure-callout>}}_1}=\frac{{\mathrm{kR}}_b}{R_a+R_b}-\frac{{\mathrm{kR}}_c}{R_c+R_d}\&CenterDot;\frac{R_s+R_m+{\mathrm{sL}}_s}{R_m+R_s+R_m+{\mathrm{sL}}_s}$

$={\mathrm{kk}}_{\mathrm{ab}}-{\mathrm{kk}}_{\mathrm{cd}}\frac{R_s+R_m+{\mathrm{sL}}_s}{R_m+R_s+R_m+{\mathrm{sL}}_s}$

$=k\frac{\mathrm{\&delta;}(R_m+R_s)+k_{\mathrm{ab}}{R}_m+L_s\mathrm{\&delta;s}}{R_m+R_s+R_m+{\mathrm{sL}}_s}---\left(6\right)$

Wherein δ is defined as the dissymmetry factor of two parallel asymmetric sampling resistor networks, δ=k _Ab-k _Cd

S when power switch pipe ₁, S ₄Conducting, S ₂, S ₃The principle of equivalence of two parallel asymmetric sampling resistor networks is as shown in Figure 4 during shutoff.As can be seen from the figure, it has the complete symmetrical structure with Fig. 3, therefore can get S ₁, S ₄Conducting, S ₂, S ₃The phase-frequency characteristic of power amplification system and S during shutoff ₂, S ₃Conducting, S ₁, S ₄Phase-frequency characteristic during shutoff is just the same.Therefore, which kind of conducting state no matter the bipolarity close power amplifier for having two parallel asymmetric sampling resistor networks be in, with U ₁Be input, U ₂For the ssystem transfer function of exporting can be expressed as G _p(s)=G ₁(s).

Fig. 5 is the equivalent structure figure with bipolarity close power amplifier of two parallel asymmetric sampling resistor networks, wherein G _p(s) expression is with U ₁Be input, U ₂Be the ssystem transfer function of output,

The transfer function of representing digital control time-delay, i _CoIt is the electric current loop feedback factor; k _AmpIt is the proportionality coefficient of power amplification system controller (employing proportional regulator); k _aIt is the ratio magnification factor of PWM MOD.The open loop transfer function G (s) of then whole close power amplifier system is:

$G\left(s\right)=k_a{k}_{\mathrm{amp}}{i}_{\mathrm{co}}{e}^{-T_{d}s}{G}_p\left(s\right)---\left(7\right)$

So,

$G\left(s\right)=k_{\mathrm{amp}}{k}_a{\mathrm{ki}}_{\mathrm{co}}{e}^{-T_{d}s}\frac{\mathrm{\&delta;}(R_m+R_s)+k_{\mathrm{ab}}{R}_m+L_s\mathrm{\&delta;s}}{R_m+R_s+R_m+{\mathrm{sL}}_s}---\left(8\right)$

Symmetric sampling resistance network for traditional has δ=k _Ab-k _Cd=0, its open loop transfer function G ₀(s) can be expressed as:

$G_0\left(s\right)=k_{\mathrm{amp}}{k}_a{\mathrm{ki}}_{\mathrm{co}}{e}^{-T_{d}s}\frac{k_{\mathrm{ab}}{R}_m}{R_m+R_s+R_m+{\mathrm{sL}}_s}---\left(10\right)$

Then its phase-frequency characteristic

is:

When δ ≠ 0, the phase-frequency characteristic of G (s) can be expressed as:

Therefore, when δ＞0, two parallel asymmetric sampling resistor networks are at specified nutation frequency ω _nThe place can provide phase lead compensation, its offset angle θ _cCan be expressed as:

Can obtain dissymmetry factor according to (13):

$\mathrm{\&delta;}=\frac{k_{\mathrm{ab}}{R}_m\tan {\mathrm{\&theta;}}_c}{L_s{\mathrm{\&omega;}}_n-(R_m+R_s)\tan {\mathrm{\&theta;}}_c}---\left(14\right)$

Can get according to formula (14), obtain dissymmetry factor δ, need to confirm sampling resistor R _m, asymmetric divider resistance ratio k _Ab, magnetic bearing winding equivalent inductance L _sAnd resistance R _sAnd the specified nutation frequency ω of magnetic suspension control torque gyroscope _nWherein, L _sBig I get through actual test; R _m, k _AbSelection to combine the scope of magnetic bearing winding current, the size and the current sample in power amplifier voltage source to take all factors into consideration with the sample range of AD chip.

On this basis, take into account the power consumption and the signal to noise ratio of asymmetric sampling resistor network, according to the divider resistance R of the two parallel asymmetric sampling resistor networks of dissymmetry factor δ configuration _a, R _b, R _cAnd R _dFor the power consumption that reduces system as far as possible and improve signal to noise ratio, R should be arranged _a＞＞R _m, and choose R for simplicity usually _a=R _d, according to

With

Obtain R _bAnd R _c

The present invention is with the advantage that the method for existing compensation magnetic bearing power amplification system phase delay is compared: the inventive method had not both increased any hardware circuit, need not to adopt the control algorithm of various complicacies yet, therefore was convenient to practical applications.

Description of drawings

Fig. 1 is the diagram of circuit of phase compensating method of the current mode switch power amplifier of a kind of magnetic suspension control torque gyroscope magnetic bearing of the present invention;

Fig. 2 is the magnetic bearing power amplification system functional block diagram based on two parallel asymmetric sampling resistor networks;

Fig. 3 works as S for the inventive method ₂, S ₃Conducting, S ₁, S ₄Topology diagram during shutoff;

Fig. 4 works as S for the inventive method ₁, S ₄Conducting, S ₂, S ₃Topology diagram during shutoff;

Fig. 5 is an electromagnetic bearing switch power amplification system equivalence closed loop controlling structure block diagram of the present invention;

Fig. 6 is the open loop frequency performance diagram of the electromagnet bearing power amplification system before and after employing the present invention.

The specific embodiment

As shown in Figure 1, in the practical implementation process, practical implementation step of the present invention is following:

1, according to the rated speed of rotation Ω of magnetic suspension control torque gyroscope rotor _nCalculate specified nutation frequency ω with system parameter _n:

${\mathrm{\&omega;}}_n=\frac{1}{2}[\left(\frac{J_a}{J_r}{\mathrm{\&Omega;}}_n\right)+\sqrt{{\left(\frac{J_a}{J_r}{\mathrm{\&Omega;}}_n\right)}^2+\frac{{8l}_m(l_s{k}_i{\mathrm{\&lambda;}}_k-l_m{k}_x)}{J_r}}]$

Wherein, J _a, J _rBe respectively the polar moment of inertia and the equator rotor inertia of rotor; l _m, l _sIt is respectively the magnetic suspension rotor center reaches radial transducer to the radial direction magnetic bearing center distance; k _i, k _xBe respectively the current stiffness and the displacement rigidity of magnetic bearing; λ _kBe control current amplitude versus frequency characte i (j ω _n) amplitude.

2, the phase-frequency characteristic of test magnetic bearing power amplification system obtains the phase delay θ of magnetic bearing power amplification system at specified nutation frequency place _n

Adopt digital signal analyser (like Agilent 35670A etc.) to carry out frequency-response analysis, can obtain system at specified nutation frequency ω through the phase frequency characteristic curve to the magnetic bearing power amplification system _nThe phase delay θ at place _n

3, confirm the phase theta that magnetic bearing power amplifier electric current loop need compensate at specified nutation frequency place _c

The magnetic bearing control system phase delay mainly comprises the phase delay that magnetic bearing power amplification system phase delay and various LPF link cause.Existing MSCMG magnetic bearing control system adopts two closed loop controlling structures of position ring and electric current loop usually, and wherein position ring is an outer shroud, adopts Decentralized PID to add the intersection controlling schemes usually, and Decentralized PID adds intersection and controls the phase place that can compensate usually in 90 °.Electric current loop is interior ring, i.e. the close power amplifier link.For the system that makes keeps stable, control system should have certain phase margin θ at specified nutation frequency place _a(common θ _a＞30 °).Because the leading in phase that control system can provide is limited, so the phase delay of close power amplifier system will limit within the specific limits.The limit value of phase delay is designated as θ at specified nutation frequency place ₀, the inventive method is used for compensating that part of phase delay that the magnetic bearing power amplification system surpasses limit value just.So can get the phase angle that the MSCMG power amplification system need compensate at specified nutation frequency place is θ _c=θ _n-θ ₀

4, in radially load-bearing and dropping under the state of protection on the bearing of magnetic suspension rotor, adopt multi-meter and secohmmeter can measure the equivalent resistance R of bearing coil winding respectively _sAnd inductance L _s

5, confirm the big or small R of current sampling resistor _m, current sample is with the ratio k of divider resistance _AbAnd the op amp of current sampling circuit gain k.

R _m, k _AbWant power consumption, signal to noise ratio, winding current magnitude range and the sampling of comprehensively sampling resistance network to take all factors into consideration with choosing of k with the sample range of AD chip.Too big R _mThe power consumption that makes sampling network is increased, and too small R _mMake that again the signal to noise ratio of current sample network is less, usually R _mThe size of size bearing winding equivalent resistance is on the same order of magnitude and slightly less than normal.The winding current magnitude range multiply by k _AbK should and have certain allowance (being generally about 20%) within sampling is with the sample range of AD chip, and for reaching higher signal to noise ratio, k _AbUsually be chosen between 1/8 to 1/4.Equivalent resistance such as the bearing winding is R _s=2.4 Ω, the winding current magnitude range is-3.1A-3.1A that it is-5V-5V that then sampling resistor is chosen as R that the sample range of AD chip is used in sampling _m=1 Ω, current sample use the ratio selectable of divider resistance to be k _Ab=1/6, press the sampling allowance of AD chip 20% and calculate, can get the op amp gain of current sampling circuit

6, confirm the dissymmetry factor δ of asymmetric sampling resistor network.

With R _m, k _Ab, L _s, ω _nSubstitution

In can get δ, L wherein _sSize for radial direction magnetic bearing winding equivalent inductance.

7, take into account the power consumption and the signal to noise ratio of asymmetric sampling resistor network, according to the divider resistance R of the two parallel asymmetric sampling resistor networks of dissymmetry factor δ configuration _a, R _b, R _cAnd R _d

For the power consumption that reduces system as far as possible and improve signal to noise ratio, 1000R should be arranged _m＜R _a＜10000R _mAnd 1000R＜R _a＜10000R, and choose R for simplicity usually _a=R _d, according to

With

Obtain R _bAnd R _c

For the validity of checking the inventive method, with magnetic suspension rotor rated speed of rotation Ω _nThe magnetic suspension control torque gyroscope of=20000r/min is that example is verified.System parameter is following: l _m=62.5mm, l _s=67.2mm, J _a=0.1019kgm ², J _r=0.062kgm ², k _i=269N/A, k _x=-1.9N/ μ m, R _m=1 Ω, R _s=2.4 Ω, L _s=24mH, λ _k=1505, k=7.7, k _a=3.2, i _Co=10, k _Ab=1/6; The electromagnet bearing power amplification system adopts P control, and its parameter is k _Amp=2.5, system delay T _d=150 μ s.ω _n＝600Hz，θ _n＝95°，θ ₀＝50°，θ _c＝θ _n-θ ₀＝95°-50°＝45°。

R _a＝R _d＝7.5kΩ，R _b＝1.5kΩ，R _c＝1.473kΩ。

The frequency characteristic of the magnetic bearing power amplification system before and after employing the inventive method compensates is respectively shown in fine rule and thick line among Fig. 6.Wherein abscissa is represented frequency, and unit is Hz, and ordinate is represented phase place, and unit is Degrees.Can get from Fig. 6, cooresponding ordinate is-53 ° to thick line in specified nutation frequency 600Hz place, and cooresponding ordinate is-95 ° and fine rule is in frequency 600Hz place, so the system phase compensation value is 42 °, has reached the effect of phase compensation.

The content of not doing in the specification sheets of the present invention to describe in detail belongs to this area professional and technical personnel's known prior art.

Claims (1)

1. method that compensates magnetic suspension control torque gyroscope magnetic bearing phase delay; It is characterized in that: rated speed of rotation and system parameter according to the magnetic suspension control torque gyroscope rotor are calculated specified nutation frequency; The phase angle of testing the open loop phase-frequency characteristic of power amplification system and confirming to compensate at specified nutation frequency place; And on the basis of test magnetic bearing winding equivalent inductance and resistance, draw the dissymmetry factor of pair parallel asymmetric sampling resistor networks; According to the two parallel asymmetric sampling resistor networks of dissymmetry factor configuration, specifically may further comprise the steps at last:

(1) according to the rated speed of rotation Ω of magnetic suspension control torque gyroscope rotor _nSpecified nutation frequency ω with system parameter calculating magnetic suspension rotor _n:

${\mathrm{\&omega;}}_n=\frac{1}{2}[\left(\frac{J_a}{J_r}{\mathrm{\&Omega;}}_n\right)+\sqrt{{\left(\frac{J_a}{J_r}{\mathrm{\&Omega;}}_n\right)}^2+\frac{{8l}_m(l_s{k}_i{\mathrm{\&lambda;}}_k-l_m{k}_x)}{J_r}}]$

Wherein, J _a, J _rBe respectively the polar moment of inertia and the equator rotor inertia of rotor; l _m, l _sIt is respectively the magnetic suspension rotor center reaches the radial transducer center to the radial direction magnetic bearing center distance; k _i, k _xBe respectively the current stiffness and the displacement rigidity of magnetic bearing; λ _kIt is the amplitude of control current amplitude versus frequency characte i (j ω);

(2) phase-frequency characteristic of test magnetic bearing power amplification system obtains the phase delay θ of magnetic bearing power amplification system at specified nutation frequency place _n

(3) confirm the phase theta that the magnetic bearing switch power amplification system need compensate at specified nutation frequency place _c=θ _n-θ ₀, θ wherein ₀Be the limit value of close power amplifier system at specified nutation frequency place phase delay;

(4) adopt multi-meter and secohmmeter can measure the equivalent resistance R of bearing coil winding respectively _sAnd inductance L _s

(5) confirm the big or small R of current sampling resistor _m, current sample is with the ratio k of divider resistance _AbAnd the op amp of current sampling circuit gain k;

(6) dissymmetry factor

of definite two parallel asymmetric sampling resistor networks

(7) take into account the power consumption and the signal to noise ratio of sampling resistor network, according to the divider resistance R of the two parallel asymmetric sampling resistor networks of dissymmetry factor δ configuration _a, R _b, R _cAnd R _d, R wherein _a=R _d,

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