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Publication numberUS20010002889 A1
Publication typeApplication
Application numberUS 09/735,330
Publication dateJun 7, 2001
Filing dateDec 11, 2000
Priority dateMay 13, 1997
Also published asUS5877993, US5982686, US5982687, US6011731, US6011736, US6026040, US6028799, US6052322, US6181617, US6188622, US6198676, US6226210, US6262927, US6335888, US6353564, US6418071, US6445629, US6452846, US6469944, US6600687, US6778452, US6882587, US7054208, US20010002888, US20010004333, US20010009522, US20020018381, US20030021171, US20040095822, US20040240286, US20050152195
Publication number09735330, 735330, US 2001/0002889 A1, US 2001/002889 A1, US 20010002889 A1, US 20010002889A1, US 2001002889 A1, US 2001002889A1, US-A1-20010002889, US-A1-2001002889, US2001/0002889A1, US2001/002889A1, US20010002889 A1, US20010002889A1, US2001002889 A1, US2001002889A1
InventorsKurt Beigel, Manny Ma, Gordon Roberts, James Miller, Daryl Habersetzer, Jeffrey Bruce, Eric Stubbs
Original AssigneeBeigel Kurt D., Ma Manny K., Roberts Gordon D., Miller James E., Habersetzer Daryl L., Bruce Jeffrey D., Stubbs Eric T.
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Method of stressing a memory device
US 20010002889 A1
Abstract
As part of a memory array, a circuit is provided for altering the drive applied to an access transistor that regulates electrical communication within the memory array. In one embodiment, the circuit is used to alter the drive applied to a sense amp's voltage-pulling transistor, thereby allowing modification of the voltage-pulling rate for components of the sense amp. A sample of test data is written to the memory array and read several times at varying drive rates in order to determine the sense amp's ability to accommodate external circuitry. In another embodiment, the circuit is used to alter the drive applied to a bleeder device that regulates communication between the digit lines of the memory array and its cell plate. Slowing said communication allows defects within the memory array to have a more pronounced effect and hence increases the chances of finding such defects during testing. The circuit is configured to accept and apply a plurality of voltages, either through a contact pad or from a series of discrete voltage sources coupled to the circuit.
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Claims(138)
What is claimed is:
1. A test mode drive-modifying device for a memory array having at least one sense amp coupled to a voltage-pulling mechanism, comprising:
a test voltage alteration mechanism, wherein:
said test voltage alteration mechanism is coupled to said voltage-pulling mechanism; and
said test voltage alteration mechanism is configured to receive a plurality of voltages.
2. The test mode drive-modifying device in
claim 1
, wherein said sense amp is a pulldown sense amp.
3. The test mode drive-modifying device in
claim 2
, wherein said voltage-pulling mechanism is a transistor.
4. The test mode drive-modifying device in
claim 3
, wherein said voltage-pulling mechanism is an n-channel transistor.
5. A test mode driver circuit for a voltage-pulling transistor of a sense amp, comprising:
a conductive path coupled to a gate of said voltage-pulling transistor; and
an apparatus configured to receive a range of voltages coupled to said conductive path.
6. The test mode driver circuit in
claim 5
, wherein said apparatus is a contact pad.
7. A voltage-pulling circuit for a sense amp, comprising:
a main conductive path coupled to said sense amp; and
a plurality of secondary conductive paths selectively coupled to said main conductive path, wherein each of said plurality of secondary conductive paths is configured to receive a generally discrete voltage.
8. The voltage-pulling circuit in
claim 7
, wherein said sense amp is a pullup sense amp.
9. A driving circuit for a voltage-pulling transistor of a sense amp, comprising:
a selective coupling apparatus coupled to and electrically interposed between:
a test conduit configured to accept a plurality of voltage signals; and
an output node configured to couple to a gate of said voltage-pulling transistor; wherein
said selective coupling apparatus has a first mode of operation and is configured to receive a signal that has a first value during said first mode of operation, and
wherein said selective coupling apparatus is further configured to allow electrical communication between said test conduit and said output node in response to said first value.
10. The circuit in
claim 9
, further comprising a main conduit coupled to said selective coupling apparatus and configured to receive a voltage source, and wherein:
said selective coupling apparatus has a second mode of operation;
said signal has a second value during said second mode of operation; and
said selective coupling apparatus is configured to allow electrical communication between said main conduit and said output node in response to said second value.
11. The circuit in
claim 10
, wherein said selective coupling apparatus is configured to prevent electrical communication between said test conduit and said output node in response to said second value.
12. The circuit in
claim 11
, wherein said selective coupling apparatus is configured to prevent electrical communication between said main conduit and said output node in response to said first value.
13. The circuit in
claim 12
, wherein said second value is inverse to said first value.
14. A voltage regulator for a voltage-pulling transistor of a sense amp, wherein said voltage-pulling transistor is driven by an inverter circuit having a p-channel transistor and an n-channel transistor, comprising:
a common node configured to couple to said inverter circuit;
a first electrical connection device electrically interposed between said common node and a first pathway, wherein:
said first pathway is configured to accept a first voltage, and
said first electrical connection device is configured to accept a first vector and to electrically connect said first pathway with said common node in response to said first vector; and
a second electrical connection device electrically interposed between said common node and a second pathway, wherein:
said second pathway is configured to accept a second voltage, and
said second electrical connection device is configured to accept a second vector and electrically connect said second pathway with said common node in response to said second vector.
15. The voltage regulator in
claim 14
, wherein said voltage regulator is configured to accept one of said first and second vectors at a time.
16. The voltage regulator of
claim 15
, wherein said voltage regulator is a package part of an integrated circuit.
17. The voltage regulator in
claim 16
, wherein said common node is coupled to said p-channel transistor.
18. The voltage regulator of
claim 15
, further comprising a latch device connected to said second electrical connection device, wherein:
said latch device is configured to receive a test vector; and
said latch device is further configured to transmit said second vector to said second electrical connection device in response to said test vector.
19. The voltage regulator of
claim 18
, wherein said latch device further comprises:
an inverter connected to said second electrical connection device, wherein:
said inverter is configured to receive said test vector, and
said inverter is further configured to transmit said second vector to said second electrical connection device in response to said test vector; and
an activated transistor electrically interposed between said second electrical connection device and said inverter.
20. The voltage regulator of
claim 19
, further comprising a replenish device coupled to said second electrical connection device and to said common node, wherein said replenish device is configured to substantially maintain a potential at said second electrical connection device.
21. The voltage regulator of
claim 20
, wherein said replenish device is a capacitor;
22. The voltage regulator of
claim 21
, wherein:
said replenish device comprises a second n-channel transistor having a gate, a drain and a source;
said drain and said source are coupled together at a capacitance node;
said capacitance node is coupled to said common node; and
said gate is coupled to said second electrical connection device.
23. A margin-range apparatus for a sense amp's voltage-pulling transistor driven by an inverter circuit having a p-channel transistor coupled to an n-channel transistor, comprising:
a voltage reception device, wherein:
said voltage reception device is configured to couple to said inverter circuit, and
said voltage reception device is selectively coupled to a first test voltage path and a second test voltage path.
24. The apparatus in
claim 23
, wherein said voltage reception device is selectively coupled to a ground path.
25. The apparatus in
claim 24
, wherein said voltage reception device is coupled to said n-channel transistor.
26. A current saturation test device for a sense amp having a pullup transistor gated by an inverter comprising a p-channel transistor and an n-channel transistor, wherein said device comprises:
a first terminal configured to couple to said inverter and adapted to receive a generally constant potential; and
a second terminal configured to couple to said inverter and adapted to receive a plurality of voltage potentials.
27. The device in
claim 26
, wherein said second terminal is coupled to:
a first test path adapted to receive a first test voltage;
a second test path adapted to receive a second test voltage; and
a non-test path coupled to a potential node.
28. The device in
claim 27
, further comprising:
a first selection device coupled to said first test path and electrically interposed between said second terminal and said first test voltage, wherein said first selection device is configured to activate in response to a first test signal;
a second selection device coupled to said second test path and electrically interposed between said second terminal and said second test voltage, wherein said second selection device is configured to activate in response to a second test signal; and
a third selection device coupled to said non-test path, electrically interposed between said second terminal and said potential node, and configured to activate in response to a non-test signal.
29. The device in
claim 28
, further comprising a logic unit having said first test signal as a first input, said second test signal as a second input, and having a non-test signal output coupled to said third selection device.
30. The device in
claim 29
, where said device is configured to avoid simultaneous activation of said first selection device and said second selection device.
31. The device in
claim 30
, wherein:
said first terminal is coupled to said p-channel transistor;
said second terminal is coupled to said n-channel transistor;
said potential node of said non-test path is configured to couple to ground; and
said logic unit is a NOR gate.
32. The device in
claim 30
, wherein:
said first terminal is coupled to said n-channel transistor;
said second terminal is coupled to said p-channel transistor;
said potential node of said non-test path is configured to couple to a non-test voltage source; and
said logic unit is a NAND gate.
33. A driver circuit for a voltage-pulling device comprising:
a first plurality of potential nodes;
a signaling circuit configured to couple to said voltage-pulling device and selectively communicative with one of said first plurality of potential nodes; and
a second plurality of potential nodes selectively communicative with said signaling circuit.
34. The driver circuit in
claim 33
, wherein:
said signaling circuit comprises an inverter having a p-channel transistor coupled to an n-channel transistor;
said p-channel transistor is selectively coupled to said first plurality of potential nodes; and
said second plurality of potential nodes is selectively coupled to said n-channel transistor.
35. A test circuit for a sense amp, comprising:
a driver circuit; and
a coupler circuit connected to said driver circuit and configured to selectively receive a first voltage source and a second voltage source, wherein said first voltage source is electrically discrete from said second voltage source.
36. A method for testing a semiconductor memory device having at least one sense amp, comprising:
writing original data to said memory device;
driving said sense amp with a first voltage;
reading a first sample of stored data from said memory device;
driving said sense amp with a different voltage;
reading an additional sample of stored data from said memory device; and
comparing said first sample of stored data and said additional sample of stored data with said original data.
37. A method for determining the range of margins that a voltage-pulling sense amp transistor of a memory array is capable of accommodating, comprising:
entering an input test pattern into said memory array;
reading at least two output test patterns using at least two voltage levels driving said voltage-pulling sense amp transistor; and
comparing said input test pattern with said output test patterns.
38. A method for determining a lowest reliable level of a supply voltage capable of driving a sense amp in a memory array, comprising:
writing initial test data to said memory array;
initiating a test mode;
taking a plurality of test readings from said memory array;
decreasing said supply voltage driving said sense amp for each of said plurality of test readings; and
repeating taking test readings and decreasing said supply voltage.
39. The method in
claim 38
, wherein said repeating further comprises repeating taking test readings and decreasing said supply voltage until current saturation occurs.
40. The method in
claim 39
, wherein initiating a test mode comprises providing electrical communication between a test pathway and said sense amp.
41. The method in
claim 40
, wherein initiating a test mode further comprises preventing electrical communication between a non-test operation pathway and said sense amp.
42. The method in
claim 41
, wherein decreasing said supply voltage comprises decreasing said supply voltage through a range of voltages.
43. A method for determining a highest reliable level of a supply voltage that can drive a sense amp in a memory array, comprising:
writing initial test data to said memory array;
taking a plurality of test readings from said memory array;
increasing said supply voltage driving said sense amp for each of said plurality of test readings; and
repeating taking test readings and increasing said supply voltage until capacitive coupling occurs.
44. The method in
claim 43
, wherein increasing said supply voltage comprises increasing said supply voltage through a series of discrete values.
45. A method of test driving a sense amp voltage puller for at least one memory cell, comprising:
entering an input value to said memory cell; providing a plurality of driving pathways to said sense amp voltage puller;
configuring each driving pathway of said plurality of driving pathways to accept at least a respective plurality of voltage sources;
associating said plurality of driving pathways with a plurality of test vectors;
transmitting a test vector corresponding to one driving pathway of said plurality of driving pathways;
enabling electrical communication between said sense amp voltage puller and said driving pathway corresponding to said test vector;
reading a first sample of output data;
transmitting a different test vector corresponding to a different driving pathway;
enabling electrical communication between said sense amp voltage puller and said different driving pathway; and
reading a second sample of output data.
46. The method in
claim 45
, wherein enabling electrical communication comprises enabling electrical communication through one of said driving pathways at a time.
47. A method of determining the margin that a voltage-pulling transistor of a memory array is capable of accommodating, comprising:
entering an input data pattern to said memory array;
repeatedly reading an output data pattern from said memory array;
changing a voltage level driving said voltage-pulling transistor for each reading; and
tracking any differences between said output data pattern and said input data pattern.
48. A method for determining operable drive levels for a pulldown sense amp accommodating an operations circuit, wherein said pulldown sense amp has a first portion at a first voltage level and a second portion at a second voltage level, comprising:
setting an initial drive level for said pulldown sense amp;
pulling down said first voltage level for a first time;
establishing a new drive level for said pulldown sense amp;
pulling down said first voltage level for a second time; and
selectively comparing one time span required to pull down said first voltage level with a time span for accessing one of said first and second portions by said operations circuit.
49. The method in
claim 48
, further comprising equilibrating said first voltage level and said second voltage level before each pulling.
50. A method for determining the capabilities of a pullup transistor serving external circuitry, wherein said pullup transistor is coupled to at least one sense amp, comprising:
initiating a test drive voltage for said pullup transistor;
pulling up a digit-line voltage of a component of said sense amp associated with said pullup transistor;
comparing a length of time needed to pull up said digit-line voltage with a length of time required to drive said external circuitry;
altering said test drive voltage; and
repeating said pulling and comparing.
51. The method in
claim 50
, further comprising altering said test drive voltage before every repetition of said pulling.
52. The method in
claim 51
further comprising establishing a non-test drive voltage within a range capable of pulling up said digit-line voltage within said length of time required to drive said external circuitry.
53. The method in
claim 52
, wherein said non-test drive voltage is generally constant.
54. A method of changing the capability of a sense amp driver during a test mode, comprising:
providing a first potential node for said sense amp driver, wherein said first potential node is at a first voltage;
providing a second potential node for said sense amp driver, wherein said second potential node is at a second voltage; and
changing said first voltage of said first potential node.
55. The method in
claim 54
, further comprising changing said second voltage of said second potential node.
56. A method of regulating a control device within a semiconductor device, comprising:
driving said control device with a first voltage;
performing a first operation on said semiconductor device;
driving said control device with a second voltage; and
performing a second operation on said semiconductor device.
57. A method of testing a memory device, comprising:
performing a plurality of readings on said memory device; and
initiating a voltage change within said memory device between each reading of said plurality of readings.
58. A voltage regulator for a semiconductor device, comprising:
a terminal configured to receive a first voltage potential and a second voltage potential;
an access transistor coupled to said terminal and configured to activate in response to receiving a test mode signal; and
a connection node coupled to said access transistor and configured to couple to said semiconductor device.
59. The voltage regulator in
claim 58
, wherein:
said semiconductor device is configured to operate based on a source voltage VCC signal;
said access transistor has a gate;
said test mode signal has a potential generally equal to said VCC signal having undergone a charge pumping process, and wherein said test mode signal is configured to drive said gate of said access transistor; and
said voltage regulator further comprises a bootstrap device coupled to said gate of said access transistor and to said connection node, wherein said bootstrap device is configured to prevent said gate of said access transistor from substantially discharging during a transmission of said test mode signal.
60. The voltage regulator in
claim 59
, wherein:
said semiconductor device includes a cell plate; and
said access transistor is configured to couple to said cell plate.
61. The voltage regulator in
claim 60
, wherein:
said semiconductor device includes a digit line; and
said access transistor is configured to couple to said digit line.
62. A voltage variance test circuit for a semiconductor device having a connection node, comprising:
a first terminal configured to receive a first voltage potential and coupled to a first exclusively operable access device, wherein said first exclusively operable access device is coupled to said connection node; and
a second terminal configured to receive a second voltage potential and coupled to a second exclusively operable access device, wherein said second exclusively operable access device is coupled to said connection node.
63. The voltage variance test circuit in
claim 62
, wherein:
said first exclusively operable access device comprises a first transistor configured to activate in response to a reception of a first test signal;
said second exclusively operable access device comprises a second transistor configured to activate in response to a reception of a second test signal; and
at most one of said first test signal and said second test signal are received at any time.
64. The voltage variance test circuit in
claim 63
, wherein said semiconductor device has a voltage pulling sense amp transistor, and said connection node is coupled to said voltage pulling sense amp transistor.
65. A regulator for an equilibration circuit, comprising:
a first potential node configured to receive a first voltage source;
a second potential node configured to receive a second voltage source;
a common node selectively coupled to said first potential node and to said second potential node; and
an output node coupled to said common node and configured to couple to said equilibration circuit.
66. The regulator in
claim 65
, wherein said equilibration circuit has a test mode and a non-test mode, and wherein:
said common node is coupled to said first potential node during said non-test mode; and
said common node is selectively coupled to said first potential node and to said second potential node during said test mode.
67. A test driver for a bleeder device joining a cell plate signal generator to a memory array, comprising:
a plurality of conductive paths configured to receive a plurality of voltage sources;
a plurality of communication devices respectively coupled to said plurality of conductive paths, wherein each communication device of said plurality of communication devices is configured to activate to the exclusion of all other communication devices of said plurality of communication devices; and
an output terminal coupled to said plurality of communication devices and configured to couple to said bleeder device.
68. A charge rate regulator for a cell plate generator coupled to a digit line pair of a memory array, comprising:
a main transmission device electrically interposed between said cell plate generator and said digit line pair; and
a drive device coupled to said main transmission device and configured to selectively receive a plurality of voltage sources.
69. The charge rate regulator in
claim 68
, wherein said drive device further comprises:
a plurality of secondary transmission devices, wherein each secondary transmission device of said plurality of secondary transmission devices has a first end and a second end, wherein:
said first end of each secondary transmission device is coupled to said main transmission device, and
said second end of each secondary transmission device is configured to receive a respective voltage source from said plurality of voltage sources; and
a selective communication circuit coupled to said plurality of secondary transmission devices and electrically interposed between said plurality of secondary transmission devices and said plurality of voltage sources.
70. The charge rate regulator in
claim 69
, wherein said selective communication circuit is configured to activate one secondary transmission device of said plurality of secondary transmission devices at a time.
71. A test mode driver circuit for a regulator device of a cell plate signal, comprising:
a conductive path coupled to said regulator device; and
a voltage range receiver coupled to said conductive path.
72. The test mode driver circuit in
claim 71
, wherein said voltage range receiver is a contact pad.
73. A voltage regulator for a memory circuit including an equilibration device, a digit line pair, and a memory cell, comprising:
a voltage reception device, wherein:
said voltage reception device is configured to couple to said memory circuit; and
said voltage reception device is selectively electrically communicative with a first test voltage path and a second test voltage path.
74. The voltage regulator in
claim 73
, wherein said voltage reception device is configured to electrically interpose between said equilibration device and said digit line pair.
75. A defect testing device for a memory array having a cell plate signal device, comprising:
a first terminal configured to couple to said cell plate signal device and configured to receive a voltage potential; and
a second terminal configured to couple to said cell plate signal device and configured to receive a plurality of voltage potentials.
76. The device in
claim 75
, wherein said second terminal is coupled to:
a first test path configured to receive a first test voltage; and
a second test path configured to receive a second test voltage.
77. The device in
claim 76
, further comprising:
a first isolation device electrically interposed between said first test path and said second terminal, wherein said first isolation device has an active mode and an inactive mode; and
a second isolation device electrically interposed between said second test path and said second terminal, wherein:
said second isolation device has an active mode complementary to said active mode of said first isolation device, and
said second isolation device has an inactive mode complementary to said inactive mode of said first isolation device.
78. The device in
claim 77
, further comprising a third isolation device electrically interposed between said first terminal and said cell plate signal device, wherein all but one of said first, second, and third isolation devices are configured to operate simultaneously.
79. A voltage regulator for a cell plate signal of a memory array, wherein said cell plate signal is transmitted through a conductive path, and a control device is coupled to said conductive path, comprising:
a first voltage node having a generally constant potential;
a first latching device coupled to said first voltage node and to said control device;
a second voltage node having a variable potential; and
a second latching device coupled to said second voltage node and configured to couple to said control device, wherein said first latching device and said second latching device are selectively operable.
80. The voltage regulator in
claim 79
, wherein said second voltage node is coupled to a contact pad.
81. The voltage regulator in
claim 80
, wherein said first latching device comprises a transistor.
82. The voltage regulator in
claim 81
, wherein said second latching device comprises:
a test signal path coupled to said control device;
a switching device coupled to said test signal path and to said second voltage node and configured to allow electrical communication between said second voltage node and said test signal path in response to a reception of a driving signal;
a driving device coupled to said switching device and configured to receive a test signal and transmit said driving signal in response to a reception of said test signal; and
a driving signal maintenance device coupled to said switching device and to said test signal path.
83. The voltage regulator in
claim 82
, wherein said driving device further comprises:
a test initiator configured to receive said test signal and transmit said driving signal in response to a reception of said test signal; and
an output device coupled to said test initiator and said switching device, wherein said output device is configured to receive and output said driving signal.
84. The voltage regulator in
claim 83
, wherein:
said memory array is configured to accommodate an external circuit operating on a source voltage VCC; and
said test initiator and said output device are driven by a voltage VCCP, wherein said voltage VCCP is greater than said source voltage VCC.
85. The voltage regulator in
claim 84
, wherein:
said control device is a transistor;
said switching device is a transistor;
said driving signal maintenance device is a capacitor;
said test initiator is an inverter; and
said output device is a transistor.
86. An equilibration regulator coupled to a semiconductor device having a cell plate generator circuit and a digit line pair, comprising:
a selective voltage circuit coupled to said cell plate generator circuit and said digit line pair, wherein said selective voltage circuit is configured to receive a plurality of driving voltages, and wherein:
a selection of at least one driving voltage from said plurality of driving voltages is a test voltage;
one of said plurality of driving voltages is a defect compensation voltage; and
said defect compensation voltage is a non-test voltage.
87. The equilibration regulator of
claim 86
, wherein said semiconductor device is generally driven by a voltage VCC, and wherein:
said defect compensation voltage has a potential greater than VCC and is configured to counteract a defect in said semiconductor device.
88. The equilibration regulator of
claim 87
, wherein:
said digit line pair comprises:
a first digit line having a voltage potential,
a second digit line having a complementary voltage potential;
said digit line pair has a configured equilibrate voltage generally between said voltage potential and said complementary voltage potential;
said defect has a biasing effect on said first digit line, wherein said digit line pair approaches a deviant equilibrate voltage in response to said biasing effect of said defect; and
said defect compensation voltage has a counter-bias effect on said digit line pair, wherein said digit line pair approaches said configured equilibrate voltage in response to said counter-bias effect.
89. The equilibration regulator of
claim 88
, wherein:
said test voltage has a potential greater than VCC and less than said potential of said defect compensation voltage; and
said test voltage has an alternate counter-bias effect on said digit line pair, wherein said alternate counter-bias effect is less than said counter-bias effect of said defect compensation voltage.
90. The equilibration regulator of
claim 89
, wherein said defect compensation voltage is another test voltage.
91. The equilibration regulator of
claim 90
, wherein said defect is a conductive path coupling said cell plate generator circuit to said digit line pair.
92. A method of testing a memory array storing a test data pattern, comprising:
allowing a defect to hinder an ability to accurately read said test data pattern; and
preventing said memory array from restoring said ability to accurately read said test data pattern;
reading said test data pattern; and
checking said test data pattern for changes.
93. The method in
claim 92
, wherein preventing further comprises preventing said memory array from fully restoring said ability to accurately read said test data pattern.
94. The method in
claim 93
, further comprising:
changing said test data pattern;
writing said test data pattern to said memory array; and
repeating said allowing, preventing, reading, and checking.
95. A method of detecting any short from a digit line pair to ground, comprising:
initiating an equilibration charge of said digit line pair at a charge rate;
allowing any short to discharge said digit line pair toward ground at a discharge rate; and
decreasing said charge rate.
96. The method in
claim 95
further comprising:
writing an input data bit to a cell corresponding to said digit line pair;
reading an output data bit from said cell; and
comparing said input data bit with said output data bit.
97. The method in
claim 96
, wherein:
writing is performed before initiating; and
reading is performed after decreasing.
98. The method in
claim 97
, wherein writing further comprises writing a logic 0 to said cell.
99. The method in
claim 98
wherein:
initiating further comprises providing electrical communication between a cell plate generator and said digit line pair; and
decreasing further comprises:
regulating said electrical communication using a control device, and decreasing a driving signal of said control device.
100. A method of detecting a short from a digit line to a cell plate, comprising:
charging said digit line to a first potential, wherein said first potential represents a logic value;
charging a complementary digit line to a second potential representing a complementary logic value;
allowing said short to urge said digit line to a third potential;
providing electrical communication between said digit line and said complementary digit line;
allowing said digit line and said complementary digit line to establish an initial common voltage generally between said second potential and said third potential;
equilibrating said digit line and said complementary digit line toward a final common voltage generally between said first potential and said second potential, wherein said equilibrating occurs at an equilibration rate; and
limiting said equilibration rate.
101. The method in
claim 100
, wherein charging a digit line further comprises writing said logic value to a cell.
102. The method in
claim 101
, further comprising receiving a first signal, and wherein allowing said short to urge said digit line to a third potential is performed in response to receiving said first signal.
103. The method in
claim 102
, further comprising receiving a second signal, and wherein providing electrical communication is performed in response to receiving said second signal.
104. The method in
claim 103
, wherein:
said first signal has a first voltage level;
said second signal has a second voltage level; and
said second voltage level is higher than said first voltage level.
105. The method in
claim 104
, wherein:
said first signal has a test duration and a non-test duration; and
said test duration is longer than said non-test duration.
106. The method in
claim 105
, wherein said first signal and said second signal are RAS signals.
107. A method of preparing to test a storage capacitor of a memory cell, wherein said memory cell also includes an access transistor and a potential node providing an initial DVC2 voltage signal to said storage capacitor, and wherein said method comprises:
turning off said access transistor;
providing a forced voltage signal to said storage capacitor, wherein said forced voltage signal has a potential greater than a potential of said initial DVC2 voltage signal; and
preventing said forced voltage signal from substantially affecting any digit line associated with said memory cell.
108. The method in
claim 107
, wherein providing a forced voltage signal further comprises providing said forced voltage signal through said potential node.
109. The method in
claim 108
, further comprising:
providing a main digit line for said memory cell;
providing a complementary digit line for said main digit line; and
equilibrating said main digit line and said complementary digit line to an equilibration potential generally equal to said potential of said initial DVC2 voltage signal.
110. The method in
claim 109
, wherein said preventing further comprises restricting electrical communication of said forced voltage signal to said main digit line and said complementary digit line.
111. A method of detecting a leak within a storage capacitor of a memory cell affiliated with a digit line, wherein a cell plate generator is coupled to said storage capacitor and to said digit line, and said cell plate generator is further configured to generate a voltage, comprising:
charging said storage capacitor with an input potential corresponding to a logic 0 value;
initiating a static refresh pause;
increasing said voltage generated by said cell plate generator;
limiting electrical communication between said cell plate generator and said digit line;
discharging an output potential of said storage capacitor to said digit line; and
comparing said input potential with said output potential.
112. The method in
claim 111
, wherein initiating a static refresh pause further comprises preventing electrical communication between said digit line and said storage capacitor.
113. The method in
claim 112
, wherein initiating a static refresh pause further comprises allowing electrical communication between said digit line and said cell plate generator.
114. The method in
claim 113
, further comprising allowing said leak to change said input potential to said output potential, wherein said allowing occurs during said static refresh pause.
115. The method in
claim 114
, further comprising ending said static refresh pause before discharging.
116. A method of altering a margin between a potential of a charge stored in a memory cell and a common potential of a digit line pair, wherein said memory cell and said digit line pair are configured to receive an equilibrate signal, comprising:
allowing a defect to change said common potential of said digit line pair; and
substantially isolating said digit line pair from said equilibrate signal.
117. The method in
claim 116
, wherein said defect is a short from said digit line pair to ground.
118. A method of altering a margin between a potential of a charge stored in a memory cell and a potential of a shorted digit line pair, wherein said digit line pair is configured to receive an equilibrate signal, and wherein said digit line pair comprises a first digit line having a first initial potential and a second digit line having a second initial potential, comprising:
allowing a defect to change said first initial potential of said first digit line;
shorting said first digit line to said second digit line; and
substantially isolating said digit line pair from said equilibrate signal.
119. The method in
claim 118
, wherein:
said defect is a short between an equilibrate signal node and said first digit line; and
said equilibrate signal node is configured to transmit said equilibrate signal.
120. A method of altering a margin between a potential of a charge stored in a memory cell and a potential of a digit line, wherein said memory cell and said digit line are configured to receive an equilibrate signal having a voltage, comprising:
changing said voltage of said equilibrate signal;
allowing said potential of said charge in said memory cell to change due to a defect and said equilibrate signal; and
substantially isolating said digit line from said equilibrate signal.
121. The method in
claim 120
, wherein said defect is a short within a storage capacitor of said memory cell.
122. The method in
claim 121
, wherein said defect is a defect of a dielectric layer between two cell plates of said storage capacitor.
123. The method in
claim 122
, wherein said defect is a nitride defect.
124. A method of generally identifying a defect within a memory circuit, wherein said memory circuit is connected to a cell plate, comprising:
writing an input bit to said memory circuit;
subjecting said memory circuit to at least one test stage;
associating said defect with said test stage;
associating an isolation of said cell plate with said test stage;
initiating said isolation of said cell plate in relation to said test stage;
reading an output bit from said memory circuit; and
comparing said input bit with said output bit.
125. The method in
claim 124
, wherein:
said test stage comprises a precharge cycle; and
initiating further comprises initiating said isolation of said cell plate during said precharge cycle.
126. The method in
claim 124
, wherein:
said test stage comprises a long RAS low period; and
initiating further comprises initiating said isolation of said cell plate after said long RAS low period.
127. The method in
claim 124
, wherein:
said test stage comprises a static refresh pause; and
initiating further comprises initiating said isolation of said cell plate during said static refresh pause.
128. A method of altering the drive of a cell plate to digit line bleeder device, comprising:
entering a test mode;
providing a plurality of drive voltage sources for said bleeder device; and
applying said plurality of drive voltage sources to said bleeder device.
129. The method in
claim 128
, wherein said providing further comprises providing a plurality of discrete drive voltage sources for said bleeder device.
130. The method in
claim 128
, wherein said providing further comprises providing a range of drive voltage sources for said bleeder device.
131. A method of stressing a memory device having a digit line configured to selectively charge to a high potential a mid-level potential, and a low potential, comprising:
allowing a defect to alter said mid-level potential; and
slowing a restoration of said mid-level potential.
132. A method of testing a memory cell having an equilibrate voltage, a logic voltage corresponding to a logic value, and a margin representing the difference between said equilibrate voltage and said logic voltage, comprising:
enabling any defect of said memory cell to alter said margin; and
reading data from said memory cell.
133. The method in
claim 132
, further comprising exacerbating any margin alteration from said defect.
134. A method of compensating for a defect within a semiconductor device having a digit line configured to receive a cell plate signal from a signal node, comprising:
providing a signal regulator between said signal node and said digit line;
applying a defect-countering driving voltage to said signal regulator.
135. The method in
claim 134
, wherein:
said defect comprises an unregulated electrical communication of said cell plate signal to said digit line; and
said defect-countering driving voltage is higher than a generally standard driving voltage for said signal regulator.
136. The method in
claim 135
, further comprising:
testing for said defect, wherein said testing further comprises:
providing said signal regulator with a plurality of driving voltages; and
selectively applying said plurality of driving voltages.
137. The method in
claim 136
, wherein said applying further comprises applying said defect-countering driving voltage in response to finding said defect.
138. The method in
claim 137
, wherein:
said method further comprises initiating a non-test mode after testing for said defect; and
said applying further comprises applying said defect-countering driving voltage during said non-test mode.
Description
    TECHNICAL FIELD
  • [0001]
    The present invention relates generally to semiconductor circuit devices and, more specifically, to a circuit for changing the voltage applied to selective portions of a memory array. Such portions include digit line pairs as well as the gate of a transistor used to regulate sense amplifiers.
  • BACKGROUND OF THE INVENTION
  • [0002]
    In the operation of certain semiconductor circuit devices, pullup and pulldown sense amplifiers (sense amps) detect and amplify a small charge stored within a memory cell. In general, two complementary digit lines are attached to a pullup sense amp and a pull down sense amp. At the beginning of a reading operation, both lines are at an equilibrate voltage Veq, which is generally between the potential of a voltage source used to operate the semiconductor device (VCC) and ground potential (0 volts). While Veq is changeable either intentionally or inadvertently through a defect, Veq is ideally equal to VCC/2 during non-test operations. This midpoint voltage is defined as DVC2.
  • [0003]
    One of the digit lines is coupled to a memory cell. The reading process involves a discharge from the memory cell to the corresponding digit line, which creates a slight difference in voltage between the two digit lines. This difference is then amplified by the sense amps: the digit line with the slightly lower voltage has its voltage further decreased by the pulldown sense amp, and the voltage of the other digit line is increased by the pullup sense amp. Once the voltage difference has been amplified, the digit lines can then be used to operate less sensitive circuitry.
  • [0004]
    Between reading cycles, it is necessary to return the complementary digit lines to Veq. This occurs during what is known as a precharge cycle, wherein equilibration transistors short the complementary digit lines together. Further, a signal having a potential of DVC2 is communicated from a DVC2 voltage generator to the shorted digit lines through a bleeder device.
  • [0005]
    Concerning the operation of the sense amps, it should be noted that pulling down the voltage of a digit line involves coupling the line to ground through a pulldown transistor. Because an entire row of digit line pairs often connects to the same pulldown transistor through a common node, the pulldown transistor will most likely have to draw current from one line of each of several pairs. In doing so, there is a risk that the transistor will become saturated with current and therefore become slower in pulling down the voltage of additional digit lines. This may lead to errors in reading, especially if an entire row of memory cells is storing logic 1's except for one cell storing a logic 0; for once the logic 0 is discharged, a slow pulldown may result in an improper reading of that logic 0 value.
  • [0006]
    One known way to solve this problem is to include an optional active area in the gate of the pulldown transistor. The increased size of the gate raises the threshold at which the pulldown transistor becomes saturated. However, one of ordinary skill in the art will appreciate that this solution requires a costly metal mask change. Further, any attempt to speed up the slowed pulldown raises other problems in reading, as disclosed in U.S. Pat. No. 5,042,011, by Casper, et al. The Casper '011 reference discloses that pulling down the common node too quickly may result in capacitive coupling between the sources and drains of the sense amp's transistors. During capacitive coupling, both digit lines in one sense amp are pulled down before the common node is pulled down low enough to turn on one of the sense amp transistors. When the sense amp finally turns on, it shorts out the capacitive coupling, bouncing the digit lines and, in the process, creates line noise that will interfere with the ability to read the data properly.
  • [0007]
    Early saturation and capacitive coupling could be avoided if one knew the margin— the difference in voltage between a logic 0 signal and a logic 1 signal—that the pulldown transistor was capable of accommodating. The only way to do so, as taught by the prior art, is to separate the pulldown transistor with a laser and probe the gate.
  • [0008]
    As an alternative to determining the sense amp's margin, one could simply test the sense amp's ability to operate at the given source voltage used in non-test operations. Prior art suggests entering a series of test data patterns into memory. Logic 1's are written to the cells of each memory array, with the exception of one column of logic 0's. As a result, each row contains only one cell storing a logic 0, thereby creating the most likely circumstance for an error in reading the data. The data in the array is then read and checked for errors. Once the first group of test data has been processed, a second sample of test data is entered with the logic 0's written to the next column. This process repeats until a logic 0 has been written to and read from every cell in any given row in the memory array. The results will indicate the pulldown transistor's ability to read data accurately. The problem with this process, however, is that it is time consuming to enter multiple samples of test data.
  • [0009]
    Thus, there is a need in the art for a quicker circuit and method for testing the capabilities of a sense amp. Further benefit would be derived if this test could indicate the margin of the sense amp's pulldown transistor.
  • [0010]
    In addition to inadequate pulldown transistors, other problems, such as defects arising during the processing of semiconductor devices, may contribute to reading errors. Various techniques involving equilibration of the complementary digit lines can be used during testing to detect these problems. For example, occasionally a digit line will inadvertently have a short to ground. As a result, the potential of that digit line will leak towards 0 volts. To detect this problem, prior art teaches extending the time for the precharge cycle during a test mode. If the short has a low enough resistance, the short will overcome the charging ability of the DVC2 voltage generator, which remains coupled to the digit lines, and Veq of the digit lines will decrease. Thus, a longer precharge cycle allows Veq to lower even further. As a result, line noise is more likely to register as a logic 0 discharge on the digit line when in fact the storage cell contains a logic 1 and has not yet discharged. Alternatively, assuming that a logic 1 is properly discharged and sensed, a reading error is still likely: Veq may be so low due to the short that the pullup sense amp may not be able to pull up the digit line's voltage in time to register as a logic 1 for purposes of driving external circuitry. Increasing the likelihood of error is desirable in the test mode, as it helps to identify errors that would affect non-test operations. Further, a reading error occurring after this extended precharge cycle will indicate the nature of the defect—in this case a short in at least one of the digit lines. However, this testing process can be time consuming. As an example, a 64 meg DRAM having a 16 meg×4 configuration requires approximately 170 seconds to carry out this test. It would be a benefit to the art to have a faster way to test for this problem.
  • [0011]
    A second problem that could be detected by altering the equilibration rate of the digit lines involves a short between the cell plate and the digit line. The typical technique for discovering this problem is to initiate a long RAS (Row Address Strobe) low signal. During the low RAS, the digit lines are not equilibrated. Rather, they are charged to their complementary voltage levels. Ideally, once the low RAS ends and the lines are shorted, both digit lines should approach a Veq level of DVC2. However, a short between one of the digit lines and the cell plate will allow the DVC2 generator 68 to change that digit line's voltage during the RAS low period. Thus, once the lines are shorted, their respective voltages will meet at a different Veq level. This will affect the margin between Veq and the voltage corresponding to one of the logic values and thereby increase the likelihood of a reading error. Eventually, the signal from the DVC2 voltage generator will restore the proper equilibrate voltage once the RAS low signal ends. Nevertheless, for purposes of detecting this problem before non-test operations begin, it would be desirable to slow the restoration of the proper Veq level.
  • [0012]
    A third example concerns a defect that could exist within the memory cell's storage capacitor, such as a defect in a nitride layer acting as a dielectric between the memory cell's conductive plates. Such a defect could cause a short within the storage capacitor. Because the storage capacitors are coupled to the DVC2 voltage generator, a defective capacitor “storing” a 0 volt charge, representing a logic 0, will slowly charge to the DVC2 level. The closer the storage capacitor approaches a DVC2 charge, the more likely that a logic 1 value may be misread during the next reading. One way to detect this problem in the prior art is to initiate a static refresh pause, wherein the memory cell's access transistor remains deactivated for a longer time than usual—generally 100 milliseconds. As a result, the capacitor, which should be storing a logic 0, has a longer time to charge to a higher voltage, thereby making an error in the next reading cycle more likely.
  • [0013]
    Once again, a speedier test is desired. The defect might be detected earlier if the problem were exacerbated to the point where the leaked charge for the stored logic 0 exceeded the equilibrate charge of the digit lines. As a result, a logic 1 would be read from the cell even though it was known that a logic 0 had been written. One could speed up the leakage into the storage capacitor by forcing DVC2 to a higher voltage. However, the equilibrate voltage of the digit lines would also increase accordingly and remain higher than the voltage of the charge in the storage capacitor. Thus, forcing DVC2 would not appreciably increase the ability to detect an error unless the equilibration of the digit lines could be slowed. The only way to do this in the prior art is through the use of a costly metal option to change the gate voltage of the bleeder device.
  • SUMMARY OF THE INVENTION
  • [0014]
    Given the need for regulating the drive of a sense amp, as well as the need for regulating the equilibration signal from a DVC2 voltage generator, a test circuit is provided for varying the voltage of a signal used to drive a connection device that allows electrical communication within a semiconductor circuit. One preferred circuit embodiment includes a contact pad for carrying a range of test voltage signals to said connection device. In another preferred circuit embodiment, a regulator circuit enables a series of discrete voltages to drive the connection device.
  • [0015]
    In one set of applications involving the regulation of a sense amp, the connection device comprises a sense amp's voltage pulling transistor. Any circuit embodiment covered by the present invention can be used to test drive the transistor. In a preferred method of use, a test data pattern is entered and the data is read several times, with a different voltage driving the sense amp's pulldown transistor each time. One advantage of this preferred method is that it reduces the need for entering several elaborate test data patterns and, therefore, allows for quicker testing of memory arrays. A second advantage is that the embodied method and devices allow a determination of the lowest supply voltage that can be used during normal operation without errors in reading data. Yet another advantage is the ability to determine the highest supply voltage, and therefore the fastest reading speed, that can be used during normal operations without causing capacitive coupling. In doing so, the preferred circuit embodiments and method increase the sense amp's ability to distinguish between a logic 0 voltage and a logic 1 voltage without physically altering the sense amp. Further, in the process of determining the lowest and highest voltages at which the sense amp is capable of functioning, the preferred embodiments and method also provide a way to ascertain the margin without dissecting components of the sense amp.
  • [0016]
    Concerning the specific errors that may be detected in relation to equilibrating the digit lines, the connection device comprises an isolation bleeder device coupled between the DVC2 voltage generator and a digit line pair. The circuit embodiments provide a test mode apparatus for driving the bleeder device in order to slow or quicken the equilibration of the digit line pair. Applying these embodiments provides the advantage of a quicker detection of defects such as a short from a digit line to ground, a short from a digit line to a cell plate, and a short within the storage capacitor of a memory cell. The embodiments also provide an alternative advantage of overcoming the influence of these defects during non-test modes.
  • BRIEF DESCRIPTION OF THE DRAWINGS
  • [0017]
    [0017]FIG. 1 depicts a row of n-channel pulldown sense amps with associated D, D*, and WL lines; a pullup sense amp; and a series of memory cells, as found in the prior art. FIG. 1 also shows a digit line equilibration circuit as found in the prior art.
  • [0018]
    [0018]FIG. 2 is a graph indicating the voltage of the conductive paths D and D* over time in the event that a memory cell storing a logic 0 discharges to D. FIG. 2 also demonstrates the resulting amplification of the difference in voltage.
  • [0019]
    [0019]FIG. 3 is a graph demonstrating the relationship between drive current (IDV) and the gate-source voltage of a pulldown transistor (VGS) at various levels of voltage applied to the gate (VGATE).
  • [0020]
    [0020]FIG. 4 details one exemplary circuit embodiment in accordance with the present invention as used with a sense amp.
  • [0021]
    [0021]FIG. 5 illustrates a second exemplary circuit embodiment in accordance with the present invention as used with a sense amp.
  • [0022]
    [0022]FIG. 6 shows a third exemplary circuit embodiment in accordance with the present invention as used with a sense amp.
  • [0023]
    [0023]FIG. 7a is a schematic of a portion of a memory array depicting an embodiment of the current invention as used in the digit line/cell plate region of a memory array. FIG. 7a further depicts a first type of possible defect within said memory array.
  • [0024]
    [0024]FIG. 7b is a graph illustrating the effect of the first defect and a first embodied method of the current invention.
  • [0025]
    [0025]FIG. 7c is another graph illustrating the effect of the first defect and the first embodied method of the current invention.
  • [0026]
    [0026]FIG. 8a depicts a cross-section of a portion of a memory array including a second type of defect.
  • [0027]
    [0027]FIG. 8b demonstrates the effect on a memory array of the second type of defect as well as the effect of a second embodied method of the current invention.
  • [0028]
    [0028]FIG. 8c further demonstrates the effect on a memory array of the second type of defect as well as the effect of a third embodied method of the current invention.
  • [0029]
    [0029]FIG. 8d depicts the effect of a fourth embodied method of the current invention as it relates to the second type of defect.
  • [0030]
    [0030]FIG. 9a is a schematic of a portion of a memory array depicting a third type of defect in said memory array.
  • [0031]
    [0031]FIG. 9b is a graph indicating the effect of the third type of defect.
  • [0032]
    [0032]FIG. 9c is a graph illustrating a method in the prior art for detecting the third type of defect.
  • [0033]
    [0033]FIG. 9d is a graph illustrating the effect of a fifth embodied method of the current invention.
  • DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
  • [0034]
    [0034]FIG. 1 illustrates the general configuration of sense amps in a memory array. A pulldown sense amp 20 includes cross coupled n-channel transistors Q1 and Q2, as well as a pulldown transistor Q3, which is an n-channel transistor driven by a signal designated as LENSA. These elements play a part in sensing and amplifying a voltage difference between D and D* caused by shorting a memory cell 22 to D by way of access transistor Q4. The sources of Q1 and Q2 are connected to a common pulldown node 24, and the gate of each is connected to the other's drain. The gate of Q1 also connects to the line D*, whereas the gate of Q2 connects to the line D.
  • [0035]
    As discussed above, each line D and its corresponding line D* are initially at the same voltage DVC2. For purposes of explanation, DVC2 is assumed to be 1.65 volts, or one half of the source voltage VCC, which is 3.3 volts. Lines D and D* connect to opposite sides of each sense amp 20. Common pulldown nodes 24 found in the sense amp arrays will also be at DVC2. A signal sent through the path WL will cause a storage capacitor 150 of particular memory cell 22 to discharge to a line D, thereby slightly changing D's voltage while the voltage of D* remains at DVC2. Again, for purposes of explanation, a memory cell discharge will be assumed to cause a 0.2 volt difference in D. The pulldown sense amp 20 will then turn on when the common pulldown node 24 is one transistor threshold voltage below D or D*, whichever is highest. For instance, if a memory cell 22 is storing a logic 1, a discharge to D will increase D's voltage to 1.85 volts. As a result, the pulldown sense amp transistor gated by D (Q2) turns on faster than the one gated by D* (Q1). With transistor Q2 on, D*'s voltage is pulled down from 1.65 volts towards ground as the common pulldown node 24 is pulled down as well. Further, the lowering voltage of D* serves to turn on the pullup sense amp transistor gated by D* (Q14) before the other pullup sense amp transistor turns on. The voltage supply VCC then charges line D.
  • [0036]
    On the other hand, if the memory cell 22 had been storing a logic 0, then a discharge to D would slightly lower D's voltage to 1.45 volts. The pulldown sense amp transistor gated by D* (Q1) would turn on first and D's voltage would be further decreased toward ground by the pulldown sense amp, thereby allowing the pullup sense amp to increase D*'s voltage toward VCC. In this way, a small voltage difference between D and D* is sensed and amplified. Once the voltage difference has been amplified, D and D* can drive less sensitive circuitry not shown in FIG. 1. It should be noted that, if a logic 0 is transmitted to D, then the pulldown sense amp need only pull down D from 1.45 volts. If a logic 1 is transmitted to D, then the pulldown sense amp must pull D* from the higher DVC2 level—1.65 volts.
  • [0037]
    Therefore, if many logic 1's in a memory array row are read, the extra voltage that must be pulled contributes to saturating the pulldown transistor Q3 with drive current, thereby slowing any further pulldown. The problem created by slow pulldown is illustrated in FIG. 2, where slope X denotes the initial discharge to D from a memory cell 22 storing a logic 0. FIG. 2 further illustrates the amplification of the difference in voltage between D and D*. Slope Y denotes the time required for D to drop in voltage given a situation where a row of cells contains a roughly equal number of logic 1's and logic 0's. Should there be many logic 1's read amongst a single logic 0, then the outcome changes: as the logic 0 is read, the pulldown transistor Q3, having approached saturation, takes much longer to pull down D's voltage. This result is illustrated by slope Z. Other circuitry elements (not shown) that are driven by D may read D before its transition to a lower voltage has been completed. As a result, a logic 0 value may be misread as a logic 1.
  • [0038]
    As illustrated in FIG. 3, increasing the voltage to the gate of the pulldown transistor allows the transistor to pulldown more current before saturation. One preferred embodiment of the current invention that uses this principal is detailed in FIG. 4, where the pulldown transistor Q3 is driven by a test circuit 26 through an inverter 27. In this embodiment, the inverter 27 comprises a p-channel transistor Q6 and an n-channel transistor Q8. The coupled gates of inverter transistors Q6 and Q8 form an input node 28 for receiving a signal ENSA*, which may be VCC, ground, or a signal from another driver. The coupled drains of the inverter transistors Q6 and Q8 output the LENSA signal that drives the pulldown transistor Q3. The source of Q8 is coupled to ground. The source of Q6 is coupled to a source node 30 that branches into a first conducting path 32 and a second conducting path 34. The first conducting path 32 is coupled to an n-channel transistor Q10, which has a channel width-to-length ratio of around 500/2. The drain of transistor Q10 is coupled to a contact pad 36. It should be understood that the term “contact pad” includes any conductive surface configured to permit electrical communication with a circuit or a node. The gate of transistor Q10 is coupled to an inverter 60 through another n-channel transistor Q36. Together, inverter 60 and transistor Q36 comprise a latch device, and both are coupled to VCCP. Further, inverter 60 receives a TEST* signal as an input. In addition, the gate of transistor Q10 is also coupled to a feedback capacitor 62. This feedback capacitor 62 comprises an n-channel transistor having a size of approximately 100/100, wherein the drain and source are shorted and coupled to the first conductive path 32. The second conducting path 34 is coupled to a p-channel transistor Q12, driven by a signal TEST, which is understood to be the complement of TEST*. The transistor Q12 is also coupled to VCC, although no voltage source is considered to be a part of the invention.
  • [0039]
    During testing, TEST* transmits a low voltage signal which is received by the inverter 60. In response, the inverter 60 initiates a VCCP signal, sending it through transistor Q36 which outputs the VCCP signal to the gate of transistor Q10, thereby switching on Q10. The feedback capacitor 62 serves to maintain and replenish this VCCP signal in the event of leakage. Capacitive coupling between the gate and drain of transistor Q10 allows Q10 to carry signals having a range of voltages for modifying the drive of the pulldown transistor Q3. Simultaneously, the TEST signal, applying a high voltage to transistor Q12, isolates VCC. A test data pattern is entered into the memory cells 22 and read with varying voltages driving the pulldown transistor Q3. The data read at various alternate voltages sent through bond pad 36 can be compared with the data as originally written. This series of readings indicates the range of voltages through which the pulldown transistor Q3 is capable of allowing accurate data readings. Once testing has ended, TEST* sends a high voltage signal and TEST becomes low, thereby isolating the bond pad and allowing the VCC signal to transmit to the pulldown transistor Q3.
  • [0040]
    The embodiment illustrated in FIG. 5 is a package part of the semiconductor circuit device and receives a plurality of voltage sources with different magnitudes. The test circuit 26 allows selection among these sources for driving the gate of the pulldown transistor Q3. The inverter 27 is the same as in FIG. 4. In this exemplary embodiment, however, source node 30 is coupled to three discrete voltage sources. First, source node 30 is coupled to VCCP through a p-channel transistor Q20 that is driven by a low signal A*. Source node 30 is also coupled to DVC2 through another p-channel transistor Q22 that is driven by a low signal B*. Finally, source node 30 is coupled to VCC by way of a p-channel transistor Q24. This p-channel transistor Q24 is gated by the output of a logic unit, such as a NAND gate 46, which will drive transistor Q24 in response to receiving a high signal A as a first input and a high signal B as a second input. Given the input vector scheme of this embodiment, one of the transistors Q20, Q22, or Q24 will be operable to the exclusion of the other two.
  • [0041]
    Thus, a low signal A* will drive the p-channel transistor Q20, thereby allowing VCCP to drive the pulldown transistor Q3. Simultaneously, signal B will be high, turning off p-channel transistor Q22. Further, the NAND gate output will also be high and turn off p-channel transistor Q24. If, on the other hand, signal B is low and signal A is high, then only p-channel transistor Q22 will be on, allowing DVC2 to transmit to the pulldown transistor Q3. Only when both signals A and B are high does the NAND gate 46 output a low signal and allow VCC drive the pulldown transistor Q3. The data read at these three voltage levels can then be compared with the data as originally written. It should be noted that this configuration does not require the die space needed for the contact pad 36.
  • [0042]
    Another embodiment concerns varying the voltage applied to a pullup sense amp 40. As seen in FIG. 1, the pullup sense amp 40 includes cross coupled p-channel transistors Q14 and Q16 as well as a pullup transistor Q18. As one of ordinary skill in the art understands, there is generally a pullup sense amp 40 corresponding to every pulldown sense amp. Nevertheless, for purposes of clarity, only one pullup sense amp 40 is shown. The sources of Q14 and Q16 are connected to a common pullup node 42, and the gate of each is connected to the other's drain. Further, the gate of Q14 connects to line D*, and the gate of Q16 connects to line D. Common pullup node 42 is coupled with pullup transistor Q18, which is another p-channel transistor. Pullup transistor Q18 is also coupled to the voltage source VCC. The pullup transistor Q18 is driven by a signal LEPSA*. FIG. 6 illustrates that the voltage driving pullup transistor Q18 may also be varied through the use of a test circuit 26 analogous to that used with the pulldown transistor Q3 in FIG. 5. FIG. 6 depicts an inverter 27 comprising a p-channel transistor Q26 and an n-channel transistor Q28. The coupled gates of inverter transistors Q26 and Q28 form an input pathway 48 for a control signal designated EPSA. The coupled drains transmit the inverted output signal EPSA* which, in turn, is received by a prior art device 50 that outputs the LEPSA* signal used to drive the pullup transistor Q18. The source of Q26 is coupled to VCC, whereas the source of Q28 is coupled to the test circuit 26 which, in this embodiment, includes three conductive paths. The first path 52 leads to DVC2 by way of an n-channel transistor Q30, which is driven by a signal C. The second path 54 is coupled to a voltage source VBB through an n-channel transistor Q32, as driven by a signal D. The third path 56 leads to ground by way of n-channel transistor Q34. The gate of n-channel transistor Q34 is coupled to the output of a NOR gate 58. The NOR gate 58 accepts signal C as a first input and signal D as a second input and will activate transistor Q34 only when both signals are low. Further, this embodiment is configured in a manner analogous to the embodiment in FIG. 5, in that signals C and D will never simultaneously activate their respective transistors Q30 and Q32.
  • [0043]
    The three n-channel transistors Q30, Q32, and Q34 will turn on if a high, or logic 1, signal is transmitted to their respective gates. As with the embodiment shown in FIG. 5 for the pulldown sense amp, the signals and transistors are configured to allow only selective communication between one voltage source and the pullup transistor Q18. As a result, if signal C is high, it will latch the n-channel transistor Q30 and provide electrical communication between DVC2 and the pullup transistor Q18. At the same time, the low signal from D turns off n-channel transistor Q32. Under these circumstances, the signals C and D also result in a low signal output from the NOR gate 58, thereby turning off n-channel transistor Q34. Thus, all of the other voltage sources are isolated. Similarly, if signal D is high, then only n-channel transistor Q32 is turned on and VBB electrically communicates with pullup transistor Q18. When both signals are low, the NOR gate 58 outputs a high signal, thereby grounding the source of the n-channel inverter transistor Q28. This embodiment has benefits similar to the embodiment in FIG. 5.
  • [0044]
    Returning to FIG. 1, a prior art equilibration circuit can be seen as part of the memory device. For purposes of explaining the following embodiments of this invention, VCC is now presumed to be 5 volts. A transistor Q101 is coupled between digit line D and its complementary digit line D*. The transistor is driven by an equilibration signal EQ. It should be noted that the signal EQ results from a logic function and is distinguishable from the equilibrate voltage Veq, which represents the common mid-range voltage level of the complementary digit lines before a reading operation.
  • [0045]
    The signal EQ also drives two additional transistors Q102 and Q103, which are connected together in series at a node 120. These connected transistors Q102 and Q103 are also coupled between lines D and D*. Moreover, node 120 is coupled to a cell plate 138 and a DVC2 voltage generator 68 through a bleeder device 122. The DVC2 voltage generator 68 transmits a cell plate signal CP of voltage DVC2 to the node 120. For purposes of explaining the following embodiments of this invention, DVC2 is now 2.5 volts. The bleeder device 122 is driven by a signal of voltage VCCP, wherein VCCP results from having pumped VCC to an even higher potential.
  • [0046]
    At the beginning of a precharge cycle, digit line D and its complementary digit line D* are at different voltages as a result of a discharge of the memory cell 22 during the reading cycle. One line will have a charge equal to the VCC value of 5 volts, while the other line will have a 0 volt charge. The equilibrate signal EQ is then sent, activating transistor Q101, which shorts D and D* together. Moreover, the signal EQ activates transistors Q102 and Q103, which not only provide another short between D and D* but also allow the CP signal to be communicated to those lines. As a result, the lines D and D* equilibrate, both gaining a charge of potential DVC2 (2.5 volts), which is the desired equilibrate voltage Veq in this example. Once the lines are equilibrated, they are ready for further testing.
  • [0047]
    For various reasons, a particular portion of the memory array may be defective. Hopefully, testing processes will identify those defects. As discussed above and illustrated in FIG. 7a, a first defect 124 that may exist as a short to ground of the digit line D. FIG. 7b illustrates the effect of the first defect 124. During the precharge cycle, the CP signal is trying to charge the digit lines D and D* to the 2.5 volt DVC2 level and maintain that level. However, if the resistance of the short is not too great, the first defect 124 may cause the digit lines to discharge toward ground faster than CP can charge them to 2.5 volts. As a result, once the precharge process has ended at time t1, the digit lines may be equilibrated at a potential lower than 2.5 volts, such as 1.7 volts. Having a Veq at a level other than DVC2 makes the memory array susceptible to reading errors. For example, in the present situation illustrated in FIG. 7b, where Veq is too low, line noise on D occurring at time t2 is more likely to register as a logic 0 discharge when in fact the storage cell 150 contains a logic 1 and has not yet discharged. Alternatively, assuming that a logic 1 is properly discharged and sensed at time t2′, a reading error is still likely: as seen in FIG. 7c, Veq may be so low due to the short that the pullup sense amp may not be able to sufficiently pull up the digit line's voltage by the time t3, when external circuitry accesses line D. In order to find such a reading error, prior art requires an extended precharge time, up to time t1, in order to allow the discharge from the first defect 124 to overtake the charge from CP.
  • [0048]
    The current invention, however, provides an alternative to requiring a long precharge time. FIG. 7a illustrates that the VCCP signal driving the bleeder device has been replaced with the test circuit 26 that applies a different voltage VREG to regulate the bleeder device. In the case of the first defect 124, the test circuit 26 transmits a signal having a voltage lower than VCCP to drive the bleeder device 122. This causes a slower charge rate and allows the discharge from the first defect 124 to quickly overtake the charging from CP, as seen by the dashed lines in FIGS. 7b and 7 c. With the resulting increased disparity between the charge rate and the discharge rate, the precharge period need only endure until time t1′ in order to increase the likelihood of detecting an error.
  • [0049]
    The design of test circuit 26 can be the same as those used in FIGS. 4 and 5, wherein a source node 30 has access to at least one test voltage, either through a bond pad 36 or from a discrete voltage source. In this application, however, the source node 30 is coupled to the bleeder device 122. Furthermore, VCCP is the voltage used in non-test operations to drive the bleeder device, and VCC and DVC2 are used to slow the charge rate. It should be further understood that the number of voltage options could be increased. Alternatively, the number of voltage options could be decreased to offer only one test voltage and one non-test voltage.
  • [0050]
    These circuit embodiments, as well as others falling under the scope of the invention, have uses in detecting other defects. FIG. 8a illustrates another defect 136 that might occur within a memory array. The cross-sectional view in FIG. 8a shows the cell plate 138 coupled to a first n-region 140 of access transistor Q4. Ideally, the only way for the DVC2 voltage generator 68 to charge the digit line D through the cell plate 138 is to drive the gate 142 of transistor Q4 so that the charge may pass from the first n-region 140 to a second n-region 144. From there, the charge travels through a tungsten plug 146, which serves as a contact between the second n-region 144 and the digit line D. Occasionally, however, a second defect 136 in the memory array may occur in the form of a short between the cell plate 138 and the tungsten plug 146. As discussed above, a long RAS low signal is used to detect this second defect 136. Assuming line D is charged to 0 volts, FIG. 8b shows that the long RAS signal allows line D to be charged to a higher voltage. Thus, when the low RAS signal ends at time t1 and the digit lines are shorted to begin equilibration, the digit lines will no longer have an initial tendency to reach an average potential between 5 and 0 volts (2.5 volts). Rather, because line D is now higher than 0 volts, the shorted lines will settle at a higher midpoint, such as 3.5 volts. At this point, the margin between the new equilibrate voltage and the voltage representing a logic 1 has decreased. Thus, an erroneous reading is more likely, as discussed above.
  • [0051]
    Conversely, if line D is initially charged to VCC (FIG. 8c), the short to the cell plate will cause D's voltage to lower during a long RAS low period. The resulting equilibrate voltage of lines D and D* could be lower than the preferred 2.5 volts. The lower equilibrate would again make an error in reading more likely. In either case, the CP signal will restore the equilibrate voltage to 2.5 volts by time t2. However, by decreasing the drive to the bleeder device 122, any of the embodiments of the current invention will serve to slow down the restoration of Veq to DVC2. With restoration time extended to time t2′, any circuit embodiment of the current invention increases the likelihood of detecting errors that would suggest the existence of the second defect 136. Alternatively, FIG. 8d shows that a circuit embodiment of the current invention could be used during a non-test mode to compensate for the second defect 136 by driving the isolation device 122 at a higher-than-normal level. As discussed above, the bleeder device 122 is normally driven at VCCP, a voltage level representing one or two Vt's above VCC. The potential Vt, in turn, is the threshold voltage of the bleeder device 122. A further increase in the potential of VCCP would allow the bleeder device 122 to quickly restore Veq to 2.5 volts by time t2″. The shorter restoration period reduces the chances of an erroneous reading.
  • [0052]
    [0052]FIG. 9a demonstrates yet another instance wherein the current invention could shorten test time. This instance concerns a third defect 148 comprising a short that may be caused by a nitride defect within the storage capacitor 150 of a memory cell 22. It should also be noted that one of the plates of the storage capacitor 150 is in fact the cell plate 138 and is therefore connected to the DVC2 generator. Given this third defect 148, FIG. 9b indicates that the CP signal, having a potential of DVC2, will charge the storage capacitor 150 toward that potential even though a logic 0 has been written to that cell for test purposes. During a static refresh pause, the word line WL leading to the memory cell 22 will continuously transmit a low signal, which turns off access transistor Q4 of the memory cell 22 and allows the storage capacitor 150 to take on a greater charge. With the stored charge having a higher voltage, such as 2 volts, it is more likely that the logic 0 will be misread at line D as a logic 1. In order to speed up the leakage into the storage capacitor 150, DVC2 is forced to a voltage higher than the normal 2.5 volts. Unfortunately, this would not result in much benefit under the prior art, as demonstrated by FIG. 9c: because the CP signal has a voltage of DVC2 and is in communication with D and D* during the static refresh pause, the CP signal would also charge lines D and D* to a higher voltage. With the circuit embodiments of the present invention, however, a lower voltage could be used to drive the bleeder device 122 and thereby slow the charging of the digit lines, as illustrated in FIG. 9d. Thus, while D and D* are regulated to substantially remain at 2.5 volts despite the forced DVC2 voltage, the storage capacitor may be quickly charged to a higher potential, such as 2.7 volts, which exceeds the equilibrate voltage and makes it very likely that a logic 1 will be mistakenly recognized.
  • [0053]
    One of ordinary skill can appreciate that, although specific embodiments of this invention have been described for purposes of illustration, various modifications can be made without departing from the spirit and scope of the invention. Concerning the invention as used with a sense amp, for example, a test circuit for the pullup sense amp could be configured to transmit an entire range of voltages through a contact pad, as done with the pulldown sense amp depicted in FIG. 4. In addition, the test circuit 26 in FIG. 6 could be used with a pulldown sense amp. Conversely, the test circuit 26 in FIG. 5 could be used with a pullup sense amp. Moreover, both of these test circuits could be coupled to the same inverter and used to test drive either type of sense amp.
  • [0054]
    Further, regarding the embodiments use with a cell plate, it should be noted that the embodiments may be applied for other testing. Any circuit embodiment, for instance, may be used during the precharge cycle discussed above in order to detect a short between a row line and a column line. Moreover, a circuit embodiment of the current invention could also be used during a non-test mode to overcome other defects in addition to the short between a digit line and cell plate, as described above.
  • [0055]
    It should also be noted that, given a particular voltage source used in an embodiment, that source can be independent of VCC rather than a mere alteration of VCC, such as VCCP or DVC2. Accordingly, the invention is not limited except as stated in the claims.
Classifications
U.S. Classification365/201, 365/189.11
International ClassificationG11C29/44, G11C7/00, G11C29/12, G11C29/56, G11C29/50, G11C29/02, G11C29/48
Cooperative ClassificationG11C2029/5004, G11C29/12, G11C29/50, G11C29/48, G11C29/025, G11C29/028, G11C29/02, G11C29/026, G11C29/56, G11C29/44, G11C29/50012, G11C11/401
European ClassificationG11C29/02F, G11C29/02E, G11C29/50C, G11C29/02H, G11C29/12, G11C29/44, G11C29/02, G11C29/50, G11C29/48, G11C29/56
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Effective date: 20140903