The present invention relates generally to voltage shifters, and more specifically to dual-level voltage shifters.
One of the most challenging problems confronting the designer in deep submicron circuits in today's technological society is leakage power which has increasingly become accountable for a significant portion of the total power consumption of such devices. Recent circuit techniques to reduce the leakage power in circuits have employed higher supply voltage to the circuit in an attempt to drive a leakage control device. One common type of leakage reduction technique comprises a sleep transistor whose gate node is driven by the additional supply voltages. A second type of leakage reduction technique changes the application circuit's body bias using the additional supply voltages. To switch between the standby and active modes, the sleep transistor's gate bias in the case of the first type of leakage reduction or the body bias of the circuit in the case of the second type of leakage reduction need to be switched between different voltage levels. This is accomplished through the use of a voltage shifter. Traditional voltage shifters, however, typically flow a significant amount of static current or support only single level voltage shifting.
For example, FIG. 1 shows a voltage shifter 100 with static current flowing. Voltage shifter 100 comprises a series connection of p-type transistor 102 and n-type transistor 104 in series between voltage Vcc and ground. The gates of transistors 102 and 104 are connected to each other. Transistors 106 and 108 are connected in series between Vpp and Vnn, where Vpp=Vcc+ΔV and Vnn=ΔV. The gates of transistors 106 and 108 connected to each other and to a node 110 between transistors 102 and 104. An input voltage Vin is supplied to the gates of transistors 102 and 104, and an output is generated from between transistors 106 and 108. Vin varies from a logical one (Vcc) to a logical zero (0 V). Vout varies from Vcc+ΔV to Vcc−ΔV.
When Vin is set to Vcc, transistor 104 is on and transistor 102 is off, but the ground voltage at node 110 is insufficient to completely shut off transistor 106, and a static current flows in path 112. When Vin is set to 0 V, transistor 104 is off and transistor 102 is on, but the voltage Vcc at node 110 is insufficient to completely shut off transistor 108, and a static current flows in path 114. Shifter 100 cannot be used for low power circuits.
Another example of a prior art voltage shifter 200 is shown in FIG. 2. Voltage shifter 200 comprises p-type transistor 202 and n-type transistor 204 connected in series between voltage Vpp and Vnn, which is ground, and p-type transistor 206 and n-type transistor 208 also connected in series between Vpp and Vnn. (ground). An input voltage Vin is connected to the gate of transistor 204, and through inverter 210 to the gate of transistor 208. The gate of transistor 202 is connected at an output node defined between transistors 206 and 208. The gate of transistor 206 is connected between transistors 202 and 204.
In this configuration, when Vin is equal to Vcc, transistors 204 and 206 are on, and transistors 202 and 208 are off, and Vout is pulled to Vpp, which equals Vcc+ΔV, through transistor 206. When Vin is set to 0 V, transistors 202 and 208 turn on, and transistors 204 and 206 turn off. In this instance, Vout is pulled to Vnn (ground) through transistor 208. In this single level voltage shifter, no static current flows. However, the circuit 200 is only a single level voltage shifter. If Vnn is set to −ΔV to achieve dual-level voltage shifting, transistors 204 and 208 cannot be fully shut off and static current will flow. Shifter 200 cannot be used for leakage reduction.
For the reasons stated above, and for other reasons stated below which will become apparent to those skilled in the art upon reading and understanding the present specification, there is a need in the art for a dual-level voltage shifter with no static current flow, and which is capable of use with low power circuits.
In some embodiments, the invention includes first and second transistors connected in series, and third and fourth transistors connected in series, each series connection between a first potential and a second potential. The gates of the first and third transistors are connected to complementary inputs. Fifth and sixth transistors are also connected in series, and seventh and eighth transistors are also connected in series, each series connection between a third potential and the second potential. The gates of the fourth and sixth transistors are connected to a first node between the first and second transistors, and the gates of the second and eighth transistors are connected to a second node between the third and fourth transistors.
In other embodiments, a level shifter includes a first transistor configured as a diode, a first CMOS inverter, and a second transistor configured as a diode, the first transistor, first inverter, and second transistor connected in series between a first voltage connection and a second voltage connection. Further, a second inverter and a third inverter are connected in parallel between a third voltage connection and a fourth voltage connection. The second and the third inverters are cross coupled, and the output of the first inverter and the input of the second inverter are operatively coupled together. A pass gate is in the feedback loop between the second and the third inverters.
BRIEF DESCRIPTION OF THE DRAWINGS
Other embodiments are described and claimed.
FIG. 1 is a circuit diagram of a prior art voltage shifter;
FIG. 2 is a circuit diagram of a second prior art voltage shifter;
FIG. 3 is a circuit diagram of an embodiment of the present invention;
FIG. 4 is a diagram showing operation of the embodiment of FIG. 3;
FIG. 5 is a circuit diagram of an alternative embodiment of the present invention;
FIG. 6 is a diagram showing operation of the embodiment of FIG. 5; and
DESCRIPTION OF EMBODIMENTS
FIG. 7 is a block diagram of an integrated circuit embodiment according to one embodiment of the present invention.
In the following detailed description of the embodiments, reference is made to the accompanying drawings which form a part hereof, and in which is shown by way of illustration specific embodiments in which the invention may be practiced. It is to be understood that other embodiments may be utilized and structural changes may be made without departing from the scope of the present invention.
FIG. 3 shows an embodiment of a dual-level voltage shifter 300 of the present invention. Embodiment 300 comprises a first transistor 302 connected in series with a second transistor 304 between a first node 306 which is at a first potential, a second node 308 which is at a second potential, a third transistor 310 and a fourth transistor 312 also connected in series between first node 306 and second node 308, a fifth transistor 314 and a sixth transistor 316 connected in series between a third node 318 at a third potential and the second node 308, and a seventh transistor 320 and an eighth transistor 322 also connected in series between the third node 318 and the second node 308. The gates of first transistor 302 and third transistor 310 are coupled to complementary inputs Vin and Vin*.
A node 324 is defined at the connection of first transistor 302 and second transistor 304. Node 324 is coupled to the gates of fourth transistor 312 and sixth transistor 316. Another node 326 is defined at the connection of third transistor 310 and fourth transistor 312. Node 326 is coupled to the gates of second transistor 304 and eighth transistor 322. The gate of fifth transistor 314 is coupled to the connection between seventh transistor 320 and eighth transistor 322. The gate of seventh transistor 320 is coupled to an output node 328 defined at the connection between fifth transistor 314 and sixth transistor 316.
Transistors 302, 310, 314, and 320 are in one embodiment p-type transistor switches which are closed (on) between their source and drain when their gate is at a low potential, and which are open (off) between their source and drain when their gate is at a high potential. Transistors 304, 312, 316, and 322 are in one embodiment n-type transistors transistor switches which are closed (on) between their source and drain when their gate is at a high potential, and which are open (off) between their source and drain when their gate is at a low potential. The input potential Vin varies from a logical one (Vcc) to a logical zero (0 V). Its complement Vin* varies from a logical zero (0 V) to a logical one (Vcc). In the dual-level voltage shifter embodiment 300, first node 306 is at a supply voltage Vcc, second node 308 is at Vnn=−ΔV, and third node 318 is at Vpp=Vcc+ΔV.
In operation, embodiment 300 functions as follows. When Vin is a logical zero (0 V), transistor 302 turns on, pulling the potential at node 324 to Vcc, turning on transistors 312 and 316. Output node 328 is pulled to Vnn through transistor 316. On the transition of Vin to Vcc, transistor 302 turns off, and transistor 310 turns on. Node 326 is pulled to Vcc through transistor 310, turning transistors 304 and 322 on. Node 324 is pulled to Vnn through transistor 304. Node 330 is pulled to Vnn through transistor 322, and transistor 314 turns on, pulling output node 328 to Vpp. The transitions of node 324 and output node 328 are fast due to the cross coupled feedback connections between transistors 314 and 320 or 304 and 312.
Since at least one device in every potential static current path of embodiment 300 is completely shut off, there is no static current flow in embodiment 300. The voltage shifting accomplished by embodiment 300 is dual-level, with output node 328 voltage shifting between Vpp and Vnn. The voltages Vpp and Vnn can be set to virtually any voltages. In one embodiment, Vpp=Vcc+ΔV and Vnn=−ΔV.
A partial timing diagram showing the operation of embodiment 300 is shown in FIG. 4. As Vin rises from ground (logical zero) to Vcc (logical one), the voltage at node 328 rises from Vnn to Vpp, and the voltage at node 324 drops from Vcc to Vnn.
FIG. 5 shows another embodiment 500 of a dual-level voltage shifter. Embodiment 500 comprises a first branch comprising first transistor 502 and second transistor 504 connected in series between third transistor 506 and fourth transistor 508, a second branch comprising a fifth transistor 510 and a sixth transistor 512 connected in series and a seventh transistor 514 and an eighth transistor 516 connected in series, and a pass gate 518.
Transistors 506 and 508 are connected as current limiting diodes, transistor 506 having its gate connected between transistor 506 and transistor 502, and transistor 508 having its gate connected between transistor 508 and transistor 504. The series connection of transistors 506, 502, 504, and 508 is connected between a first node 520 at a first potential, and a second node 522 at a second potential. In one embodiment, node 520 is at Vcc (logic one) and node 522 is at ground (logic zero). The series connections of transistors 510 and 512, and 514 and 516, are connected in parallel between a third node 524 which is at a third potential, and a fourth node 526 which is at a fourth potential. In one embodiment, node 524 is at Vpp=Vcc+ΔV, and node 526 is at −ΔV.
Pass gate 518 comprises two transistors 530 and 532 connected in parallel. As shown in FIG. 5, transistor 530 is an n-type transistor and transistor 532 is a p-type transistor. The gates of the transistors 530 and 532 are connected to complementary inputs Ven and Ven*, respectively.
Pass gate 518 is connected between node 534 and node 536. Node 534 is also connected to the gates of transistors 510 and 512, and between transistors 502 and 504. Node 536 is connected between transistors 514 and 516. An output node 538 is defined between transistors 510 and 512, and is also connected to the gates of transistors 514 and 516.
Transistors 510 and 512 form a first inverter, and transistors 514 and 516 form a second inverter. The first and second inverters are cross coupled, and pass gate 518 is in the feedback loop. Transistors 502 and 504 form an inverter, and transistors 506 and 508 act as current limiting diodes. The pass gate 518 substantially instantaneously disconnects the feedback loop of the cross coupled inverters to avoid large current involved to flip the cross coupled inverters' state.
The operation of the embodiment 500 is shown in greater detail in FIG. 6. Pass gate 518 is always on when the potential Ven is at logic one. When voltage shifting occurs, pass gate 518 disconnects the feedback loop to avoid the large current involved in flipping the state of the cross coupled inverters. When Vin is logic zero and voltage Ven=logic one, node 534 is initially pulled to near Vcc through transistor 502, with current limited by diode 506, while transistor 504 is turned off. Transistor 510 is turned off and transistor 512 is turned on, and node 538 is pulled to Vnn through transistor 512. This in turn turns transistor 514 on and transistor 516 off. Node 536 is pulled to Vpp through transistor 514. The pass gate 518 passes potential Vpp to node 534. Output node 538 is held at the potential coupled to node 526.
To shift the voltage level of embodiment 500, Ven is set to logic zero, and Vin is raised to logic one (Vcc). When this occurs, transistor 502 shuts off and transistor 504 turns on. The voltage at node 534 is pulled to Vth above ground through transistor 504, with current limited by diode 508, and transistor 508 begins to enter its cutoff region. As the potential at node 534 decreases, output voltage at output node 538 approaches Vpp, turning transistor 516 on and turning transistor 514 off. Voltage at node 536 is pulled to Vnn through transistor 516. Ven is then raised to logic one, connecting the feedback loop through pass gate 518, pulling the potential at node 534 to Vnn. As the potential at node 534 drops, transistor 510 turns on and transistor 512 turns off, pulling output node 538 to Vpp through transistor 510.
When Vin is lowered to logic zero, transistor 504 shuts off, and transistor 502 turns on. This pulls node 534 to near Vcc through transistor 502, with current limited by diode 506. The potential at node 534 turns transistor 510 off and transistor 512 on. Node 538 is pulled to Vnn through transistor 512. The potential at node 538 turns transistor 514 on and transistor 516 off, pulling node 536 to Vpp through transistor 514. Pass gate 518 passes potential Vpp to node 534, and output node 538 is held at Vnn.
Although transistor 504 can conduct static current from ground potential 522 to Vnn due to its forward bias when Vin is at logic one, the current is shut off by transistor 508 which is under a reverse bias. Likewise, when Vin is lowered to logic zero, static current flow through transistor 502 is shut off by transistor 506.
No static current flows in embodiment 500. Embodiment 500 provides dual-level voltage shifting between Vnn and Vpp with no static current. The input potentials Ven and Ven* coupled to the gates of pass gate transistors 530 and 532, respectively, may be generated from the rising and falling transitions of Vin.
As supply voltage Vcc to the application circuit is lowered to reduce the active power of the circuits, the threshold voltage is lowered as well to maintain the speed of the circuit. The low threshold voltage drastically increases the leakage current up to a significant portion of the total power consumed by the application circuit. Leakage reduction techniques that shut off leakage current of idling blocks or an entire microchip in standby mode are becoming important. These techniques require low power voltage level shifting to switch the voltage level of the supplies. The voltage level shifters of the present invention provide dual-level voltage shifting required by modem components without generating static current. They are therefore well suited to leakage reduction techniques.
FIG. 7 illustrates a block diagram of an integrated circuit 700 that comprises an embodiment of the present invention dual-level voltage shifter circuit 704 driving an application integrated circuit 706. The circuit 704 receives an input voltage 702 and its complement 702*, and supply voltages 708, connected to Vpp, and 710, connected to Vnn. The circuit 704 switches its output 716's voltage level between Vpp and Vnn. The application integrated circuit 706 receives the regular supply voltages 712, connected to Vcc, and 714, connected to GND, and receives the shifted voltage 716 from the circuit 704. The dual-level voltage shifter circuit 704 can be arranged as described above with respect to FIGS. 3 and 5. The application integrated circuit may be of any type, including but not limited to a processor, memory, memory controller, or application-specific integrated circuit (ASIC).
As shown in FIG. 7, the integrated circuit 700 may be part of a computer system 750. Such a system could include a desktop system, a portable system, or the like.
It is to be understood that the above description is intended to be illustrative, and not restrictive. Many other embodiments will be apparent to those of skill in the art upon reading and understanding the above description. The scope of the invention should, therefore, be determined with reference to the appended claims, along with the full scope of equivalents to which such claims are entitled.