US 20040263111 A1 Abstract A method and apparatus for use with a controller that uses a flux angle position value to control a three phase induction machine, the method for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D
_{H }that is at least two, the method comprising the steps of injecting a high frequency voltage signal having a high frequency value into the machine thereby generating a high frequency current within the stator windings, obtaining a high frequency feedback signal from the machine, mathematically combining the high frequency value and the dominant harmonic number D_{H }to provide an instantaneous modified angle, using the feedback signal to identify X consecutive calculating instances during each Y consecutive feedback signal cycles where Y is at least two, at each of the X different calculating instances, identifying an instantaneous flux angle position value by mathematically combining a shift angle with the instantaneous modified angle where the shift angles corresponding to each of the X different calculating instances are all different and providing the instantaneous flux angle position value to the controller. Claims(25) 1. A method for use with a controller that uses a flux angle position value to control a three phase induction machine, the method for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D_{H }that is at least two, the method comprising the steps of:
injecting a high frequency voltage signal having a high frequency value into the machine thereby generating a high frequency current within the stator windings; obtaining a high frequency feedback signal from the machine; mathematically combining the high frequency value and the dominant harmonic number D _{H }to provide an instantaneous modified angle; using the feedback signal to identify X consecutive calculating instances during each Y consecutive feedback signal cycles where Y is at least two; at each of the X different calculating instances, identifying an instantaneous flux angle position value by mathematically combining a shift angle with the modified angle where the shift angles corresponding to each of the X different calculating instances are all different; and providing the instantaneous flux angle position value to the controller. 2. The method of 3. The method of _{H }to provide an instantaneous modified angle includes dividing the high frequency value by the dominant harmonic number and integrating. 4. The method of _{H }and X is four. 5. The method of _{H}. 6. The method of 7. The method of _{H }is 2 and first through fourth shift angles during one of the two consecutive feedback signal cycles include values 2π, π/4, π/2 and 3π/4 while first through fourth shift angles corresponding to the other of the two consecutive feedback signal cycles include values π, 5π/4, 3π/2 and 7π/4, respectively. 8. The method of _{H }is 4 and first through fourth shift angles during one of the four consecutive feedback signal cycles include values 2π, π/8, π/4 and 3π/8, first through fourth shift angles corresponding to another of the four consecutive feedback signal cycles include values π/2, 5π/8, 3π/4 and 7π/8, first through fourth shift angles corresponding to another of the four consecutive feedback signal cycles include π, 9π/8, 5π/4 and 11π/8 and first through fourth shift angles corresponding to another of the four consecutive feedback signal cycles include 3π/2, 13π/8, 7π/4 and 15π/8, respectively. 9. The method of 10. The method of _{H }and X is 4. 11. The method of _{H }is one of 2 and 4. 12. The method of 13. A method for use with a controller that uses a flux angle position value to control a three phase induction machine, the method for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D_{H }that is at least two, the method comprising the steps of:
injecting a high frequency voltage signal having a high frequency value into the machine thereby generating a high frequency current within the stator windings; obtaining one of a high frequency zero sequence voltage feedback signal and a high frequency zero sequence current feedback signal from the machine; dividing the high frequency value and the dominant harmonic number D _{H }to provide an instantaneous modified angle; using the feedback signal to identify four consecutive calculating instances during each of Y consecutive feedback signal cycles where Y is at least two; at each of the calculating instances during the Y consecutive feedback signal cycles, identifying an instantaneous flux angle position value by mathematically combining a shift angle with the instantaneous modified angle where the shift angles corresponding to each of the calculating instances during the Y consecutive feedback signal cycles are all unique shift angles; and providing the instantaneous flux angle position value to the controller. 14. The method of _{H }and the shift angles are multiples of 2π/4D_{H}. 15. The method of _{H }is one of two, four and eight. 16. The method of 17. A method for use with a controller that uses a flux angle position value to control a three phase induction machine, the method for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D_{H }that is at least two, the method comprising the steps of:
injecting a high frequency voltage signal having a high frequency into the machine thereby generating a high frequency current within the stator windings; obtaining one of a high frequency zero sequence voltage feedback signal and a high frequency zero sequence current feedback signal from the machine; integrating the feedback signal to generate a quadrature signal; identifying the zero crossing times of each of the feedback signal and the quadrature signal; dividing the high frequency by the dominant harmonic number D _{H }to provide an instantaneous modified angle; at each of the zero crossing times during D _{H }consecutive feedback signal cycles, identifying an instantaneous flux angle position value by mathematically combining a shift angle with the instantaneous modified angle where the shift angles corresponding to each of the zero crossing times during the D_{H }consecutive feedback signal cycles are all unique shift angles and are multiples of 2π/4D_{H}; and providing the instantaneous flux angle position value to the controller. 18. The method of _{H }is one of two and four. 19. An apparatus for use with a controller that uses a flux angle position value to control a three phase induction machine, the apparatus for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D_{H }that is at least two, the apparatus comprising:
a programmed processor performing the steps of: injecting a high frequency voltage signal having a high frequency value into the machine thereby generating a high frequency current within the stator windings; obtaining a high frequency feedback signal from the machine; mathematically combining the high frequency value and the dominant harmonic number D _{H }to provide an instantaneous modified angle; using the feedback signal to identify X consecutive calculating instances during each Y consecutive feedback signal cycles where Y is at least two; at each of the X different calculating instances, identifying an instantaneous flux angle position value by mathematically combining a shift angle with the instantaneous modified angle where the shift angles corresponding to each of the X different calculating instances are all different; and providing the instantaneous flux angle position value to the controller. 20. The apparatus of 21. The apparatus of _{H }to provide an instantaneous modified angle by dividing the high frequency value by the dominant harmonic number and integrating. 22. The apparatus of _{H}, X is four and the shift angles are multiples of 2π/4D_{H}. 23. The apparatus of _{H }is one of two and four. 24. The apparatus of 25. The apparatus of Description [0001] Not applicable. [0002] Not applicable. [0003] The field of the invention is AC induction motor drives and more specifically the area of injecting high frequency voltage signals into an AC induction motor and using high frequency feedback signals to identify stator flux position. [0004] Induction motors have broad application in industry, particularly when large horsepower is needed. In a three-phase induction motor, three phase alternating voltages are impressed across three separate motor stator windings and cause three phase currents therein. Because of inductances, the three currents typically lag the voltages by some phase angle. The three currents produce a rotating magnetic stator field. A rotor contained within the stator field experiences an induced current (hence the term “induction”) which generates a rotor field. The rotor field typically lags the stator field by some phase angle. The rotor field is attracted to the rotating stator field and the interaction between the two fields causes the rotor to rotate. [0005] A common rotor design includes a “squirrel cage winding” in which axial conductive bars are connected at either end by shorting rings to form a generally cylindrical structure. The flux of the stator field cutting across the conductive bars induces cyclic current flows through the bars and across the shorting rings. The cyclic current flows in turn produce the rotor field. The use of this induced current to generate the rotor field eliminates the need for slip rings or brushes to provide power to the rotor, making the design relatively maintenance free. [0006] To a first approximation, the torque and speed of an induction motor may be controlled by changing the frequency of the driving voltage and thus the angular rate of the rotating stator field. Generally, for a given torque, increasing the stator field rate will increase the speed of the rotor (which follows the stator field). Alternatively, for a given rotor speed, increasing the frequency of the stator field will increase the torque by increasing the slip, that is the difference in speed between the rotor and the stator fields. An, increase in slip increases the rate at which flux lines are cut by the rotor, increasing the rotor generated field and thus the force or torque between the rotor and stator fields. [0007] Referring to FIG. 1, a rotating phasor [0008] These two components [0009] Accordingly, in controlling an induction motor, it is generally desired to control not only the frequency of the applied voltage (hence the speed of the rotation of the stator flux phasor [0010] Generally, it is desirable to design FOC strategies that are capable of driving motors of many different designs and varying sizes. Such versatility cuts down on research, development, and manufacturing costs and also results in easily serviceable controllers. Unfortunately, while versatile controllers are cost-effective, FOC controllers cannot control motor operation precisely unless they can adjust the division of d and q-axis currents through the stator windings to account for motor-specific operating parameters. For this reason, in order to increase motor operating precision, various feedback loops are typically employed to monitor stator winding currents and voltages and/or motor speed. A controller uses feedback information to determine how the inverter supplied voltage must be altered to compensate for system disturbances due to system specific and often dynamic operating parameters and then adjusts control signals to supply the desired inverter voltages. [0011] To this end, in an exemplary FOC system, two phase d and q-axis command currents are provided that are calculated to control a motor in a desired fashion. The command currents are compared to d and q-axis motor feedback currents to generate error signals (i.e., the differences between the command and feedback currents). The error signals are then used to generate d and q-axis command voltage signals which are in turn transformed into three phase command voltage signals, one voltage signal for each of the three motor phases. The command voltage signals are used to drive a pulse width modulated (PWM) inverter that generates voltages on three motor supply lines. To provide the d and q-axis current feedback signals the system typically includes current sensors to sense the three phase line currents and a coordinate transformation block is used to transform the three phase currents to two phase synchronous dq frame of reference feedback currents. [0012] In addition to requiring two phase signals and three phase signals to perform 2-to-3 and 3-to-2 phase transformations, respectively, a precise flux position angle estimate θ′ [0013] In an effort to reduce system costs and increase reliability, the controls industry has recently developed various types of sensorless or self-sensing induction machine systems that, as the labels imply, do not include dedicated speed sensing hardware and corresponding cabling but that, nevertheless, can generate accurate position, flux and velocity estimates. Techniques used for operating parameter estimation can be divided into two groups including techniques that track speed dependent phenomenon and techniques that track spatial saliencies in system signals. These techniques generally use disturbances in d and q-axis feedback currents to identify the operating parameters of interest and hence provide additional functionality which, in effect, “piggy-backs” on feedback signals that are obtained for another purpose (i.e., dq current components are already required for FOC). [0014] Because speed dependent techniques depend on speed in order to generate an identifiable feedback signal, these techniques ultimately fail at zero or low (e.g., below 5 Hz) excitation frequency due to lack of signal. In addition, because these methods estimate operating parameters from voltage and current, these techniques are sensitive to temperature varying system parameters such as stator resistance, etc. [0015] One type of saliency tracking technique includes injecting or applying a known high frequency “injection” voltage signal in addition to each of the command voltage signals used to drive the PWM inverter and using feedback current (or voltage) signals to identify saliencies associated with the flux angle. To this end, an exemplary system converts a high frequency command signal into a high frequency phase angle and generates a first injection signal that is the product of a scalar and the sine of the high frequency phase angle. Second and third injection signals are also generated, each of the second and third signals phase shifted from the first signal by 120 degrees. A separate one of the first, second and third signals is then added to a separate one of the three voltage command signals that are used to drive the PWM inverter. [0016] One injection type saliency tracking algorithm to generate a flux position angle estimate without a rotor speed sensor employs a negative sequence of the high frequency current component and is described in an article that issued in the IEEE Transactions on Industry Applications publication, vol. 34, No. 5, September/October 1998 by Robert Lorenz which is entitled “Using Multiple Saliencies For The Estimation Of Flux Position, And Velocity In AC Machines” (hereinafter “the Lorenz article”). The algorithm in the Lorenz article is based on the fact that when a high frequency voltage signal (referred to in the Lorenz article as a “carrier signal”) is injected into a rotating system, a resulting high frequency field interacts with system saliency to produce a “carrier” signal current that contains information relating to the position of the saliency. The carrier current consists of both positive and negative-sequence components relative to the carrier signal voltage excitation. While the positive sequence component rotates in the same direction as the carrier signal voltage excitation and therefore contains no spatial information, the negative-sequence component contains spatial information in its phase. The Lorenz article teaches that the positive sequence component can be filtered off leaving only the negative-sequence component which can be fed to an observer used to extract flux angle position information. [0017] Unfortunately, algorithms like the algorithm described in the Lorenz article only works well if an induction machine is characterized by a single sinusoidally distributed spatial saliency. As known in the art, in reality, motor currents exhibit more than a single spatial saliency in part due to the fact that PWM inverters produce a plethora of harmonics. As a result, the phase current negative sequence comprises a complicated spectrum that renders the method described in the Lorenz article relatively inaccurate. [0018] Injection type saliency tracking algorithms employ a zero sequence high frequency current or voltage component instead of the negative sequence current component. One such technique is described in an article that issued in the IEEE IAS publication, pp. 2290-2297, Oct. 3-7, 1999, Phoenix Ariz., which is entitled “A New Zero Frequency Flux Position Detection Approach For Direct Field Orientation Control Drives” (hereinafter “the Consoli article”). The Consoli article teaches that the main field of an induction machine saturates during system operation which causes the spatial distribution of the air gap flux to assume a flattened sinusoidal waveform including all odd harmonics and dominated by the third harmonic of the fundamental. The third harmonic flux component linking the stator windings induces a third harmonic voltage component (i.e., a voltage zero sequence) that is always orthogonal to the flux component and that can therefore be used to determine the flux position. Unfortunately, the third harmonic frequency is low band width and therefore not particularly suitable for instantaneous position determination needed for low speed control. [0019] The Consoli article further teaches that where a high frequency signal is injected into a rotating system, the injected signal produces a variation in the saturation level that depends on the relative positions of the main rotating field and high frequency rotating field. Due to the fundamental component of the main field, the impedance presented to the high frequency injected signal varies in space and this spatial variance results in an unbalanced impedance system. The unbalanced system produces, in addition to the fundamental zero sequence component of air gap flux and voltage, additional high frequency components having angular frequencies represented by the following equation: ω [0020] where: [0021] ω [0022] ω [0023] ω [0024] where the sign “±” is negative if the high frequency “injected” signal has a direction that is the same as the fundamental field direction and is positive if the injected signal has a direction opposite the fundamental field direction. [0025] In this case, referring to FIGS. 2 [0026] For instance, referring to in FIGS. 2 [0027] At time t2 where voltage V [0028] Unfortunately, as in the case of the negative current component signal employed by Lorenz, high frequency zero sequence feedback signals contain a complicated harmonic spectrum mostly due to the PWM technique employed where the spectrum can be represented by the following equations: ω ω ω ω [0029] where: [0030] ω [0031] U.S. patent application Ser. No. 10/092,046 (hereinafter “the '046 reference”) which is entitled “Flux Position Identifier Using High Frequency Injection With The Presence Of A Rich Harmonic Spectrum In A Responding Signal” which was filed Mar. 5, 2002 and which is commonly owned with the present invention is incorporated herein by reference. Consistent with the comments above, the '046 reference teaches that when a high frequency injection signal is injected into an induction based system which is operating at a stator fundamental frequency, the high frequency signal interacts with the stator field to generate a resulting high frequency current (and corresponding voltage) that has a complicated initial high frequency spectrum that includes a component at the injection frequency as well as components (hereinafter “sideband components”) at various frequencies within sidebands about the injection frequency that are caused by inverter harmonics as well as interaction between system saliencies and the injected signals. The sideband components are at frequencies equal to the injection frequency plus or minus multiples of the fundamental frequency. For instance, where the injection frequency is 500 Hz and the fundamental frequency is 2 Hz, the sideband components may include frequencies of 494 Hz, 496 Hz, 498 Hz, 502 Hz, 504 Hz, 506 Hz, etc. [0032] In addition, the '046 reference recognizes that, given a specific motor control system configuration (i.e., specific hardware and programmed operation), a dominant sideband frequency has the largest amplitude. This dominant sideband frequency for the system configuration always corresponds to the sum of the injection frequency and a specific harmonic of the fundamental where the specific harmonic number is a function of system design and operating parameters. For instance, given a first system configuration, the system specific dominant sideband frequency may be the sum of the injection frequency and the 4 [0033] Moreover, the '046 reference recognizes that during a commissioning procedure, the system specific D [0034] In light of the above realizations, the '046 reference teaches a system designed to strip the injection frequency value out of each initial spectrum frequency thereby generating a low frequency spectrum including a separate frequency corresponding to each of the initial spectrum frequencies. For instance, in the above example where the fundamental and injection frequencies are 2 Hz and 500 Hz, respectively, and assuming sideband frequencies within the initial spectrum at 494 Hz, 496 Hz, 498 Hz, 502 Hz, 504 Hz and 506 Hz, after stripping, the low frequency spectrum includes modified sideband frequencies at −6 Hz, −4 Hz, −2 Hz, 2 Hz, 4 Hz and 6 Hz. [0035] After the low frequency spectrum value has been generated, the '046 reference teaches that the low frequency spectrum can be divided by the system specific dominant harmonic number D [0036] More specifically, at least one embodiment disclosed in the '046 reference filters out the positive sequence components of the high frequency feedback currents and generates stationary high frequency α and β-axis negative-sequence components. These stationary components are orthogonal and together include the noisy initial spectrum about the high injection frequency. [0037] As well known in the art, in the case of any stationary to synchronous component signal conversion, an angle that corresponds to the rotating components must be known. Where the angle is accurate, the resulting synchronous d and q-axis components are essentially DC values. However, where the angle is inaccurate, the resulting components fluctuate and the resulting d and q-axis components are not completely synchronous. [0038] The '046 reference teaches that a phase locked loop (PLL) adaptively generates a high frequency angle estimate that includes components corresponding to all high frequencies in the stationary α and β-axis negative sequence components. The angle estimate is used to convert the stationary high frequency α and β-axis negative-sequence components to synchronous d and q-axis negative-sequence components. Thereafter, one of the d or q-axis components is negated and the resulting negated or difference value is fed to a PI controller or the like to step up the difference value and generate the low frequency spectrum. [0039] The angle estimate is adaptively generated by adding the high injection frequency and the low frequency spectrum to generate a combined frequency spectrum and then integrating the combined frequency spectrum. Thus, the angle estimate is accurate when the combined frequency spectrum matches the actual frequency spectrum that exists in the stationary α and β-axis negative sequence components and, where there is a difference between the combined frequency spectrum and the stationary α and β-axis components, that difference is reflected in the synchronous d and q-axis components which adaptively drive the PI regulator and adjusts the low frequency spectrum. [0040] The low frequency spectrum is combined mathematically with the system specific dominant harmonic number to generate a stator fundamental frequency estimate. After the stator frequency is identified, the stator frequency can be integrated to generate an air gap flux angle estimate F [0041] According to another embodiment described in the '046 reference, instead of employing the three phase feedback currents to identify the complex frequency spectrum, a zero sequence voltage or current signal is employed. To this end, unlike the case where the high frequency current is resolved into quadrature d and q-axis components, the zero sequence embodiment includes a feedback loop that only senses and feeds back a single common mode component. With the zero sequence voltage (or current) feedback signal being a stationary α-axis signal, an artificial stationary β-axis signal is generated by integrating the α-axis signal to generate an integrated signal, low pass filtering the integrated signal to generate a filtered signal and subtracting the filtered signal from the integrated signal thereby providing the high frequency component of the integrated signal as the β-axis signal. [0042] Consistent with the high frequency current example described above, after the α and artificial β-axis components are generated, the stationary α and β-axis signals are converted to synchronous high frequency d and q-axis signals and one of the d or q-axis signals is used to drive the PLL. Operation of the PLL in this embodiment is similar to operation of the embodiment described above. [0043] While the concepts described in the '046 reference are advantageous and suitable for certain applications where PLL capabilities are supported, in other cases such capabilities are not supported or preferably are not supported and therefore some other method for determining the main field flux angle in a rich harmonic system would be advantageous. [0044] It has been recognized that the clearest and most accurately recognizable component of the high frequency zero sequence feedback signal and hence the component optimally used to identify the flux angle is the dominant harmonic component. It has also been recognized, however, that the dominant harmonic zero sequence feedback component cannot be employed directly in a Consoli type flux angle determining algorithm to yield accurate instantaneous flux angle values. To this end, assume that the fundamental frequency is 1 Hz, the frequency of the injected voltage is 500 Hz, a high frequency first harmonic is 501 Hz and that a high frequency second harmonic is 502 Hz. In addition, for the purposes of this explanation, assume that the high frequency second harmonic is the dominant harmonic component. Referring to FIGS. 3 [0045] According to Consoli, each instance corresponding to a zero crossing, maximum or minimum of the high frequency first harmonic V for points for points for points for points [0046] respectively. Similarly, during the next cycle of the injected high frequency voltage, at points [0047] Points [0048] Referring still to FIGS. 3 [0049] Thus, Consoli's Equations 6-9 above do not work with the high frequency second harmonic. Although not illustrated, the Consoli Equations also do not work with other dominant harmonics including the 4th, the 8 [0050] Referring yet again to FIG. 3 [0051]FIG. 3 [0052] Referring specifically to waveforms θ [0053] While using equations 6-9 at times corresponding to maximum, positive to negative zero crossing, minimum and negative to positive zero crossing instances of modified second harmonic (V for points a: θ for points b: θ for points c: θ for points d: θ for points e: θ for points k: θ for points m: θ for points n: θ [0054] Thus, for the first four points a, b, c and d the maximum, minimum and zero crossing times of a leading or first period of the second harmonic zero sequence feedback signal are used to determine main field flux angle θ [0055] In the case of a system that generates a dominant fourth harmonic feedback signal (i.e., DH=4), angle Θ [0056] Thus, one object of the invention is to provide a method and apparatus that identifies the main flux angle in rich harmonic systems that has performance characteristics similar to the characteristics of Consoli. As described above, the present invention performs as well as Consoli despite rich harmonics and irrespective of which harmonic is dominant in a feedback signal. [0057] Another object is to provide a method and apparatus that accurately provides flux angle values in a rich harmonic system. Here instead of providing angle θ [0058] Consistent with the above, the invention includes a method for use with a controller that uses a flux angle position value to control a three phase induction machine, the method for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D [0059] The invention also includes a method for use with a controller that uses a flux angle position value to control a three phase induction machine, the method for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D [0060] In addition, the invention includes a method for use with a controller that uses a flux angle position value to control a three phase induction machine, the method for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D [0061] Furthermore, the invention includes an apparatus for use with a controller that uses a flux angle position value to control a three phase induction machine, the apparatus for determining an instantaneous flux angle position value in the machine where the machine is characterized by a system specific dominant harmonic frequency number D [0062] These and other objects, advantages and aspects of the invention will become apparent from the following description. In the description, reference is made to the accompanying drawings which form a part hereof, and in which there is shown a preferred embodiment of the invention. Such embodiment does not necessarily represent the full scope of the invention and reference is made therefore, to the claims herein for interpreting the scope of the invention. [0063]FIG. 1 is a schematic view in cross section of an induction motor showing instantaneous locations of a rotor flux, a stator mmf and the torque and flux components of the stator mmf; [0064]FIGS. 2 [0065]FIG. 3 [0066]FIG. 4 is a flow chart illustrating an inventive method according to the present invention; [0067]FIG. 5 is a schematic diagram of a motor control system according to the present invention; [0068]FIG. 6 a schematic diagram illustrating one embodiment of the flux angle determiner of FIG. 5; [0069]FIGS. 7 and 8 are schematic diagrams that together illustrate the components of one embodiment of the flux angle estimator of FIG. 6; [0070]FIG. 9 is a phasor diagram illustrating the relationship between various system operating parameters in a system including a high frequency injection voltage; [0071]FIG. 10 is similar to FIG. 9 albeit at a different instant in time; [0072]FIG. 11 is a flow chart illustrating an exemplary method performed by the components of FIGS. 6 and 7 above; [0073]FIG. 12 is a flow chart illustrating an exemplary method performed by the components of FIG. 8; and [0074]FIG. 13 is a graph and timing diagram illustrating a high frequency zero sequence feedback signal and a corresponding quadrature signal and timing diagrams corresponding to the components illustrated in FIG. 8. [0075] In the description that follows, an “*” superscript denotes a command signal, an “f” subscript denotes a feedback signal, an “h” subscript denotes a high frequency signal, an “i” denotes that a corresponding signal relates to a current signal, a “V” denotes that a signal relates to a voltage signal, a “d” subscript denotes that a signal corresponds to a synchronous d-axis, a “q” subscript denotes that a signal corresponds to a synchronous q-axis, “u”, “v” and “w” subscripts denote that corresponding signals relate to each of first, second and third system phases, a “zs” subscript also denotes a zero sequence signal, a “sw” subscript denotes a square wave, an “ox” subscript denotes a stationary α-axis signal and a “β” subscript denotes a stationary β-axis signal. [0076] While the following description details various blocks, steps, and functions, it should be remembered that all of these elements are meant to be implemented in software as computer programs and represent algorithms for execution by a conventional-type digital processor adapted for industrial applications. Hereinafter a general inventive method will first be described and thereafter a more detailed exemplary method will be described in the context of a control system. [0077] Referring now to the drawings wherein like reference characters represent similar elements and signals throughout the several views and, more specifically, a referring to FIG. 4, a general method [0078] At block [0079] Now, for a more detailed description of an exemplary embodiment of the invention, reference is made to FIG. 5 where the present invention will be described in the context of an exemplary motor control system [0080] Generally, system [0081] Command currents i [0082] Transformer [0083] Referring still to FIG. 5, in addition to command currents i* [0084] Sine table [0085] The peak high frequency amplitude signal V [0086] Referring still to FIG. 5, the feedback currents from the two of the three motor phases are provided to the analog to digital converter [0087] Notch filter [0088] The three phase currents output by notch filter [0089] Referring still to FIG. 5, some embodiments will include identifier [0090] Referring still to FIG. 5, identifier [0091] Referring now to FIG. 6, exemplary components of flux angle determiner [0092] Referring now to FIGS. 7 and 8, components that comprise one embodiment of the flux angle estimator [0093] Each of signals α [0094] Referring now to FIG. 8, estimator [0095] Flip-flop module [0096] Referring still to FIG. 13, the third flip-flop FF- [0097] Each one of the first through fourth flip-flops provides its output to two additional flip-flops which, in the fashion similar to that described above, generates two other square waves where each of the other square waves has a frequency which is half that of the received signal. For example, referring still to FIG. 13, waveform β [0098] Referring still to FIG. 13, importantly, waveforms β [0099] Referring again to FIGS. 8 and 13, pulse generator [0100] Referring still to FIG. 8, switch module [0101] Referring still to FIG. 8, switch [0102] As illustrated in FIG. 8, each of first inputs ζ [0103] Summing module [0104] During periods when value θ [0105] As its label implies, hold module [0106] Referring now to FIG. 11, an exemplary more detailed method [0107] Continuing and, referring to FIGS. 5 and 11, at block [0108] Referring now to FIG. 12; a continuation [0109] As described above, where the dominant harmonic number D [0110] As another instance, where D [0111] It should be understood that the methods and apparatuses described above are only exemplary and do not limit the scope of the invention, and that various modifications could be made by those skilled in the art that would fall under the scope of the invention. [0112] To apprise the public of the scope of this invention, the following claims are made: Referenced by
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