US20070235135A1

US20070235135A1 - Plasma processing apparatus

Info

Publication number: US20070235135A1
Application number: US11/513,233
Authority: US
Inventors: Ryoji Nishio; Tsutomu Iida
Original assignee: Hitachi High Technologies Corp
Current assignee: Hitachi High Tech Corp
Priority date: 2006-04-07
Filing date: 2006-08-31
Publication date: 2007-10-11
Also published as: JP4928817B2; JP2007281205A; KR100749169B1

Abstract

A plasma processing apparatus including a vacuum vessel, a lower electrode provided in the vacuum vessel to place a sample thereon, a matcher connected to the lower electrode, and a power supply for supplying power to the lower electrode via the matcher includes an electrostatic chuck electrode provided within the lower electrode to hold the sample, and a voltage measurement circuit provided within the lower electrode to measure a voltage at the electrostatic chuck electrode and output the measured voltage as a DC voltage.

Description

BACKGROUND OF THE INVENTION

The present invention relates to a technique for manufacturing semiconductors. In particular, the present invention relates to a plasma processing apparatus suitable for conducting plasma processing on semiconductor wafers by using plasma.
As the degree of integration of semiconductor elements becomes higher in recent years, the circuit pattern goes on becoming finer. Accordingly, demanded working dimension precision is becoming stricter and stricter. Furthermore, the diameter of the wafer has become as large as 300 mm with the object of reducing the manufacturing cost of semiconductor elements. However, it is demanded to make plasma uniform in a wide range between the center of the wafer and the vicinity of the outer periphery and make high-quality uniform working possible with the object of increasing the yield. In the product processing, it is typical to apply a high-frequency bias in order to form a fine circuit pattern by using anisotropy working. At this time, values of a high frequency voltage and a self bias voltage generated on the wafer are important parameters in working. It becomes important to monitor them accurately.
In order to achieve such an object, it has heretofore been conducted to detect a high frequency voltage between a wafer and a matcher for a high frequency power supply (see, for example, JP-A-2003-174015 and JP-A-2002-203835 corresponding to U.S. Pat. No. 6,771,481).
Apart from this, as to the influence of a high frequency transmission path upon the high frequency voltage, current and phase difference, it is known that a high frequency waveform in an output part of the high frequency coupling circuit differs from that on the wafer and consequently a wafer potential probe technique of directly measuring a wafer potential to obtain information of the wafer potential is effective (see, for example, JP-A-2001-338917).
Furthermore, in the conventional technique, in a parallel plate plasma generation apparatus including an upper plate electrode of a metal material and a lower wafer (functioning as an electrode), a high frequency bias having the same frequency is applied to each of the upper electrode and the lower electrode (wafer). A technique of monitoring the voltages and phases of the upper electrode and the lower electrode in order to control the high frequency voltage phase between those biases is known (see, for example, JP-A-8-162292).

SUMMARY OF THE INVENTION

A phenomenon that poses a problem in the plasma processing apparatus is resonance caused by an inductance and a stray capacitance in a high frequency power feeding system or a capacitance of an ion sheath generated on a front face of an electrode capacitance-coupled to plasma, such as the wafer. Resonance caused by the stray capacitance and the inductance of the power feeding system and resonance caused by the capacitance of the ion sheath and the inductance of the power feeding system are independent of each other. In other words, the two resonance phenomena occur at the same time. This poses a problem that information such as a voltage obtained from a measurement point indicates a value that is widely different from a state such as a voltage that is being actually generated on the wafer or the electrode. A problem of the conventional technique is that these resonance phenomena are not taken into consideration essentially.
Clearly, the technique described in JP-A-2003-174015 has a precondition that information obtained from a measurement point, such as a voltage, is the same as the information concerning the wafer or has the same quality as the information concerning the wafer. If this precondition is not satisfied, the precision of the present technique is degraded remarkably.
In JP-A-2002-203835 corresponding to U.S. Pat. No. 6,771,481, attention is paid to the fact that the precondition is not satisfied in the typical plasma processing apparatus. In accordance with the present invention, it is possible to obtain information such as a load impedance seen from the wafer, besides the voltage, current and phase on the wafer, from the information at the measurement point by precisely specifying an equivalent circuit between a measurement point for the voltage or the like and the wafer. Even if the present technique is used, however, the influence of the resonance phenomenon in question cannot be avoided. Because the inductance component and the stray capacitance causing the resonance are incorporated in an equivalent circuit according to the present technique, but the capacitance of the paired ion sheath is not incorporated into the equivalent circuit. This resonance phenomenon caused by plasma is a phenomenon that cannot be predicted from the view point of the present technique.
In addition, it is very difficult, and substantially impossible, to incorporate the capacitance of the ion sheath into the equivalent circuit and evaluate it accurately. Because the capacitance of the sheath depends upon plasma characteristics (an electron density, an electron temperature and a gas density and distribution of them on the wafer), which in turn depend upon a large number of parameters such as the gas pressure and gas component and high frequency power for plasma generation, and high frequency power for bias applied to the wafer, and consequently the capacitance value of the sheath cannot be calculated accurately. As a matter of course, there is a theory for calculating the capacitance. However, it is not possible to know accurate values of numerical values to be substituted into the theory. In other words, precision assurance cannot be conducted.
Furthermore, the capacitance of the ion sheath is a major element that determines a value of the load impedance seen from the wafer. A high frequency voltage generated on the wafer depends upon a combination of a circuit ranging from a matching circuit to the wafer and the load impedance. However, the capacitance of the ion sheath has a property that it depends upon the high frequency voltage generated on the wafer. In other words, the capacitance and the wafer voltage depend on each other and they are related by a nonlinear relation. As for determination of the capacitance and the wafer voltage, therefore, it cannot be solved by using ordinary equivalent circuit simulation. They cannot be determined without executing a recursive calculation using a numerical computation method. It is very difficult to conduct the present calculation in real time from the viewpoint of both aligning numerical values of basic data for calculation start and the calculation time.
A conclusion obtained from the foregoing description is that the resonance phenomenon in question cannot be solved by using the technique of using the equivalent circuit. As a result, calculation cannot be conducted even if the equivalent circuit is used, or the precision assurance cannot be conducted.
As compared with the technique disclosed in JP-A-2003-174015 or JP-A-2002-203835 corresponding to U.S. Pat. No. 6,771,481, the technique disclosed in JP-A-2001-338917 is a technique of directly measuring the wafer potential and the resonance phenomenon in question can be avoided in principle. However, the present technique has a problem of reliability, and it is difficult to put the present technique to practical use. According to the present technique, an oxide film or a nitride film located on the back of the wafer is broken through by a hard needle of WC (tungsten carbide), and direct measurement of the wafer voltage is implemented. A problem is that it cannot be ensured to break through the film on the back of the wafer certainly and implement stable measurement in a semiconductor manufacturing apparatus for processing five hundred thousand to one million wafers one after another. It is very difficult to design such a structure.
As regards the phase as well, it is well known that there is a large change between phases before and after the resonance point and the phase is inverted in an extreme case. Also in the technique of conducting the phase control as described in JP-A-8-162292, the control performance is crippled by the resonance in question. The resonance in question is a phenomenon that the inductance of the high frequency transmission path and the capacitance of the ion sheath cause resonance. The resonance in question is a phenomenon that occurs not only when a high frequency bias is applied to the wafer but also when a high frequency bias is applied to the electrode opposed to the wafer as described in JP-A-8-162292. In JP-A-8-162292 as well, the resonance in question is not taken into consideration as regards the phase measurement point, and it is appreciated that the resonance phenomenon in question constitutes a serious hindrance in the same way as JP-A-2003-174015, JP-A-2002-203835 corresponding to U.S. Pat. No. 6,771,481, and JP-A-2001-338917.
Hereafter, a resonance phenomenon found by the present inventors will be described in detail. Here, an electrode mounting the wafer is taken as an example. However, these two resonance problems occur entirely in the same way, as regards any electrode capacitance-coupled to plasma. First, it will now be described that a resonance phenomenon is seen even if plasma is not present, by representing the electrode structure by the use of an equivalent circuit and taking measurement of a voltage (here, a peak-to-peak voltage Vpp) as an example. This is first resonance, i.e., resonance caused by a stray capacitance and an inductance of the high frequency transmission system. Subsequently, a resonance phenomenon in the case where there is plasma will be described. This is second resonance, i.e., resonance caused by the capacitance of the ion sheath and the inductance of the high frequency transmission system. The entirely same conclusion is obtained as regards the phase measurement as well.
The first resonance, i.e., resonance caused by the stray capacitance and the inductance of the high frequency transmission system will now be described. FIG. 1 schematically shows a block diagram of components ranging from a wafer bias RF power supply to an electrode. Beginning with an output of the wafer bias RF power supply, a matching circuit, a Vpp detector, a power feeding cable, and an electrode are included in the cited order. The components ranging from the RF power supply to the power feeding cable are in the air, and the electrode for mounting the wafer is in the vacuum. A circuit shown in FIG. 2 is obtained by replacing each of the blocks shown in FIG. 1 with an equivalent circuit. The power feeding cable is an ordinary coaxial line, and it includes an inductance (L1+L2) of a central conductor and a stray capacitance (C1). The electrode is divided into a high frequency transmission part (having an equivalent circuit that is the same as the coaxial structure has) and a spray deposit (C3+R1) for electrostatic chucking on a wafer. A high voltage probe (8 pF and 10 MΩ) for voltage measurement is connected to the wafer. Since the impedance is very high and it can be neglected, however, it is not written in the equivalent circuit. The equivalent circuit shown in FIG. 2 is a typical one. As for the actual electrode, a large number of contrivances, such as a focus ring, have been executed and in addition, stray capacitances represented by Cs1 and Cs2 are included. Therefore, the equivalent circuit becomes more complicated than that shown in FIG. 2.
A result obtained by measuring frequency characteristics with the configuration shown in FIG. 1 by using the actual electrode is shown in FIG. 3. Its abscissa indicates a frequency applied as a bias, and its ordinate indicates a ratio between voltages in positions V1 and V2 shown in FIG. 2. It is appreciated that some resonance points appear at frequencies of 4 MHz or above. Therefore, the inductance and capacitance of the electrode are measured to generate an equivalent circuit, and simulation is conducted. This result is shown in FIG. 4. It is found that the measured resonance phenomenon can be reproduced. This can be understood by means of the typically known resonant frequency represented by the following expression (1).
$\begin{matrix} fo = \frac{1}{2 π \sqrt{L C}} & (1) \end{matrix}$
In the equivalent circuit shown in FIG. 2, a total inductance Lt of the transmission line is approximately 1.7 μH and a total stray capacitance Ct of the transmission line and the electrode is approximately 908 pF. Substituting them into the Expression 1 yields 4.1 MHz, and the result of the measurement can be explained well. Although the resonance phenomenon itself can be reproduced by the simulation, the voltage ratio cannot be reproduced. This is because it is scarcely possible to replace electrical characteristics of the actual structure by such an accurate equivalent circuit that the measurement precision can be assured.
If resonance occurs at 4 MHz as heretofore described, then the reliability of voltage measurement conducted when using a frequency lower than the resonant frequency but higher than at least 2 MHz in this case, although it also depends on the resonance bandwidth (Q value). It is important that the inductance Lt and the stray capacitance Ct are respectively 1.7 μH and 908 pF, and consequently they are not extremely large values. They are the inductance and stray capacitance generated easily by connecting a high frequency transmission path having a length of several meters to the electrode. According to the experience of the present inventors, it is necessary to take this resonance phenomenon into consideration when using a bias having a frequency of at least 1 MHz, although it depends upon the design technique and the apparatus configuration.
The second resonance, i.e., the resonance caused by the capacitance of the ion sheath and the inductance of the high frequency transmission system will now be described. If there is plasma, the wafer is capacitance-coupled to the plasma. Therefore, it becomes necessary to take a new capacitance generated by the plasma into consideration. In addition, when there is plasma, it is conceivable that there is a case where the resonant frequency further falls as compared with the case shown in FIG. 3 or 4. In this new capacitance, the capacitance of the ion sheath formed on the front face of the wafer becomes dominant. A thickness d_shof this ion sheath is theoretically given by the following expression (2).
$\begin{matrix} d_{sh} = 1.36 \frac{\sqrt{2}}{3} {λ_{db} (\frac{2 {eV}_{sh}}{k_{B} T_{a}})}^{0.75} & (2) \end{matrix}$
Here, λ_dbis the Debye length, e is the elementary charge, k_Bis the Boltzmann's constant, and T_eis the electron temperature. An average voltage V_shof the sheath can be defined by the following expression (3).
$\begin{matrix} V_{sh} = \frac{1}{2 π} \int_{0}^{2 π} (V_{S} (τ) - V_{B} (τ)) \partial τ & (3) \end{matrix}$
Here, τ is an angular frequency of the bias, V_s(τ) is a plasma space potential, and V_B(τ) is a bias potential.
The final capacitance of the ion sheath is represented by the following expression (4) using the thickness d_shof the ion sheath.
$\begin{matrix} C_{sh} = \frac{ɛ_{0} S_{W}}{d_{sh}} & (4) \end{matrix}$
Here, ε₀is the dielectric constant of the vacuum, and S_Wis an area of the wafer.
Since the area of the wafer is constant in the Expression 4, it is appreciated that the capacitance of the ion sheath is in inverse proportion to the thickness of the ion sheath. In other words, a condition under which the thickness of the ion sheath becomes thin is equivalent to a condition under which the resonant frequency becomes low. The Debye length is the basic length of the electric field shielding capability, and it becomes short in inverse proportion to the density of the plasma. In the plasma, the electron temperature changes only by several tens percents at most. Neglecting the electron temperature change accordingly, it is appreciated from the Expression 2 that a condition under which the thickness of the ion sheath becomes thin is satisfied when the plasma density is high and the bias voltage is low. A conclusion obtained from this is that the resonant frequency in question is not constant but it changes depending upon the plasma generation condition and wafer working condition even in the same apparatus or if the apparatus is different.
Typically, in plasma used for working on semiconductor products, the electron temperature is approximately 3 eV and the plasma density is in the range of 10¹⁰to 10¹²cm⁻³. Furthermore, the bias voltage is in the range of 100 to 4,000 Vpp. The capacitance of the ion sheath obtained from this is in the range of approximately 200 to 8,000 pF. By using these values, the resonance is simulated. A schematic equivalent circuit is shown in FIG. 5. In the equivalent circuit shown in FIG. 5, a plasma load is added to the equivalent circuit shown in FIG. 2. Letting C5=2000 pF and R3=160Ω (corresponding to the wafer of 300 mm) as a typical plasma circuit, a result shown in FIG. 6 is obtained. It is appreciated from the result shown in FIG. 6 that the resonant frequency falls to 3 MHz. As appreciated from FIG. 5, there is C3, which is a capacitance of the electrode spray deposit, in series with C5. A composite capacitance of C3 and C5 causes resonance with inductances (L1 to L4) on the transmission line. Supposing that C3=7500 pF (corresponding to the wafer of 300 mm), the composite capacitance becomes equal to 1579 pF. Substituting this value and 1.7 μH which is the inductance Lt of the transmission line into the Expression 1, a value of 3.1 MHz is obtained and the simulation result is explained well. This indicates that the resonant frequency at the time when there is plasma depends on the composite capacitance of the capacitance of the ion sheath and the capacitance of the electrode spray deposit and the inductance of the transmission line. Since the capacitance of the electrode spray deposit assumes a value unique to the apparatus, it can be concluded that the resonance phenomenon is generated by the inductance of the transmission line and the capacitance of the ion sheath.
This is verified by actually using the apparatus. FIG. 7 shows frequency characteristics obtained when the wafer bias power supply is output so as to make Vpp on the electrode equal to a constant voltage of 20 V. As predicted from the theory, the resonant frequency becomes extremely low. In this case, the resonant frequency becomes 2 MHz or below. Supposing that the inductance Lt of the transmission line is 1.7 μH in calculation, the composite capacitance is estimated to be approximately 4,300 pF. Since Vpp is extremely low in this case, the capacitance of the sheath amounts to approximately 10,000 pF. It is appreciated that the resonant frequency falls remarkably when the bias voltage is low, as predicted in accordance with the theory described heretofore.
Conclusions and problems obtained heretofore are put together. First, there are two resonance phenomena in question. The first resonance phenomenon is generated by the inductance and the stray capacitance of the high frequency transmission line. The second resonance phenomenon is generated by the inductance of the high frequency transmission line and the capacitance of the ion sheath. On the basis of this principle, it is impossible that the resonance phenomenon itself disappears. The resonant frequency based on the capacitance of the ion sheath has strong dependence on the bias voltage and the plasma density. The resonant frequency based on the capacitance of the ion sheath changes greatly depending upon the wafer processing condition.
As a matter of course, it is more advantageous to lower these inductances and capacitances because the resonant frequency is raised in accordance with the (Expression 1). When the frequency of the high frequency used as the bias is in the vicinity of the resonant frequency, the voltage value measured at the measurement point becomes widely different from the voltage that is actually generated on the wafer. Furthermore, the ratio between the voltage at the measurement point and the voltage on the wafer changes according to the wafer processing condition, and the ratio does not assume a constant value. It is substantially impossible to quantitatively calculate the voltage generated on the wafer, by using the equivalent circuit. As regards the phase and the current measurement as well, the conclusion is the same.
Dimensions of wafers in the semiconductor processing apparatuses, liquid crystal substrates, and so on have been extended from the past to the present time. This aims at reducing the manufacturing cost. This tendency can be expected to continue hereafter as well, although it depends on the development of the technique as well. An increase in dimensions, i.e., in area of the substrate such as the wafer lowers the resonant frequency, because it increases the capacitance of the sheath as represented by the Expression 4. Therefore, the technique provided by the present invention becomes a technique indispensable to applying a high frequency in future semiconductor manufacturing.
An object of the present invention is to provide a technique by which voltage and phase measurement can be easily set to arbitrary goal precision even under presence of the resonance phenomenon.
In order to achieve the object, in accordance with a first aspect of the present invention, a plasma processing apparatus including a vacuum vessel, a lower electrode provided in the vacuum vessel to place a sample thereon, a matcher connected to the lower electrode, and a power supply for supplying power to the lower electrode via the matcher includes an electrostatic chuck electrode provided within the lower electrode to hold the sample, and a voltage measurement circuit provided within the lower electrode to measure a voltage at the electrostatic chuck electrode and output the measured voltage as a DC voltage.
In accordance with a second aspect of the present invention, a plasma processing apparatus including a vacuum vessel, a lower electrode provided in the vacuum vessel so as to incorporate an electrostatic chuck electrode for holding a sample, a matcher connected to the lower electrode, and a power supply for supplying power to the lower electrode via the matcher includes a voltage measurement circuit provided under atmospheric pressure to measure a voltage at the electrostatic chuck electrode and output the measured voltage as a DC voltage, and a coaxial line for connecting the electrostatic chuck electrode to the voltage measurement circuit.
According to the present invention, it is possible to implement a detection circuit that is not susceptible to the influence of resonance even if the resonance is present. As a result, the high frequency voltage and phase can be detected accurately. Furthermore, it becomes possible to run the operation of the plasma processing apparatus stably in an optimum state.
Other objects, features and advantages of the invention will become apparent from the following description of the embodiments of the invention taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of components included in the range of a wafer bias RF power supply to an electrode;

FIG. 2 is an equivalent circuit diagram for the block diagram shown in FIG. 1;

FIG. 3 is a diagram showing frequency characteristics in the configuration shown in FIG. 1;

FIG. 4 shows a result of simulation conducted by using the equivalent circuit shown in FIG. 2;

FIG. 5 is an equivalent circuit diagram for a range from the wafer bias RF power supply to plasma;

FIG. 6 shows a result of simulation conducted by using the equivalent circuit shown in FIG. 5;

FIG. 7 is a frequency characteristic diagram obtained when Vpp on the electrode is set equal to a constant voltage of 20 V;

FIG. 8 is an equivalent circuit diagram obtained when a Vpp detector is incorporated into the electrode;

FIG. 9 is a schematic diagram showing a first embodiment of a plasma etching apparatus;

FIG. 10 is a schematic diagram showing a second embodiment of a plasma etching apparatus;

FIG. 11 is an equivalent circuit diagram for the configuration shown in FIG. 10;

FIGS. 12A-12C show results of simulation conducted by using the equivalent circuit shown in FIG. 11;

FIG. 13 is an equivalent circuit diagram obtained when a phase detector is incorporated into the electrode;

FIG. 14 is an equivalent circuit diagram obtained when the phase detector is disposed outside the electrode;

FIGS. 15A and 15B show results of phase difference simulation conducted using the equivalent circuit shown in FIG. 14; and

FIG. 16 is a schematic diagram showing a plasma etching apparatus according to a third embodiment.

DETAILED DESCRIPTION OF THE INVENTION

As described above, the resonances do not disappear, and correction using a calculation or calibration cannot be conducted. Therefore, it is appreciated that it is important in attaining the object to configure the apparatus so as to cause the voltage and phase information at the measurement point to be equivalent to or have the same quality as the voltage and phase information at an electrode of the measurement subject (an electrode capacitance-coupled to plasma on the wafer or the like). Specifically, it is important to form a configuration having a detection circuit that is not susceptible to the influence of resonance even if the resonances are present.
Such a configuration can be achieved by incorporating the Vpp detector incorporated in the matcher shown in FIGS. 1 and 2 into the electrode. This configuration is shown in FIG. 8. According to this configuration, the Vpp detector becomes unsusceptible to the influence of L1 to L4 causing the resonance and it becomes possible to convert a voltage generated directly at the electrode to a DC voltage and output the DC voltage.
Hereafter, a first embodiment obtained by making the structure shown in FIG. 8 concrete will be described.
FIG. 9 is a longitudinal section diagram of an etching chamber used in the present invention. In the present embodiment, an example of a VHF plasma etching apparatus for forming plasma by utilizing a VHF (Very High Frequency) and a magnetic field is shown. An upper opening part including a cylindrical processing vessel 104, a platelike antenna electrode 103 formed of a conductor such as silicon, and a dielectric window 102 formed of quartz and sapphire capable of transmitting electromagnetic waves is placed on a vacuum vessel 101 via a vacuum seal material 127, such as an O-ring, so as to be hermetically sealed. A processing chamber 105 is formed inside. A magnetic field generating coil 114 is provided on an outer periphery part of the processing chamber 104 so as to surround the processing chamber. The antenna electrode 103 has a perforated structure for letting an etching gas flow. A flon gas such as CF₄, C₄F₆, C₄F₈, C₅F₈, CHF₃or CH₂F2, an inert gas such as Ar or N₂, or O₂or a gas containing an oxide such as CO is controlled by a flow rate adjuster (not illustrated) including an MFC (mass flow controller) provided in a gas supplier 107, and led into the processing chamber 105 via the gas supplier 107. Furthermore, a vacuum exhauster 106 is connected to the vacuum vessel 101. The inside of the processing chamber 105 is kept at a predetermined pressure by a vacuum exhauster (not illustrated) including an MP (turbo-molecular pump) provided in the vacuum exhauster 106 and a pressure governor (not illustrated) including an APC.
A coaxial line 111 is provided over the antenna electrode 103. A high frequency power supply for plasma generation (first high frequency power supply) 108 (having, for example, a frequency of 200 MHz) is connected to the antenna electrode 103 via the coaxial line 111, a coaxial waveguide 125 and a matcher 109. A substrate electrode 115 on which a wafer 116 can be disposed is provided in a lower part in the vacuum vessel 101. In the same way as the antenna electrode 103, a coaxial line 151 is provided under the substrate electrode 115. A wafer bias power supply (second high frequency power supply) 119 (having, for example, a frequency of 4 MHz) is connected to the substrate electrode 115 via the coaxial line 151, a coaxial waveguide 152, a power feeding cable 153, and a matcher 118. The coaxial line 151 and the coaxial waveguide 152 are, for example, the high frequency transmission part in the electrode shown in FIG. 2, and they are in the vacuum. The power feeding cable 153 is on the atmospheric pressure side. An electrostatic chuck electrode 124 having an electrostatic chuck function for adsorbing the wafer 116 electrostatically is buried in the substrate electrode 115. An electrostatic chuck power supply 123 is connected to the electrostatic chuck electrode 124 via a filter 122. The filter 122 passes through DC power from the electrostatic chuck power supply 123, and effectively cuts off power from the plasma generation high frequency power supply 108 and the wafer bias power supply 119.
In the present configuration, a wafer voltage measurement circuit 154 is incorporated right under the electrostatic chuck electrode 124 in the vacuum. The influence of the resonance is eliminated by thus attaching the measurement circuit directly to a place where the voltage to be measured is generated, converting the measured voltage to a DC voltage on the spot, and taking out a resultant signal to the outside of the vacuum. A composite impedance of C6 and C7 in the voltage measurement circuit 154 shown in FIG. 8 must be sufficiently high. To which degree the composite impedances must be high will be described with reference to a second embodiment. However, this method has several problems. The problems are: (1) electric parts (such as resistors, capacitors, coils and diodes) in use are premised on use in the atmosphere, and the performance is not assured for use in the vacuum; (2) since heat generation from the electric parts is inevitable and little heat is radiated in the vacuum, continuous use is impossible; the possibility that part degradation will be caused by a corrosive gas is high; (4) when film deposition occurs, the possibility that circuit operation will be affected is high; (5) the possibility that the circuit will be damaged by turnaround of the high frequency for plasma generation is high; and (6) the possibility that the circuit will be damaged or the circuit operation will be affected by plasma generated around the circuit because of turnaround of the high frequency for plasma generation is high. Each of these problems is not insoluble. For example, the problems can be solved by burying the whole of the voltage measurement circuit 154 into resin, housing the whole of the voltage measurement circuit 154 into a hermetically sealed structure to protect the voltage measurement circuit 154 from the corrosive gas, and housing the whole of the voltage measurement circuit 154 into a hermetically sealed vessel that can be shielded electromagnetically.
A second embodiment in which the problems of the first embodiment are solved more thoroughly is shown in FIG. 10.
In the present configuration, the voltage measurement point is the electrostatic chuck electrode 124 in the same way as FIG. 9. This voltage is taken out to the outside of the vacuum by using a coaxial cable 157. The voltage taken out to the outside of the vacuum is converted to a DC voltage signal by using the voltage measurement circuit 154. This configuration has a merit that the demerit of the configuration shown in FIG. 9 is eliminated because the voltage measurement circuit 154 can be disposed on the atmosphere side. As regards the voltage measurement, the above-described resonance phenomenon loses no relation because it suffices that the voltage at the electrostatic chuck electrode 124 is equal to the voltage at the voltage measurement circuit 154.
A special contrivance becomes necessary in the coaxial cable 157 and the voltage measurement circuit 154 in order to make the voltage at the electrostatic chuck electrode 124 equal to the voltage at the voltage measurement circuit 154.
An equivalent circuit for the apparatus shown in FIG. 10 is shown in FIG. 11. The equivalent circuit shown in FIG. 11 differs from the equivalent circuit shown in FIG. 8 in that a coaxial cable is inserted between the electrode and the voltage measurement circuit. The above-described special contrivance is to make a composite impedance Zs of the coaxial cable and the voltage measurement circuit sufficiently higher than a load impedance Zp inclusive of the plasma. If Zs is small, then a voltage drop is caused by Zs and large reactive current flows, resulting in a heavy burden on the transmission system. If an RF power supply shown in FIG. 11 is controlled to output constant power, then such a demerit is not eliminated completely, but it can be suppressed to a negligible level in an allowable range.
The relation between Zp and Zs will now be described in detail. When seen from the RF power supply shown in FIG. 11, Zp and Zs are connected in parallel as a load circuit. Therefore, a load impedance Z at the time when Zs is not coupled is Z=Zp, whereas a load impedance Z′ at the time when Zs is coupled becomes Z′=Zp·Zs/(Zp+Zs). On the other hand, when the RF power supply is used in power control, V1 which is the voltage to be measured is determined by V1=(WZ)̂0.5 where W is RF power. As a result, the ratio between a voltage V1′ at the time when Zs is coupled and a voltage V1 at the time when Zs is not coupled is represented by V1′/V1=(Zs/(Zp+Zs))̂0.5. Letting V1′/V1=α, α represents precision of the measured voltage value in the state in which the voltage measurement circuit is coupled. Thereafter, a is a value in the range of 0 to 1. From the foregoing description, the relation between Zp and Zs is represented by the following expression (5) using α.
$\begin{matrix} Zs = \frac{α^{2}}{1 - α^{2}} Zp & (5) \end{matrix}$
If, for example, the voltage detection precision is made at least 95%, then it is appreciated from this expression that Zs must have an impedance that is at least 9.3 times as large as Zp. It is also possible to replace C6 and C7 in the voltage measurement circuit by resistors. Unless resistances of the resistors are sufficiently high (for example, at least 10 MΩ), however, power loss is caused in the resistors. Accordingly, care should be taken.
This will now be described by using concrete numerical values. A composite impedance Zp obtained by seeing the plasma side from the place of V1 is calculated by using C5=2000 pF, R3=160Ω and other constants. As a result, |Zp|=approximately 15Ω is obtained.
A composite impedance Zs obtained by seeing the voltage measurement circuit side from the place of V1 will now be found. Supposing a coaxial cable corresponding to 3D2V from the viewpoint of the withstand voltage, inductance and capacitance per unit length become 0.27 μH/m and 103 pF/m, respectively. These correspond to L5, L6 and C9 shown in FIG. 11. Supposing that composite capacitance of C6 and C7 is 8 pF and the coaxial cable has a length of 1 m, it follows that Zs=−355i Ω, where i is an imaginary number. It follows that |Zs/Zp|=24, and the measurement precision can be made sufficiently high.
Equivalent circuit simulation results obtained by using circuit constants heretofore described are shown in FIGS. 12A-12C. FIG. 12A shows a voltage ratio between V1 and V2 (indicated in FIG. 11) obtained when the voltage measurement circuit is not connected, and shows the same result as that of FIG. 6. V1/V2 ratio and V1/V3 ratio obtained when the voltage measurement circuit having the above-described circuit constants is connected are shown in FIGS. 12B and 12C, respectively. Comparing the V1/V2 ratio shown in FIG. 12A with that shown in FIG. 12B, it is appreciated that an influence of the voltage measurement circuit is noticeable at 40 MHz or above, but the influence of the voltage measurement circuit is hardly noticeable at 10 MHz or below.
In the V1/V3 ratio shown in FIG. 12C, a drop in voltage ratio caused by resonance in the vicinity of 40 MHz is noticeable. Calculation of the resonant frequency in the voltage measurement circuit is conducted as described below.
First, a composite impedance of L6, C6 and C7 shown in FIG. 11 will now be found. L6 corresponds to a coaxial cable having a length of 50 cm. Therefore, it follows that L6=0.135 μH. Since a composite capacitance of C6 and C7 is 8 pF, the composite impedance of L6, C6 and C7 becomes −5i kΩ. Since this is a capacitive impedance, 8.005 pF is obtained by converting the capacitive impedance into a capacitance.
A composite capacitance of this and C9 becomes 103 pF+8.005 pF=111.005 pF. Since this composite capacitance and L5 (=0.135 μH) cause serial resonance, the resonant frequency (hereafter referred to as Reso_Measure) becomes 41.113 MHz according to the Expression 1.
Because of a voltage variation caused by the resonance in the voltage measurement circuit, there must be a definite relation between the frequency of the voltage to be measured and the resonant frequency.
In order to hold down the voltage measurement precision to ±5%, a frequency satisfying the relation V1/V3>0.95 is checked particularly in the graph representing the V1/V3 ratio shown in FIG. 12C. As a result, the frequency is 8.9 MHz or below. Denoting the frequency of the voltage to be measured by fB, therefore, it follows that Reso_Measure/fB>41.113/8.9=4.6.
Denoting an inductance and a capacitance that determine the resonant frequency of the voltage measurement circuit respectively by L and C, therefore, it is necessary to satisfy the following expression (6) on the basis of the Expression (1).
LC<(9.2 τfB)⁻² (6)
The coefficient 9.2 need not be this value necessarily. Since this coefficient depends on the voltage measurement precision, this coefficient should be determined with respect to a required measurement precision by using simulation or actual measurement. For example, if the measurement precision is set to +10% under the same condition as that in FIG. 12C, the frequency satisfying the relation V1/V3>0.90 becomes 12.6 MHz or below, and the coefficient becomes 6.5 (=41.113/12.6*2).
Phase measurement will now be described. If a rectifier circuit using a diode D1 included in the voltage measurement circuit shown in FIGS. 8 and 11 is replaced by a phase detection circuit, the voltage phase can be measured. Block diagrams corresponding to FIGS. 8 and 11 are shown in FIGS. 13 and 14, respectively. Results obtained by simulating phase differences of V1/V2 and V1/V3 in FIG. 14 under the same conditions as FIGS. 12A-12C are shown in FIGS. 15A and 15B, respectively. The phase difference between V1 and V2 exhibits complicated behavior. The phase difference between V1 and V3 suddenly changes from 0° to 180° at a resonant frequency of 41 MHz. This is because the phase detection circuit is formed of only an inductance and a capacitance without using resistances. If resistances are used, then unadvantageously the phase difference exhibits a comparatively gently-sloping change. It is appreciated from this result that there are no problems as regards the phase measurement as long as the restriction represented by the Expression 6 is observed.
The circuit concerning the voltage measurement and phase measurement heretofore described can be applied to not only the electrode having a wafer mounted thereon, but also all electrodes capacitance-coupled to plasma. This embodiment will now be described.
FIG. 16 is a longitudinal section diagram of an etching chamber used in the present invention. FIG. 16 differs from FIG. 10 in that not only the high frequency power supply for plasma generation (the first high frequency power supply) 108 (having a frequency of, for example, 200 MHz) is connected to the antenna electrode 103 via the matcher 109 but also an antenna bias power supply 113 which is a third high frequency power supply is connected to the antenna electrode 103 via a matcher 112. The antenna bias power supply 113 and the wafer bias power supply 119 are connected to a phase controller 120. As a result, phases of the high frequencies output from the antenna bias power supply 113 and the wafer bias power supply 119 can be controlled. In this case, the antenna bias power supply 113 and the wafer bias power supply 119 are made to have the same frequency (for example, 4 MHz). In this system, a difference in phase (for example, 180°) between the antenna biasing high frequency appearing on the antenna electrode 103 and the wafer biasing high frequency appearing on the wafer 116 is controlled, and a bias can be applied to each of the antenna electrode 103 and the wafer 116 effectively. For implementing such a system, the voltage and phase at the electrostatic chuck electrode 124 are detected by pulling out the voltage to the atmospheric pressure side by the use of the coaxial cable 157 and providing a phase measurement circuit 155. In order to detect the voltage and phase at the upper antenna electrode 103, the voltage at the antenna electrode 103 is taken out to the atmospheric pressure side by using a coaxial cable 159 and a phase measurement circuit 156 is provided, in the same way as the lower electrode. A phase controller 120 compares phases obtained from the two phase measurement circuits 155 and 156, and determines a phase difference in high frequencies to be sent to the antenna bias power supply 113 and the wafer bias power supply 119 so as to generate a predetermined phase difference.
In order to raise the reliability of the control, the matcher 109 incorporates a filter 110 for cutting off the frequency of the antenna bias power supply 113. In the same way, the matcher 112 incorporates a filter 121 for cutting off the frequency of the high frequency power supply 108 for plasma generation. Outputs of the two matchers 109 and 112 are combined by using a coaxial cable 158, and a resultant signal is coupled to the coaxial line 111 which is the high frequency transmission system for the antenna electrode.
It should be further understood by those skilled in the art that although the foregoing description has been made on embodiments of the invention, the invention is not limited thereto and various changes and modifications may be made without departing from the spirit of the invention and the scope of the appended claims.

Claims

1. A plasma processing apparatus including a vacuum vessel, a lower electrode provided in the vacuum vessel to place a sample thereon, a matcher connected to the lower electrode, and a power supply for supplying power to the lower electrode via the matcher, the plasma processing apparatus comprising:

an electrostatic chuck electrode provided within the lower electrode to hold the sample; and

a voltage measurement circuit provided within the lower electrode to measure a voltage at said electrostatic chuck electrode and output the measured voltage as a DC voltage.

2. The plasma processing apparatus according to claim 1, wherein said voltage measurement circuit is installed within a vessel that intercepts at least a corrosive gas.

3. The plasma processing apparatus according to claim 1, wherein said voltage measurement circuit can detect a phase signal.

4. A plasma processing apparatus including a vacuum vessel, a lower electrode provided in the vacuum vessel so as to incorporate an electrostatic chuck electrode for holding a sample, a matcher connected to the lower electrode, and a power supply for supplying power to the lower electrode via the matcher, the plasma processing apparatus comprising:

a voltage measurement circuit provided under atmospheric pressure to measure a voltage at the electrostatic chuck electrode and output the measured voltage as a DC voltage; and

a coaxial line for connecting the electrostatic chuck electrode to said voltage measurement circuit.

5. The plasma processing apparatus according to claim 4, wherein a composite impedance of said voltage measurement circuit and said coaxial line is greater than a load impedance between the electrostatic chuck electrode and plasma.

6. The plasma processing apparatus according to claim 4, wherein said voltage measurement circuit can detect a phase signal.

7. A plasma processing apparatus including a vacuum vessel, a lower electrode provided in the vacuum vessel so as to incorporate an electrostatic chuck electrode for holding a sample, an upper electrode provided in a position opposed to the lower electrode, a first matcher connected to the lower electrode, a first power supply for supplying power to the lower electrode via the first matcher, a second matcher connected to the upper electrode, and a second power supply for supplying power to the upper electrode via the second matcher, the plasma processing apparatus comprising:

a first phase measurement circuit provided under atmospheric pressure to measure a phase of a voltage applied to the electrostatic chuck electrode;

a first coaxial line for connecting the electrostatic chuck electrode to said first phase measurement circuit;

a second phase measurement circuit provided under atmospheric pressure to measure a phase of a voltage applied to the upper electrode;

a second coaxial line for connecting the upper electrode to said second phase measurement circuit; and

a controller for controlling the first power supply and the second power supply based on output signals of said first phase measurement circuit and said second phase measurement circuit.