US 2092709 A Abstract available in Claims available in Description (OCR text may contain errors) Sept. 7, 1937. H. A. WHEELER BAND PASS FILTER 2 Sheets-Sheet 1 %REOE|VR Filed Nov 19, 1935 BAND PASS FILTER S 1937- H. A. wHEELEE 2,092,709 ' I BAND PASS FILTER I Filed Nov. 19, 1935. 2 Sheets-Sheet 2 F|e.5u. 33 FIG.60. Fl,G.6b g RL g RL h m o f1 f2 f3 5 n f2 f3 w FlG.6c. Q need; fl -f2 f3 E f1 f2 fa FIG. 66. g I INVENTOR. o V E. v HAROLD A. WHEELER. ATTORNEY. Patented Sept. 7, 193 7 UNITED STATES BAND-PASS rim-En Harold A. Wheeler, Great Neck, N. 1., r to Hazeltine Corporation, a corporation of Delaware Application November 19, 1935, Serial No. 50,503 7 Claims. ' This invention relates to band-pass filters and more particularly to composite filters including a plurality of individual band-pass filters cooperating to pass a more extended band than any of the filters separately. While this invention is of general application, it is especially suitable for coupling an antenna, adapted to receive or transmit a wide band or a plurality of bands of the radio-frequency spectrum, to a signal-translating circuit. In many installations, particularly in radiofrequency receiving and transmitting circuits, it is desired to pass a wideband of frequencies, or selectively to pass any, of a plurality of frequency bandsaggregating a wide portion of the radiofrequency spectrum. However, certain difliculties are presented in the'de'sign of a band-pass filter operating between extreme frequency limits, both as to the complexity or number of filter elements required and the actual circuit design as well as to the procurement of reasonably uniform responsiveness over the band. In my United States Patent No. 2,064,775 granted December 15, 1936, on an application, Serial No. 25,736, filed June 10, 1935, there is disclosed a composite band-pass filter for interconnecting an antenna through a transmission line to a radio receiver, the connections being effective to provide for operation of the antenna as a doublet over the high-frequency band and as a simple fiat-top antenna over the low-frequency band. Thepresent invention is particularly use ful in such a system and constitutes essentially improvements on the band-pass filter disclosed therein, enabling the use of a substantially shorter doublet antenna and a substantial reduction in the number of circuit elements of the component filters, at the same time procuring improved performance. The circuit elements in this application are given the same reference numerals as in the copending application, in so far as they correspond. It is an ,object of my invention, therefore, to provide a composite band-pass filter capable of passing a wide band of frequencies, which will overcome the above-mentioned difliculties of the arrangements of the prior art, and which will require a minimum of circuit elements. More specifically, it is an object of my invention to provide a composite band-pass filter capable of covering a wide band of frequencies and including a plurality of co-operating bandan over-all characteristic exhibiting a substantially uniform responsiveness over the entire frequency band. In accordance with my invention, there are provided two or more component band-pass filters designed to pass relatively wide contiguous frequency bands 15-}: and fz-fs, respectively. Each component filter comprises at least two half-sections of band pass filters of particular types which may be combined to form a com posite filter requiring a minimum number of reactance elements. The terminal sections of at least one end of the filters are of such a type and are so proportioned that they may be directly interconnected, as by a series connection, without substantially affecting the operation of either filter over its respective band. 'Ihe terminal sections of the other end of the filters are of a type posite hand-pass filter according to the present invention; Fig. 2 is a modified form of the lowfrequency hand-pass filter component; Figs. 3a,-3d are circuit diagrams illustrating the circuit transformations in developing the high band filter component; Figs. la-4d are corresponding diagrams for the low band filter component; Figs. 5H8 are corresponding diagrams for the modified low hand filter component of Fig. 2; while Figs. fine-Be are graphs representing image impedance characteristics on the line side of the. filter networks of Figs. 1-5, inclusive. Referring now more particularly to Fig. 1, there is shown schematically a wave-signal-collecting system, to which this invention is particularly applicable and inwhich the invention is embodied as a composite filter for coupling an antenna, for operation over an extended frequency band or a plurality of individual frequency bands, to a signal-translating or load device, such as a radio receiver, either directly or indirectly, by means of a transmission line. The general system of Fig. 1 is disclosed and claimed in my United States Patent No. 2,064,774 granted December 15, 1936, on an application, Serial No. 25,735, filed June 10, 1935; so that a detailed description thereof is considered unnecessary. In general, the system includes an antenna Illa, lllb preferably designed as a doublet for balanced operation in the high-frequency band fzfa but including connections for operation as a fiat-top antenna over the low-frequency band )1-)2. The antenna Illa, lob is coupled to a transmission line l2 by a high-band filter Ila for balanced operation of both the doublet antenna and the line over the high-frequency band f2f3, which may, for example, extend from 5 to 18 megacycles. Similarly, the low band filter I lb couples the antenna, for unbalanced operation as a fiat-top antenna, with the balanced line over a low-frequency band f1-fz, for example, 055 to 5 megacycles. The term "balanced" is used herein in its usual sense to denote that the circuit or device so described is electrically symmetrical with respect to ground. The two filters are uncoupled on the antenna side because of the balanced and unbalanced operation of the antenna over the high and low-frequency bands, respectively. They are joined together on the line side and their operation is interdependent at frequencies in the neighborhood of the dividing frequency, for example, 5 megacycles. The two filters Ho, Ho are designed independently and then combined, as will be explained more fully hereinafter. It will be noted that both the high band and low band are relatively wide bands; that is, the ratio of upper to lower cut-off frequencies is substantially greater than unity, being 3.6 in the case of the high band and 9.1 in the case of the low band, for the boundary frequencies given. The other end of the line l2 is coupled by a band-pass filter l3, which may be of any suitable type, to a signal-translating or load device l4 having an input circuit the impedance of which is represented at IS. A composite extended band filter particularly suitable for this purpose is disclosed and claimed in United States Letters Patent No. 2,081,861, granted May 25, 1937. It is desired that the compositefilter shall match the impedance of the doublet antenna to the constant image impedance of the line over the high-frequency band and that it shall connect the antenna as a flat-top antenna and match its impedance to that of the line over the lowfrequency band. It is also desirable that the composite filter circuit shall include a transformer section which avoids direct connections between the primary and secondary circuits of the filter and permits impedance transformation. It has become common practice in the design of filters to make certain of the computations on the basis of half-sections of standard types, whose design formulas are available. It is also customary to base the preliminary computations on an arbitrary value for the image impedance at each junction of the sections or half-sections. For the constant-k type, generally used as a standard of reference, the mid-series and midshunt image impedances of band-pass half-sections have the same value at the geometric mean frequency of the band. This value is indicated by the symbol R, which may be assumed to have a value of 100 ohms, for example, for the purposes of computation. (For a more complete description of the several types of band-pass filter halfsections utilized in the preferred embodiment of this invention and discussed herein, reference s made to a textbook by T. E. Shea: Transmission Networks and Wave Filters, D. Van Nostrand Co., 1929.) The development of the high band filter is shown in Figs. 3a3d. In Fig. 3a is illustrated at the left-hand side a filter half-section having the constant-k characteristics referred to above and hereinafter designated as a type A filter halfsection. (This type is referred to at page 315 of the Shea reference as type IVK.) Such a type A half-section comprises a mid-series condenser l6 and inductance l1 and mid-shunt condenser l8 and inductance l9. In computing the circult constants of the section A, it will be assumed that it is to be designed for an arbitrary value of R at both input and output terminals. The values of the circuit reactances may then be computed, in terms of R and the boundary frequencies f2 and f: of the pass band, from the formulas given by Shea for the IVK type on page 316. These formulas are for the so-called fullseries" and full-shunt" arms and must be modified in the well-known manner for computation of the mid-series or mid-shunt" arms used to form a half-section. In order to connect adjacent band filters together at one end to form a composite filter, it is desirable to include at that end of each component filter a half-section terminated in a midseries reactance arm including a parallel-resonant circuit for which may be substituted reactance elements of the other adjacent band filter. There is represented at B, Fig. 3a, a type of half-section filter by which these characteristics can be procured. The type B filter halfsection includes mid-series inductance 'lll, midseries parallel-connected condenser 20 and inductance 2|, and mid-shunt condenser 22 and inductance 23. (The type B filter section just described is referred to at' page 318 of the Shea reference as type V3.) In this type of filter, for the purposes herein described, the values of the mid-series elements 20, 2| are not critical and these elements can be replaced by reactance elements'forming a part of, and critically proportioned for, a band filter designed for an adjacent band, without substantially afiecting the. operation of the filter. The filter half-section B has at its left-hand terminals a constant-k midshunt image impedance similar in form to that at the right-hand end of the filter half-section A, so that if these two image impedances are made equal, the two adjacent ends may be directly interconnected. The circuit constants of the type B filter section may be computed, assuming for R the same value as for the half-section A, in terms of R and the boundary frequencies f2 and is of the pass band, by using the formulas given by Shea for type V3 half-sections. By the use of well-known equivalent circuit transformations, the half-sections A and B of Fig. 3a can be combined into the reactance network of Fig. 3d. For example, the adjacent terminals of sections A and B may be interconnected, the condensers l8 and 22 combined into a single condenser. 25 and the inductances l9 and 23 into a single inductance 24, since these elements are all connected in parallel. This transformation is shown in Fig. 31). Similarly, it is well known that the inverted-L connection of inductances l0 and 24 of Fig. 3b is the equivalent of a transformer in which the inductances l0 and 24 provide the self-inductance of the secondary circuit and the inductance 24 provides the mutll l. inductance and the self-inductance of the primary circuit, in this case the two latter inductances being of equal value. The result of this transformation is the circuit of Fig. 3c, in which the transformer comprising inductances 21 and by the ratio which determines the nominal impedance Rn of the doublet antenna in terms of the effective inductance Lee of the doublet antenna, the equivalent reactance network of which is illustrated in Fig. 3d. Similarly, in order to match the secondary circuit to the line if it is necessary to multiply the circuit reactances of the secondary circuit by the ratio of, the image impedance of the line H, Rn, to the assumed image impedance R. The filter section of Fig. 3c is then efiective approximately to match the nominal impedance of the antenna to the image impedance of the line over the frequency band for which it is designed, for example, from -18 megacycles. The midserles-termination 28, 29 of Fig. 3c is resonant at a frequency somewhat below the lower cut-off frequency of the band f2--f3; that is, in the low band f1-f2. given by Shea for the types NE and V3 in conjunction with the circuit transformations indicated above. The circuit of Fig. 3c is rearranged as shown in Fig. 3:1, for balanced operation, by giving to the inductances' 21a and 2Tb a combined value equal to that of the inductance 2i and, similarly, by dividing the inductance 3| into two portions represented by the elements 3m and 3"). In gen-, eral, the inductanccs 21a, Zl'b will not each have a value half that of the inductance 21, nor the inductances 3m, 3") half that of the inductance 3|, because of the mutual inductance between' 69b and 30a. form an electrical network equivalent in reactance to a. doublet antenna having a fundamental frequency equal to the geometric mean frequency of the high band, and are replaced thereby. The equivalence of this network determines L69 which is the effective inductance of the doublet antenna. In making this substitution the left-hand terminals of Fig. 3c are effectively short-circnited except for the relatively small resistance of the doublet antenna. The fundamental frequency of the doublet antenna is determined mainly by its length, which is so selected that the fundamental frequency thereof is approximately equal to the geometric mean of the cut-off frequencies of the high band filter. The doublet antenna so designed has a minimum value of impedance at the geometric mean of the cut-off frequencies 12, is of the high band and has a substantially greater value of capacitive reactance at the lower cut-off frequency f2. The input circuit 21a, 21b, 30b of the high band filter, designed as described, has maximum impedance approximately at the geometric mean frequency and has at the lower cut-off frequency an induc- The formulas for the circuit constants of Fig. 30' may be derived from the formulas to the antenna reactance. Similarly, the component band-pass filter lib, for operation over the low-frequency band f1f2, for example, 0.55-5 megacycles, may be designed to couple the antenna Illa, b, operating as a simple, unbalanced, fiat-top antenna, to the balanced transmission line [2 and to match the impedances of the antenna and line over the lowfrequency band. This filter is required to operate over the entire low-frequency band, which is substantially contiguous to the high-frequency band. At the same time, it is desired to prevent unbalanced coupling through the low band filter at frequencies within the high band. Such coupling would permit unbalanced noise picked up on the antenna at such frequencies to be couplcd to the'transmission line as balanced current. Therefore, it is desirable to provide means for securing great attenuation in the low band filter at frequencies within the high band. The design of the component low band filter is approached from the same standpoint as that of the high band filter described above. Referring to Fig. 4a, the starting point in thisinstance is the selection of a. standard type filter halfsection including as a mid-series element a parallel-resonant circuit, or trap, resonant at a frequency in the lower portion of the high band and contributing to the attenuation in the low band filter at frequencies in the high band. The type C filter of Fig. 4c (referred to at page 318 of the Shea reference as type V4) has been found suitable for this purpose. The type C filter halfsection comprises a mid-series condenser 32 and a mid-series trap circuit comprising inductance 33 and condenser ll, connected in parallel, and mid-shunt condenser 34 and inductance 35. The formulas for computing the circuit constants of such a filter section in terms of the nominal iniage impedance R and the boundary frequencies f1 and f2 are given by Shea at page 318. In the case of the right-hand filter half-section of Fig. 4a., a type D filter half-section (described by Shea at page 317 as type 1V3) has been found particularly suitable. This filter half-section comprises mid-series parallel-connected condenser 36 and inductance 31 and mid-shunt parallelconnected condenser 38 and inductance 39. The formulas for computing the circuit constants of the type D filter section in terms of the same parameters are given at page 317 of Shea. The final provision of a transformer requires the addition of a filter section such as a type E filter section of Fig. 4a. The filter half-sections C and D both have constant-k mid-shunt image impedances and can, therefore, be properly joined by the symmetrical section E of mid-shunt termination. This section comprises a mid-shunt condenser 40 and inductance 4| a full-series inductance 42, and a mid-shunt condenser 43 and tive reactance approximately equal in magnitude inductance M. The formulas for computing the circuit constants of the type E section of Fig. 4c, in terms of the parameters R, 11 and f2, are given at page 316 of the Shea reference wherein this filter section is identified as type IIIz. The components C, D and E of Fig. 4a can be merged into their electrical equivalents, as indicated in Figs. 4b, 4c and 4d. In Fig. 4b the midshunt condensers 34 and All are combined into the single condenser 45, the mid-shunt induct- ,ances 35 and 4! into the inductance 48, the mid shunt condensers 38 and 43 into the condenser 41, and the mid-shunt inductances 39 and 44 into the inductance 48. It is seen that the inductances 46, 42 and 48 comprise a pi-section which may be replaced by an equivalent transformer. Such a transformation is shown in Fig. 4c, in which these inductances are converted into a transformer comprising inductances 50 and 52. The transformer 50, 52 is effective to match the impedances of the antenna and the line over the low-frequency band 11-42. The circuit of Fig. 4c is also modified in that the values of the reactance elements of its primary circuit are multiplied by such a factor that the condenser 53 has a capacitance equal to that of the fiat-top antenna, effective at the lowest frequency ii. The nominal value of the antenna impedance and the image impedance RE of the antenna end of the filter of Fig. 4c is then determined by multiplying the assumed nominal image impedance R by the ratio of the capacitance of the condenser 53 to that of condenser 32. Similiarly, the values of the reactance elements of the secondary circuit are multiplied by the ratio of the image impedance Rr. of the line to the assumed nominal image impedance R of the half-section D. This circuit of Fig. 4c is modified to that of Fig. 401 in order that the secondary circuit may operate into the balanced line. To this end the mid-series condenser 55 and inductance 56 are divided into equalparts represented, respectively, by the condensers 55a and 55b and the inductances 56a and 56b of Fig. 4d. These latter midseries terminations 55a, 56a and 55b, 5612 are resonant somewhat above the upper cut-off frequency of the low band fifz; that is, in the high band ,f2f3. Fig. 4d shows also a modification of the primary circuit of Fig. 4c in which the midseries condenser 53 is connected across the input terminals and represents the value of the antenna capacitance effective at the lowest frequency f1, so that the antenna may be substituted for this condenser. The high band and low band filters of Figs. 3d and 4d, respectively, are combined in Fig. 1 to form a composite filter covering the entire band from h to f3. I'he input terminals of the high band filter are connected directly to the terminals of the doublet antenna, while the input terminals of the low band filter are connected respectively to the mid-tap between the inductances 27a and 27b and the ground connection 59 to be described presently. The primary circuits are unchanged, excepting only for the substitution for the doublet antenna and the fiat-top antenna for the equivalent circuit elements of Figs. 3d and M, respectively. The secondary circuits are combined in a particular manner, the midseries elements 28 and 29 of Fig. 3d being replaced by the elements 5|, 52a and 52b,. respectively, of the low band filter, while the mid-series elements 55a, 56a, 55b and 56b of the low band filter are replaced by the elements 3|a and 3|b of the high band filter. In other words, each filter circuit proper forms a mid-series reactance arm for the other filter. While the circuit constants of each of the filters may not be ideal for terminating the other, the values of these terminating reactances are not critical so that these constants may be selected primarily to satisfy the design requirements of their respective filter sections and the responses ofthe filters over their respective bands will be satisfactory, while the cut-off frequencies will be unaltered. Undesired noise at frequencies in the low band may be minimized by the use of a quiet ground connection in the neighborhood of the antenna to a point which is relatively free from coupling to sources of noise. The doublet antenna is preferably located high above the ground and the antenna filters Ila, ||b located at, and connected directly to, the doublet terminals. This requires the use of a relatively long ground lead 59 to a ground connection 6|. Such a long lead, if grounded directly, has a relatively high impedance at certain frequencies, especially at the frequency corresponding to a wavelength four times the length of the ground lead. Therefore, the ground lead may be made more effective if its impedance is made substantially uniform by the use of a damping resistor '80 of about 500 ohms inserted at the ground end of the lead. A similar expedient comprising a ground lead 62 connected to the junction between inductances 52a and 52b, a damping resistor 53 and a similar ground connection 64 may be used advantageously for grounding the line through the antenna filter. The image impedance characteristic of the type B filter section of Fig. 3a is shown in Fig. 6a.. The right-hand portion of this characteristic, near the upper cut-off frequency 13, has the gradual curvature characteristic of a constant-k mid-shunt image impedance. The curvature at the lower'cut-off frequency f2, however, is considerably more abrupt than that of a constant-k section and is characteristic of the mid-series termination of the type B filter. Similarly, the image impedance characteristic of the type D section of Fig, 4a is shown in Fig. 61). It is seen that these two filter characteristics are substantially complementary for the entire band ji-fi. The effect of connecting together the high and low band filters at the terminals of the line I2, as shown in Fig. 1, is to merge the image impedance curves of the two filters into a single curve substantially continuous over the band 11-;3, as shown in Fig. 6c. The latter characteristic has substantially the same shape as the mid-series image impedance characteristic of a constant-k continuous band filter designed to pass the band f1f3 and is nearly uniform over the entire frequency range of the high and low bands, its departure from uniformity at the extreme frequencies not being a serious disadvantage. It is believed that the general principles of operation of the above-described system will be clear from the foregoing detailed description of the circuit arrangement and the principles involved in its design. However, the operation in the two frequency bands may be summarized briefly as follows: At the frequencies of the high band, the balanced operation of the doublet antenna causes no desired balanced currents to be coupled to the low band filter from the center tap between 21a and 21b. Undesired unbalanced currents at the same frequencies are attenuated in the low band filter by the impedance of the trap 51, 58 and the by-passing effect of condensers 49, 5|. This attenuation is so great that the unbalancing effect of these currents isminimized. Similarly, at the frequencies of the low band the impedance of the elements 21a and 21b effectively in parallel is so low that it may be neglected, resulting in the equivalent of a direct connection from a flat-top antenna to the input circuit of the low band filter. At the frequencies near the boundary between the high and low bands, both filters contribute to the action of the composite filter. Over the high-frequency band the antenna lfla, |||b operates as a balanced doublet and the high band filter Ila is effective to couple the balanced antenna currents to the balanced line I2 as balanced circulating currents therein, at the same time approximately matching the impedance of the doublet antenna with that of the line over the high-frequency band 12-43. The circulating currents in the line l2 are coupled by the filter l3 to the input circuit l5 of the receiver l4. When operating over the low-frequency band the low band filter llb serves to couple the unbalanced currents of the antenna Illa, lnb,,operating as a simple fiat-top antenna, as balanced circulating currents in the line l2. The filter l3, similarly, couples the balanced circulating currents of the line l2 to the input circuit l5 of the receiver 84. The modified form of the low band filter shown in Fig. 2 is designed to take advantage of two added condensersand a comminuted iron core in the coupling transformer, thus increasing the coefficient of coupling and facilitating the reduction of the lower cut-off frequency of the low band f1.-fa, which, in this case, may extend from 0.15 to 5 megacycles. The development of the low band filter of Fig. 2 is shown in Figs. 5a. to 5e, inclusive. The method of attack is as described above. Starting, as in Fig. 4a, with the left-hand input half-section of type C (type V4 of Shea), a type C is also chosen for the output filter halfsection in this case and a type -E transformer section (type III: of Shea) is utilized to interconnect the input and output.sections to allow the insertion finally of a coupling transformer. The filter half-sections C, E and C of Fig. 5a are, therefore, of the same :types and. comprise the same elements as the corresponding types of Fig. 4a and the elements are identified by the same reference characters. The transformation from Fig. 5a to Fig. 5b is similar to that from Fig. 4a to Fig. 4b and involves merely the combination of the condensers 34 and 40 of the input side and 34 and 43 of the output side to the condensers 45, and of the inductances and 4i of the input side and 35 and 44 of the output side to the inductances 46. The transformation of the network of Fig. 5b to that of Fig. 5c involves interchanging the order of the series condenser 32 and the shunt condenser 45 on the output side. This transformation may be effected by proper proportioning of the values of these condensers to the values of the condensers l5 and 16, respectively, and properly multiplying all of the impedance values to the right thereof by a constant of the transformation, without otherwise affecting the filter characteristics. (This type of transformation is illustrated and quanti-' tatively analyzed at page 137 of the Shea reference with respect to Fig. 68). The ultimate purpose of this transformation is to reduce the total section 46, 42, 46 by the transformer 68, I9 and at the same time multiplying the impedance values of, each side by an impedance ratio. All impedances on the right side are multiplied by the ratio of the image impedance R1. of the line to the nominal image impedance Rc. All im-' ratio pedances on the left side are multiplied by the depending on the effective capacitance 8| of the antenna to ground at the lower cut-off frequency of the low band, which also determines the nominal image impedance RF in Fig. 5d. Fig. 5e constitutes merely a rearrangement of Fig. 5d in which the condenser BI is shown as representing the effective capacitance of the antenna to ground across the input terminals. In this latter case, also, the elements l8, i9, 85 and 86 are divided into parts of equal reactance and the condenser 11 is interposed between the parts of the inductances 19a, 19b, providing a balanced circuit for connection to the balanced line I 2. Considering the mid-series terminatin ances 85a, 85b, 86a and 8% as part of he high band filter and the condenser 8| as replaced by the antenna capacitance, as in the preceding embodiment of the invention, it is seen that the circuit of Fig. 5c is the same as that of Fig. 2. The transformer comprising the inductances 68, 19a. and 19b is preferably provided with a comminuted iron core 80 which, with the series condenser contributes to lowering the lower cut-oif frequency of the filter, enabling it to cover a wider frequency band. The ground lead 62 on the line side of the filter is connected to the center 3d to forma composite filter asin Fig. 1, the resultant image impedance characteristic of the composite filter is as represented in Fig. 6c. the same shape as that of a constant-k continuous-band filter. with mid-series termination, Therefore, this combination has particular utility imped- .This latter characteristic curve has substantially in cases where it is desired to operate contiguous band filters into a continuous band filter terminated in such a way that its image impedance has this simple form. The general. principles of operation of the modified form of low band filter shown in Fig. 2 are substantially the same as those of the low band filter lib of Fig. l and need not be reconsidered. While the filter sections of this invention have been described with reference to primary or input circuits and secondary or output circuits, it will be clear that these designations are by way of illustration only and that power may be transferred through the-filters in either direction so that either end may be considered as the input or output circuit. The invention described above is of general application and may be designed. for operation over a wide range of conditions. In general, it is preferable to follow the indicated formulas for computing filter sections as given by the reference Shea, above, in the preliminary design of the filter network, and to modify this design as necessary to obtain the exact operating characteristics and circuit configuration desired. The following circuit values are given for an antenna filter suitable for interconnecting a particular antenna and a particular line. The antenna is the double-V doublet 15 meters long with the outer-ends spaced 2 meters and suspended at a height of 10 meters from the ground. This type of doublet is described in the above-mentioned Patent No. 2,064,774. The line is a twisted pair of rubbercovered conductors. The following circuit values were realized as closely as possible and include such effects as inherent capacitance or inductance of other related circuit elements and wiring. Fig. 1 (11a) and Figs. Fig. 1 (lit) and Figs. Fig. 2 (11c) and Figs. 3c, 3d 4c, 4d 5d, 56 Cu 45 141 .1 on 40 pp! Cu 190 mi! Can 47 pp! Cu 470 up! Cu 7 mil Cau- 10 pp! C5; 190 up! C61 12 up! C30 37 up! Can 27ml! C11 7000 up! kimi w, n= 0-8 1: 450 #1 L 6.7 ph Lao uh Imm 0.94 Ln -0 1 Li: 27 es 100 u Lug 6.8 #11 L51 I 26 ph Ln 2400 ph Rn =400 ohms R; =l,500 ohms Ln 07 uh R1, =ll ohms R 115 ohms R1: 4600 ohms j. 5 Me {1 I 0.56 Mc Rx. F 115 Ohms [a 18 Mo 1': 5 Mo [1 0.15 Me While I have described what I at present consider the preferred embodiments'of my invention, it will be obvious to those skilled in the art that various changes and modifications may be made therein without departing from my invention, and I, therefore, aim in the appended claims to cover all such changes and modifications as fall within the true spirit and scope of my invention. What is claimed is: 1. In a wave-signal-translating system for signals of any frequency within two substantially contiguous frequency bands and including a. pair of devices, both of said devices being designed for balanced operation over one of said bands and at least one of said devices being designed for unbalanced operation over the other of said bands, a. composite band-pass filter for interconnecting said devices comprising a pair of individual band-pass filters proportioned to pass individually said contiguous bands, the one of said individual filters passing said other band including a transformer having an unbalanced winding for coupling to said one device and a balanced winding for coupling to the other device, and including also means providing maximum attenuation at a frequency in the pass band of the other filter for minimizing the unbalancing effect of said one filter at frequencies in the pass band of said other filter. 2. In a wave-signal-translating system for signals of any frequency within two substantially contiguous frequency bands and including a pair of devices, both of said devices being designed for balanced operation over one of said bands and at least one of said devices being designed for unbalanced operation over the other of said bands, a composite band-pass filter for interconnecting said devices comprising a. pair of individual bandpass filters proportioned to pass individually said contiguous bands, the one of said individual fil ters passing said other band including a transformer having an unbalanced winding for coupling to said one device and a balanced winding for coupling to the other device, and including also a trap circuit connected in series with said unbalanced winding and resonant at a frequency in the pass band of the other filter for minimizing the unbalancing effect of said one filter at frequencies in the pass band of the other filter. 3. In a wave-signal-translating system for signals of any frequency within two substantially contiguous frequency bands and including a pair of devices designed, respectively, for balanced operation over both of said bands and for balanced operation over the higher of said bands and unbalanced operation over the lower of said bands, a composite band-pass filter for interconnecting said devices comprising a pair of individual band-pass filters proportioned to pass individually said contiguous bands, the lower band filter including a transformer having balanced and unbalanced windings for coupling respectively to said devices, capacitance means connected across each of said transformer windings for greatly attenuating frequencies in said higher frequency band, and additional means providing maximum attenuation at a frequency in the higher band for minimizing the unbalancing effect of said lower band filter at frequencies in the higher band. 4. A composite band-pass filter comprising two individual band-pass filters passing two substan-' tially contiguous frequency bands and connected at one end to a common terminal circuit, the higher band and lower band filters being terminated at said end in filter half-sections respectively of types B and D, as defined herein, each of said half-sections having at its terminal end end a mid-series termination normally including in series a parallel-resonant circuit resonant at a frequency in the band of the other filter, each of said parallel-resonant circuits being replaced by reactance elements of the other filter, and said filters being relatively proportioned to present across said terminal circuit a resultant image impedance approximating the mid-series image impedance of a constant-k continuous-band filter passing the band comprising said contiguous bands. 5. A composite band-pass filter comprising two individual band-pass filters passing respectively two substantially contiguous frequency bands and connected at one end to a common terminal circuit, said filters individually including transformers with windings toward said common terminal circuit connected in series thereacross, the lower band filter including capacitance means effectively in series and capacitance means effectively in parallel with its respective corresponding transformer winding, and said filters being relatively proportioned to present across said terminal circuit a resultant image impedance approximating the mid-series image impedance of a constant-k continuous-band filter passing the band comprising said contigous bands. 6. In a wave-signal-translating system for signals of any frequency within a given band bounded by upper and lower cut-off frequencies having a ratio substantially greater than unity and including an antenna having a minimum value of impedance at approximately the geometric mean frequency of said band and a substantially greater value of capacitive reactance at said lower cut-off frequency and a signal-translating device operative over said band, a band-pass filter adapted to pass said band for interconnecting said antenna and said device, said filter including a terminal circuit for connection with said antenna and having at approximately said mean frequency a maximum impedance and at said lower cut-off frequency an inductive reactance of the same order of magnitude as the antenna reactance. 7. In a wave-signal-translating system for signals of any frequency within a given band bounded by upper and lower cut-off frequencies having a ratio substantially greater than unity and including an antenna having a minimum value impedance at approximately the geometric mean frequency of said band and a substantially mm adapted'tc passsaid band and having terminals for connection respectively to said antenna and said device; said filter including a transformer having one winding connected beproportioned to present across the antenna terminals at approximately said mean frequency a maximum impedance and at said lower cut oft frequency an inductive reactance of the same order of magnitude as the antenna reactance. 5 " HAROLD A. WHEELER. Referenced by
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