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Publication numberUS2986707 A
Publication typeGrant
Publication dateMay 30, 1961
Filing dateJul 13, 1959
Priority dateJul 13, 1959
Publication numberUS 2986707 A, US 2986707A, US-A-2986707, US2986707 A, US2986707A
InventorsBlecher Franklin H
Original AssigneeBell Telephone Labor Inc
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Prevention of overload instability in conditionally stable circuits
US 2986707 A
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Description  (OCR text may contain errors)

y 30, 1961 F. H. BLECHER 2,986,707

PREVENTION OF OVERLOAD INSTABILITY IN CONDITIONALLY STABLE CIRCUITS Filed July 13, 1959 3 Sheets-Sheet 1 FIG.

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PREVENTION OF OVERLOAD INSTABILITY IN CONDITIONALLY STABLE CIRCUITS Filed July 13, 1959 3 Sheets-Sheet 3 r0 TRIGGEA 5% if: r 22 Es cmcu/r 48 l LL 2- Ll INPUT T0 TRIGGER lNl/ENTOR f. H. BL E CHE R BVKZCZHQA;

A TTOPNEV United States Patent PREVENTION OF OVERLOAD INSTABILITY IN CQNDITIONALLY STABLE CIRCUITS Franklin H. Blecher, Plainfield, 'N.J., assiguor to Bell Telephone Laboratories, Incorporated, New York, N.Y., a corporation of New York Filed July -13, 1959, Ser. No. 826,718

18 -Claims. (Cl. 330-86) laced system disclosed .in Patent No. 2,909,623 which issued to applicant on October 20, 1959.

The problem 'to whichthepresent invention is directed, viz., overload instability in multiple-loop feedback amplifiers, will be firmly established if, at the outset, we

consider it in connection with a particular system, say, that of Greendwood. If additional insight concerning the nature of the problem is desired, 'itis suggested'that reference be made to the landmark work of H. W. Bode, Network Analysis and Feedback Amplifier Design, pages 162-164, D. Van 'Nostrand Co., Inc, (1945); to an article by J. Oizumi and M. Kimura, entitled, Design of Conditionally Stable Feedback Systems, which appears at 'page 157 of Transactions of the Institute of Radio Engineers, volume (ft-.4, No. 3 (September 1957); and to an article entitled Transistor Multiple'LoopFeedback Amplifiers by 'F. H. Blec'her which appears in volume '13 of Proceedings of the National Electronics Conference, page 19 (1957).

For greater clarity and ease vof narration, the Greenwood system will be limited, as it is .in'his patent, to two stages having respective forward gains A and A two local positive feedback paths having respective feedback factors of 13 and and one negative over-all feedback path having a feedback factor of a The loop gains A 5 and A 5 are therefore positive, while the loop gain A A 3 is negative. Note, further, that both A 5 and A 6 are constrained to equal unity.

Since the amplifier has two positive feedback loops whose gain magnitudes are constrained to equal unity, it is conditionally stable. Conditionally stable feedback amplifiers may become unstable by such disturbances as variations of supply potentials, variations in load, and saturation of amplifying elements by large input signals which overload the amplifier. It is the last-named cause of instability that is denominated overload instability and to which the invention is vdirccted. If the negative feedback represented by the gain product A A B should be diminished ,or discontinued, then the amplifier will burst into oscillation. The negative feedback loop will, in fact, as Will later be explained, be effectively opened if an active element, contributing to the gain A should become overloaded. The amplifier will then go into oscillation for, while the gain product A A fl effectively will have been rendered impotent, the unity gain product A 9 will continue withfull vigor. Oscillation will also result an active element contributing to the gain A should Patented May 30, 1961 "ice 2 become overloaded, for then, the unity gain product A e, would continue unopposed by the product 13 13 13 It is therefore an object of the present invention to automatically disable positive feedback'in a multiple-loop feedback amplifier in the event that any amplifying element divulges the imminent prospect of becoming overloaded. 7

Reflection will show that if the prospective overload is not swiftly acted upon, the amplifier will go into oscillation notwithstanding the warning of impending overload. Accordingly, a further object of the invention is toensure fast-acting and effective response to the overload warning.

In accordance with the invention, the alternating-current output signal of the amplifier is detected and converted to a direct-current signal. The latter signal controls the state of a trigger circuit to which it is supplied. If this control signal is below a certain critical level, .the trigger circuit is vunafiiected and remains in its normal state. If, however, the critical level is surpassed, the trigger circuit will be flipped to its abnormal state. Provision is made in the detector circuit so that the critical level of the trigger circuit corresponds to a prescribed maximum output level of the amplifier, which imports imminent instability due to the rapidly approaching overload point of one of the amplifiers active elements. The tn'ggercircuit, in turn, controls switches which eifectively interrupt all regenerative propagation whenever the trigger circuit is flipped to its abnormal state.

Perhaps the most outstanding featureof .the invention is the utter simplicity by which "the perplexing problem of overload instability is overcome. The invention will doubtless be welcomed wherever extremely -reliable trans mission is desired, to say nothing of applications where reliability is the pre-eminent consideration: e. g., transoceanic cable systems, and the many military applications in which failures due to instability cannot be tolerated.

The following description, when read in conjunction with the drawings, will offer a more comprehensive ,understanding of the invention. 'In the drawings:

Fig. l-is a block schematic diagram of the tandem feedback amplifier, referred to above, and very generally, of a circuit, arranged in accordance .WithLthe invention, for preventing overload instability;

Fig. 2 illustrates, also very generally, application .of the principles of the invention to the interlaced feedback amplifier spoken of above;

Fig. 3 is a detailed circuit diagram ofa preferred. embodiment of the invention;

Figs. 4 and 5 show the manner in which the control switches may be connected in the tandem and interlaced feedback amplifiers, respectively; and

Figs. 6 and 7, respectively dealing with shunt and series control switches, concern alternative configurations of the control switches shown in Figs. 3 to 5, transistors being used in lieu of diodes.

Fig. 1 broadly shows, in :block schematic form, the detailed circuit of Fig. 4. As stated above, Fig. lis aso-called tandem multiple-loop feedback amplifier of the type disclosed in the Greenwood jpatent. The stages -14 and 16 have gains of A and A respectively. The amplifier has probably come to be known as .one of the tandem multiple-loop variety because of the successive local feedback paths, here represented by the 13 and fi circuits, which provide local feedback respectively around the stages 14 and 16. Over-all negative feedba k is prothe stabilizing loop gain A A B the amplifier of Fig. 1

would be normally unstable, ,i.e., the amplifier would continually oscillate. If the magnitude of the external feedback provided by 19 is large compared to the internal feedback provided by B, and ,6 the internal feedback does not affect the stability of the amplifier under normal conditions. Proof of this is given in an article entitled Design Principles for Single Loop Transistor Feedback Amplifiers, by F. H. Blecher which appears in Transactions of the Institute of Radio Engineers, volume Ct4 (September, 1957). The loop gain A A fi is therefore chosen to offset the regenerative loop gains A 5 and A 5 in order to render the amplifier stable under normal conditions of operation. But in view of the positive feedback loops and the possibility of abnormal operating conditions, the amplifier is only conditionally stable. Thus, should an active element in stage 14 become overloaded, that is to say, saturated, the negative feedback loop will be effectively opened and the amplifier will oscillate.

It will be instructive to elaborate upon the last statement. It should first be understood that the overloading of a stage vitally affects its amplifying capability. Its gain is, in fact, drastically reduced. This fact can be discerned from a consideration of the characteristic curves of a typical amplifying element. Since the active element, which was assumed to have become overloaded, contributes to the negative feedback represented by the loop gain A A fi the negative feedback will decrease as a consequence of the decrease in A due to the overload. The positive feedback represented by A 18 will also decrease. The positive feedback represented by A 5 will, however, continue unabated. The phase and gain margins of the amplifier will therefore be reduced and, since the Greenwood system ideally calls for an A 6 of unity, oscillation will result. The above analysis can also be used to show that oscillation will occur, as Well, if an active element in stage 16 should become overloaded.

But in accordance with the principles of the invention, whether or not active elements in stage 14, or 16, or in both stages become overloaded, the tandem multipleloop feedback amplifier will remain stable. To be sure, the advantages of positive feedback will not be available so long as the overload condition persists, but the amplifier will, nevertheless, continue operation as a stable, single loop, negative feedback amplifier. Stability is insured by the protection circuit consisting of detector 18 Schmitt trigger circuit 20, and the switches 22 and 24.

In Fig. l, as in Fig. 2, switch 22 is shown as a shunt switch, whereas switch 24 is shown to be of the series variety. The difference is only meant to show that alternate methods of switching may be used. Thus, the

switches 22 and 24 may both be of the same type or, if

desired, respectively of the series and shunt varieties. The protection circuit will be explored in much greater detail when the description of Fig. 3 is given.

Fig. 2 is a block diagram of an interlaced feedback amplifier of the type described in the above-cited copending application. For ideal performance, the loop gains A A fi and A A B are each constrained to equal unity. The 13 and 3 circuits are positive feedback paths. The unity gain constraint is advantageous in that the over-all gain of the amplifier is thereby rendered insensitive to changes in the gain A so long as the gains A and A have not varied, and to changes in the gain A so long as the gains A and A have not varied. To obtain this result, there are no restrictions on the normally-stabilizing loop gain A A A fi other than the practical requirement that the feedback be negative and that its absolute magnitude be greater than unity by an amount suflicient to render the system stable in the sense that it will not oscillate.

Another advantageous property of the interlaced amplifier shown in Fig. 2 is that output distortion introduced by the last stage A is drastically reduced if the loop gain A A 3 is constrained to equal unity. Moreover, this reduction of distortion is effectively independent of the loop gains A A fi and A A A p except that the magnitude of the latter gain, as mentioned above, should be greater than unity by an amount sufficient to render the system stable. Theoretically, it should be noted, the output distortion is completely eliminated, but it is impracticable to constrain A A 5 to exactly unity at all times. Nevertheless, reduction of output distortion in the interlaced feedback amplifier closely approaches this theoretical ideal.

The overload protection circuit of Fig. 2, shown very generally in block schematic form, is identical to that of Fig. 1. Again, it should be noted that it is not necessary to use the shunt and series switches as they are shown in the diagram. A shunt or a series switch may be used in conjunction with either feedback network, depending upon the specific design of the network.

It is well to note that the use of both positive and negative feedback in an amplifier enables a designer to obtain transmission characteristics that are not obtainable in amplifiers using only negative feedback. Such amplifiers, however, are only conditionally stable, as we have seen. Since a vacuum tube requires a substantial warm-up period, it is, in a very real sense, persona non grata so far as conditionally stable amplifiers are concerned. Positive feedback, therefore, is usually not employed in vac uum tube amplifiers; although it should be understood that it can be, if adequate provision is made for warmup time. Transistors, on the other hand, do not present this warm-up problem and are, consequently, particularly suitable for use in conditionally stable circuits. Thus, the reader will note that the feedback amplifiers and overload protection circuits presently under discussion employ transistors exclusively.

Fig. 3 shows in detail the overload protection circuits shown very generally in Figs. 1 and 2. Ultimately, as we have seen, the regenerative portions of Figs. 1 and 2 switch (change states) whenever an imminent overload condition is forecast. The method used to accomplish this end has been broadly treated above. The means for doing so will now be elaborated upon.

The overload protection circuit consists of a directcurrent detector 18, a Schmitt trigger circuit 20, and a switching circuit 26. The symbols E E E represent absolute magnitudes of direct-current voltages at various points in Fig. 3. Of these, -E --E and +E are power supply voltages; -E is the direct-current voltage developed at the output 34 of the direct-current detector; and E is the direct-current output voltage of the Schmitt circuit.

The direct-current detector circuit 18 is coupled through capacitor 36 to the output 12 of the multipleloop feedback amplifier of either Fig. 1 or Fig. 2. The detector circuit 18 consists of a level control, an emitter follower (otherwise known as a common collector stage), and a peak detector.

The level control is the potentiometer R It is adjusted so that the maximum alternating-current output voltage of, say, the amplifier of Fig. 2, after which an overload condition is imminent, is converted by the detector to the threshold or fiipping" voltage of the Schmitt circuit.

The emitter follower 38 is preferred for at least two reasons. First, it has a high input impedance and, consequently, the output of the feedback amplifier, to which it is connected, will not be loaded down. Second, and more important, it has a low output impedance which makes it possible for the peak detector, the diode D and capacitor 27, to respond to sudden increases in the level of the feedback amplifier output voltage. It is not sufficient that the Schmitt circuit 20 and the switching circuit 26 be fastacting, since a forecasted overload condition will have been in vain if detector circuit 18 cannot quickly convey this information to the Schmitt circuit 20. The rapid responsive capability of detector circuit 18 to abrupt increases in the feedback amplifier output level operation, i.e., two states of equilibrium, and will be in one or the other, .depending upon the magnitude of the detector output voltage ---E in its normal state, i.e., when the feedback amplifier has forecast no danger of imminent -.ov. erload, the Schmitt circuit has its first stage 40 nonconduetive and its second stage -42 conductive. Conversely, stages 40 and 42 are respectively rendered conductive and nonconductive when the output level of the feedback amplifier has informed the detector circuit 18 that an overload is immediately prospective and, in turn, voltage -E has been brought to the flipping level of Schmitt circuit 20. The amplifier output voltage at which an .overload is imminent thus corresponds to a specified level of E beyond which Schmitt circuit 20 changes state. So long as the magnitude of -E .does not fall below this level, the Schmitt circuit output voltage -E will be at a constant peak value substantially equal to .-E as we shall see.

When the Schmitt circuit 20 is in its normal state, then, its output voltage -E has arelatively small negative value since transistor 42 is conductive. In its abnormal state, however, Schmitt circuit 20 has a negative output voltage of relatively high value, since transistor 42 is then cut off (has zero collector current) and E is very nearly equal to the supply voltage E Thus, in the face of an imminent prospect of amplifier overload, the output voltage E of Schmitt circuit 20 will be relatively large, negatively, and very nearly equal to the supply voltage E The Schmitt circuit has been the subject of many papers, beginning with A Thermionic Trigger by O. H. Schmitt, which appeared in The Journal of Scientific Instruments, volume 15, page 24 (1938), published by the Institute of Physics, London, England. The vacuum-tube analog of trigger circuit 20, is shown at page 468 of Reference Data for Radio Engineers (4th ed. 1956).

In discussing switching circuit 26, it will be helpful to do so with reference, at appropriate times, to the p, and p networks of, say, Fig. 5. While Schmitt circuit 20 is in a state of normalcy-no overload in the feedback amplifieris imminentthe diode D of the shunt switch 22 is reverse-biased by the negative supply voltage -E It is apparent, therefore, that the voltage E is more negative than the Schmitt output voltage E when transistor 42 is conductive. The diode D has an extremely high impedance when it is reverse-biased, since, in practice presently at least-D is a silicon junction diode. Thus, the overload protection circuit has no effect on the makeup of the ,8 circuit while the multiple-loop feedback amplifier operates at normal signal levels. The 13 circuit of Fig. 5, for example, normally comprises two series resistors, R and R and two shunt resistors, R and R When, however, an overload condition is immediately prospective, diode D of shunt switch 22 is forwardbiased by the change-of-state of Schmitt circuit 20, since the voltage E now very nearly equal to E is greater, negatively, than the supply voltage E The shunt resistance in the [-3 network now includes, in addition to resistors R and R the resistor R which interconnects diode D and the source of potential E The resistance of R is sufficiently small so that the positive feedback due to the 5 network is now negligible enough to be ignored. That is to say, A A B L Regeneration by way of the B network thus etfectively ceases.

Again, while Schmitt circuit is in a state of normalcy, the diode D of the series switch 24 is forwardbiased by the voltage E which is manifest at juncture 44 of Fig. ,5. The voltage E is the positive direct-current voltage component of the feedback amplifier output signal. Regenerative propagation through the p3 circuit is thus unaffected by diode D while the feedback-amplifier is operating at permissible signal levels. While D is forward-bi d, i can se n th 14 2 i ui zeonsists of the series re i tors RB an 7R7 and the two sshunt' resistors R and R 10 3R8 When, however, in 'responseto tin-imminent amplifier overload, ,Schmitt circuit 20 changes state, rendering transistor 42 ,-nonconductive, diode D becomes reverse- .biased. The reverse-biasing of D is insured by propertioning E (it will be recalled that E, very nearly'a sumes this value when transistor .42 is out off) E :R and R so that E /R E /.R The series impedance of the p now additionally includes the extremely high impedance of the reverse-biased diode D Consequently, A A e 1 and, in fact, regeneration by way of the o circuit is effectively interrupted.

'In all of what has been said :above,'it should have been apparent that when the overload condition is removed, the feed-back ampiifier'returns to normal operation.

Figs. 4 and 5, already referred to inthe above discussion, show the manner in which the switching'networks 22 and 24may be connected to typical -/3 networks. The connections are self-explanatory. :Sufiice it to say that capacitor 4610f the shunt switch 22 intercouples juncture 28 of the 13 network and juncture 48 of Etheswitch. Also, as can readily be seen, diode'D 'of the series switch 24 interconnects juncture .32 of the 5 network and juncture .50 of :the switch. Juncture {'50, in turn, is connected-to resistor R The operationof the switching networks 122 and 24 has already been described in connection with Fig. 3. The amplifying portions 10f Figs. 4 and 5 are similar. It will be sufiicie'nt, therefore, in pointing up the interplay between the 8 circuits and .the amplifying elements, to briefly describe one of these figures, namely Fig. 5.

'Fig. 5, as has been mentioned above, is an interlaced multiple-loop feedback amplifier. As shown, it comprises three cascaded NPN junction transistors 60, 62, and 64, each connected in the common emitter configuration. The 13 circuit interconnects the collector output 66, of transistor :62 and the base input .68 of :transistor'60. The J9 circuit interconnects the collector output 70 of transistor 64 and the collector output 72'of transistor 60 which, in turn, is coupled to the base input 74 of transistor .62. The 5 circuit interconnects the collector output 70 ,of transistor 64 and the base input 68 of transistor 60. Appropriate bias potentials for each'of. the transistors 60, 62 and 64 are provided by the negative voltage power .supply 76 and the positive voltage power supply 104.

It should be noted that resistor 78 is inserted between the collector output 66 of transistor 62 and the base input 80 of transistor .64 so that the amount of current fed back by way of B circuit is efiectivelyindependent of any degradation that may occur in the output stage 64. Consequently, the value of resistor 78 should be at least "an order of magnitude greater than the input impedance of transistor 64.

Shaping of the gain characteristic, i.e., controlling the rate at which the current or voltage gain of the amplifier falls off with frequencies outside the useful frequency band, is an important consideration in the design of feedback arnplifiers. Low frequency shaping of loop current transmission is accomplished by capacitors 82, 84-and 86, each of which is connected between the emitter of its respective transistor and ground. Low frequency shaping is also provided by capacitor 88 and its associated resistors 90 and 92. 'High frequency shaping oft-he loop gain trans- In practice, R is small competed mission is accomplished by the B circuit and the 'interstage network which comprises the series combination of capacitor 94, resistor 96, and inductor 98. High frequency shaping is also provided by the network consisting of inductor and resistors 90 and 92.

The aforementioned shaping circuits serve to shape the loop gain of each of the positive feedback loops as well as the negative feedback loop and, consequently,

there is no need for shaping elements in the 9 and 18 circuits. Accordingly, the p, and fig circuit may be purely resistant, as they are in Fig. 5.

In order to stabilize the positive feedback provided by the B and ,8 circuits, the B circuit is designed for relatively large phase and gain margins. In addition, at frequencies outside the useful frequency band, the magnitude of the loop gain of each of the positive feedback loops should decrease at a rate which is greater than or at least equal to the rate at which magnitude of the loop gain of the negative feedback loop decreases. The interlaced amplifier illustrated in Fig. 5 is advantageously designed to have a 45-degree phase margin at both the low and high ends of its useful frequency band.

The input resistor 102 is serially inserted in the connections between the input terminal and the base input 68 of transistor 60, primarily to insure that substantially all of the current fed back through 3 and B circuits finds its way into the base input 68 of transistor 60. This precaution is not needed in the output circuit of transistor 60, since the relatively high output impedance of this transistor insures that substantially all of the current fed back through the B circuit finds its way into base input 74 of transistor 62.

Fig. 6 illustrates an alternative arrangement for the shunt switch 22 of Fig. l. PNP transistor 52 here serves as the fundamental switching element. The collector junction of the transistor is reverse-biased by the negative voltage --E-,. E-; is more negative than the voltage -E The relationship between the negative voltages E and E the Schmitt circuit output voltage shown in Fig. 3, remains the same as before. That is, so long as transistor 42 is conductive E E but when the Schmitt trigger circuit changes state and transistor 42 is rendered nonconductive, E E With this relationship in mind, it can be seen that so long as the emitter junction of transistor 52 is reverse-biased by the voltage E propagation through the 8; network will be unaffected, for the collector-emitter path of transistor 52 presents a very high impedance. But when an impending amplifier overload has caused transistor 42 to be cut off and, consequently, the voltage -E to become substantially equal to E the emitter junction of transistor 52 becomes forward-biased. Since transistor 52 is then driven into current saturation, its collector-emitter path presents a very low impedance. Under this state of affairs, the positive feedback due to the ,6 network is negligible and A 13 (Fig. l) or A,A 5 (Fig 2) is very much less than unity. Stability of the respective feedback amplifiers, pending cessation of the overload condition, is thus insured.

The series switch 24, shown in Fig. 7, employs an NPN transistor 54. This is in lieu of the diode D which -is included in the series switch 24 as it is shown, for

. more negative than --E thus forward-biases the emitter junction of transistor 54, driving it into saturation. The

collector-emitter path of the transistor is, therefore, normally a very low impedance to current passing through the 5 network. But when, in response to an impending overload condition, transistor 42 of Fig. 3 is cut off and the voltage -E increases negatively to the value of -E which is more negative than -E the emitter junction of transistor 54 becomes reverse-biased. The collectoremitter path of transistor 54 is then a very high impedance. Positive feedback by way of the a, circuit is therefore effectively discontinued so long as the overload condition persists.

Countervailing factors will, in various applications of the invention, determine the designers choice of switching element for the switches 22 and 24 of Fig. 1. Diode switches have the advantages of relative circuit simplicity, speed of operation, and extremely high reverse impedance. Transistor switches, on the other hand, are advantageous in view of their sensitivity (they require a relatively small amount of driver power) and their extremely low forward impedance.

While specific embodiments have been shown and described, they should not be construed as circumscribing the Scope of the invention.

What is claimed is:

1. In combination, a multiple-loop feedback amplifier comprising at least one regenerative feedback loop and an overall negative feedback loop, said regenerative feedback loop having associated therewith a switching circuit for disabling the regenerative loop; and a protection circuit for preventing overload instability in said multipleloop feedback amplifier, said protection circuit including and interconnecting said switching circuit with the output of said amplifier, and further including a direct-current detector for detecting signals at the output of said amplifier and converting them to direct current; bistable means responsive to the amplitude of said detected current to change state at a certain critical amplitude level of said current, at which level an overload in said multiple-loop feedback amplifier is imminent, said bistable means being in its normal state of equilibrium while said detected current is below said critical level and in its abnormal state of equilibrium while said detected current is above said critical level; means for supplying said detected current to said bistable means; and means interconnecting said bistable means and said switching circuit, said switching circuit disabling its associated regenerative feedback loop when said bistable means is in its abnormal state of equilibrium and enabling said associated loop when said bistable means is in its normal state.

2. The combination, in accordance with claim 1 in which said switching circuit comprises a diode.

3. The combination in accordance with claim 1 in which said switching circuit comprises a diode serially inserted within its associated regenerative feedback loop.

4. The combination in accordance with claim 1 in which said switching circuit comprises a diode connected in parallel circuit relationship with its associated regenerative feedback loop.

5. The combination in accordance with claim 1 in which said switching circuit comprises a transistor.

6. In combination, a multiple-loop feedback amplifier and a protection circuit for preventing overload instability in said amplifier, said amplifier comprising an overall negative feedback loop and a plurality of regenerative feedback loops, each of said regenerative loops having associated therewith a switching circuit for interrupting positive feedback whenever an overload in said amplifier is immediately prospective, said protection circuit including and interconnecting each of said switching circuits with the output of said amplifier and further including a directcurrent detector for converting said output of said amplifier to direct current, a bistable circuit responsive to the amplitude of said direct current and changing from its normal to its abnormal state of equilibrium at a predetermined amplitude level of said direct current, at which level said amplifier overload is immediately prospective, said detector including a level control for correlating the amplifier output level at which said overload is immediately prospective and said predetermined direct-current level at which said bistable circuit changes from its normal to its abnormal state of equilibrium, said bistable circuit reverting to its normal state when said direct-current output of said detector falls below said predetermined amplitude level, and means for supplying the output of said bistable circuit to each of said switching circuits,

' each of said switching circuits including means for interrupting regenerative propagation in its associated feedback loop whenever said bistable circuit is in its abnormal state.

7. The combinaion in accordance with claim 6 in which said amplifier is a tandem multiple-loop feedback amplifier having a local regenerative feedback loop around each stage of said amplifier.

8. The combination in accordance with claim 6 in which said amplifier is an interlaced multiple-loop feedback amplifier having a plurality of regenerative feedback loops, each including more than one stage of said amplifier.

9. The combination in accordance with claim 6 in which said bistable circuit is a Schmitt trigger circuit.

10. In combination, a multiple-loop feedback amplifier comprising a plurality of regenerative feedback loops each having unity loop gain; means to detect the imminent prospect of an overload condition in said amplifier; bistable circuit means, responsive to said detector means, and changing state when an imminent overload has been detected; and switching means, responsive to said bistable circuit means and associated with each of said regenerative feedback loops, to interrupt regeneration in said feedback loops upon said change of state of said bistable circuit means.

11. In combination, a conditionally stable circuit including at least one regenerative feedback loop and having a prescribed maximum output voltage beyond which said circuit will tend to become unstable, detector means to detect the output voltage of said conditionally stable circuit, bistable circuit means responsive to said detector means and changing from a normal to an abnormal state when said prescribed voltage has been surpassed, and switching means responsive to said bistable circuit means to interrupt regeneration in said feedback loop when said bistable circuit means has assumed its abnormal state.

12. The combination in accordance with claim 11 in which said detector means comprises a level control, an emitter-follower stage, and a peak detector, interconnected in the order named.

13. The combination in accordance with claim 11 in which said bistable circuit comprises a Schmitt trigger circuit having a pair of emitter-coupled transistor stages.

References Cited in the file of this patent UNITED STATES PATENTS 2,281,238 Greenwood Apr. 28, 1942 2,379,768 Wynen July 3, 1949 2,581,953 Hecht Jan. 8, 1952 2,752,432 Richter June 26, 1956

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Referenced by
Citing PatentFiling datePublication dateApplicantTitle
US3175158 *May 25, 1961Mar 23, 1965Flesher Gail TControlled decay feedback type comb filters
US3206685 *Jun 13, 1961Sep 14, 1965Gen Motors CorpNon-linear amplifier circuit
US3264572 *Feb 15, 1963Aug 2, 1966Tia Electric CompanyTransiently regenerative amplification
US3264573 *Jan 24, 1964Aug 2, 1966Tia Electric CompanyTransiently regenerative amplifiers with response controlling means
US3398395 *Apr 28, 1966Aug 20, 1968Texas Instruments IncSeismic amplifier system with preprogrammed gain control
US3456204 *May 27, 1965Jul 15, 1969Honeywell IncTransistor amplification circuitry
US3571625 *Jul 25, 1968Mar 23, 1971Bell Telephone Labor IncPulse amplifier with positive feedback
US4358763 *May 9, 1980Nov 9, 1982U.S. Philips CorporationContinuous-wave radar responder having two-position switches
US7365597 *Aug 19, 2005Apr 29, 2008Micron Technology, Inc.Switched capacitor amplifier with higher gain and improved closed-loop gain accuracy
US7605650Mar 29, 2008Oct 20, 2009Micron Technology, Inc.Switched capacitor amplifier with higher gain and improved closed-loop gain accuracy
Classifications
U.S. Classification330/86, 330/112, 330/9, 330/104, 330/144, 327/205
International ClassificationH03F1/34, H03F1/36
Cooperative ClassificationH03F1/36
European ClassificationH03F1/36