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Publication numberUS3345589 A
Publication typeGrant
Publication dateOct 3, 1967
Filing dateDec 14, 1962
Priority dateDec 14, 1962
Publication numberUS 3345589 A, US 3345589A, US-A-3345589, US3345589 A, US3345589A
InventorsDi Piazza Gerald C
Original AssigneeBell Telephone Labor Inc
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Transmission line type microwave filter
US 3345589 A
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Description  (OCR text may contain errors)

Oct. 3, 1967 G. c. DI PIAZZA 3,345,589

TRANSMISSION LINE TYPE MICROWAVE FILTER Filed Dec. 14, 1962 2 Sheets-Sheet 1 PIP/OR APT F/GIZ a /NVE/VTOR 6. C. D/ P/AZZA ATTORNEY,

0ct. 3, 1967 G. c. DI PIAZZA TRANSMISSION LINE TYPE MICROWAVE FILTER 2 Sheets-Sheet 2 Filed Dec. 14, 1962 United States Patent Office 3,345,589 Patented Oct. 3, 1967 3,345,589 TRANSMISSION LINE TYPE MICROWAVE FILTER Gerald C. Di Piazza, Lake Hiawatha, N.J., assignor to The present invention relates to frequency selective net- Works, and more specifically to narrow-band microwave filters.

In many microwave applications, it is desirable to transmit a narrow band of frequencies while simultaneous- 1y suppressing all other frequencies. Prior art filters for accomplishing this end have taken many and varied forms including the well known capacity-coupled half-wave resonator structures. Examples of such devices are disclosed in United States Patents Nos. 2,859,417 and 2,867,782, both granted to M. Arditi on Nov. 4, 1958, and J an. 6, 1959, respectively.

Most of the prior art filters, due largely to the inherent limitations of the resonator structures employed, suffer one or more practical disadvantages. These disadvantages often include high inband insertion loss, relatively low selective capability and relative difficulty of design or construction. It is therefore an object of the present invention to reduce the inband insertion losses in narrow-band microwave bandpass filters.

It is a further object of the present invention to increase the selective capabilities of microwave bandpass filters.

It is another object of the present invention to simplify the design and construction of high selectivity microwave bandpass filters.

The foregoing objects are accomplished in accordance with the principles of the present invention by utilizing a transmission line having a uniform characteristic admittance across which there are shunt-connected transmission line elements having characteristic admittances greater than that of the uniform line. As used herein, the term transmission line is understood to mean any structure capable of supporting propagating high frequency electromagnetic wave energy. Such structures can include coaxial lines, strip transmission lines, parallel wire transmission lines and conductively bounded waveguiding structures. The shunt transmission line elements are open-circuited at one end and short-circuited at the other end. The connection between the uniform line and each shunt line is at a point intermediate these two ends. A shunt transmission line element connected in this fashion is known as a ftapped stub. The ratio of the length of the open stub section to the total stub length is known as the tapping ratio.

By utilizing the so-called loaded Q approach, it has been found that, as the ratio of stub-to-line characteristic admittance is increased, the selectivity of the filter is likewise increased. The selectivity of the filter of the present invention can be further increased by increasing the tapping ratio mentioned above.

In another embodiment of the invention, even greater selectivity is obtained by the utilization of a branched transmission line configuration. In this embodiment, a first uniform transmission line branches into a plurality .of secondary uniform transmission lines. In general, the characteristic admittance of each branching line is substantially equal to that of' the first uniform line divided by the number of branching lines. Frequency selective stu'bs or circuits are shunt-connected across each of the secondary lines. The secondary transmission lines again converge to form a uniform transmission line having a characteristic admittance substantially equal to that of the first uniform line.

A distributed shorting element for conductively shorting a section of a strip transmission line or stub to a pair of extended conductive ground planes is also described. This element consists of a thin piece of conductive material bent into the form of an E and held in physical and electrical contact with the ground planes and center conductive strip by external clamping means.

The above mentioned and other features and objects of the present invention will become more apparent by reference to the following description taken in conjunction with the accompanying drawings in which:

FIG. 1 is a schematic illustration of a transmission line filter of the type well known in the prior art;

FIG. 2 is a schematic illustration of a two-element bandpass filter in keeping with the principles of the present invention;

FIG. 3 is a graphical representation of the tapping ratio of the shunt stubs of the embodiment of FIG. 2 plotted as a function of normalized frequency for various ratios of stub-to-line characteristic 'adunittances;

FIG. 4 is a pictorial view of the embodiment of FIG. 2 adapted for a strip transmission line configuration;

FIG. 5 is a pictorial illustration of a second embodiment of the present invention adapted for strip transmission line configuration;

FIG. 6 is a pictorial View of an improved distributedtype shorting element capable of being employed in stirp transmission line devices;

FIG. 7 is an exploded pictorial view of a portion of the embodiment of FIG. 4 showing the manner in which the distributed shorting element of FIG. 6 is utilized; and

FIG. 8 is a cross-sectional view of the embodiment of FIG. 4 showing the distributed shorting element of FIG. 6in place. 1

Referring more specifically to the drawings, FIG. 1 is a schematic illustration of a transmission line filter of the type well known in the art. In this structure, a plurality of transmission line stubs designated 1, 2, 3 n are connected at one-quarter wavelength intervals to a uniform transmission line 10. The total Q of the filter Q represents a combination of the Qs of the constituent sections. These Qs are designated Q Q Q Q and comprise theQof each stub plus the compensation for the selectivity introduced by the sections of line 10 immediately adjacent to each stub. It is well known that the Q attributable to a one-quarter wavelength section of uniform line in such a filter can be approximated, in a narrow-band filter, by the addition of the factor 1r/8 to the Q of each stub. (See US. Patent No. 2,540,488 granted to W. W. Mumford on Feb. 6, 1951.)

3 where Q Q Q Q represents the Q of each stub 1, 2, 3, n, respectively.

In the above cited patent of W. W. Mumford, it is stated that for a maximally-flat bandpass filter, the Q of any section is given by Q.=QT sin 2) where r denotes the order of the stub. (See also US. Patent No. 1,849,656 granted to W. R. Bennett on Mar. 15', 1932.)

In terms of the characteristic admittance, it can be shown that where Y is the characteristic admittance of line and Y is the characteristic admittance of the rth stub.

Since it is a useful parameter in the design of filters in accordance with the present invention, the ratio Y /Y will be designated hereinafter as k From Equation 3 it is obvious that the Q of a stub increases as its characteristic admittance Y increases. In general, therefore, high Q is characterized by high stub admittance.

It has been found that the use of a tapped stub configuration in transmission line filters yields higher Qs than the shorted stub configuration of FIG. 1. A schematic diagram of a two element tapped stub bandpass filter is shown in FIG. 2. In general, any number of stubs can be utilized in practicing the present invention; however, for the purposes of illustration, a filter utilizing two stubs is described.

The bandpass filter of FIG. 2 consists of a length of uniform transmission line having a characteristic admittance Y across which are connected tapped transmission line stubs 21 and 22. Stubs 21 and 22, which have characterstic admittances Y and Y respectively, are connected in shunt with line 20; and spaced apart onequarter wavelength at the midfrequency of the band of frequencies to be passed. The overall length of each stub 21 and 22 is also one-quarter wavelength at said midband frequency, although as will be discussed in greater detail hereinbelow, the length and spacing of these stubs can be any odd multiple of one-quarter wavelength. The connections between line 20 and stubs 21 and 22 are made at points intermediate their respective ends. Each stub is open circuited at one end and short circuited at the other end. The electrical distance from the tapped connection to the shorted end of each stub is designated 1 and the distance between this connection and the open end is designated From Equation 2 the Qs of stubs 21 and 22 are and since Y is constant, it is seen from Equation 3 that o2= o1- The value of susceptance due to stub 21 or 22 at the junction of line 20 and the stub is described by where [3 is the phase constantof the stub and is equal to 21r/)\, where A is the wavelength measured along the stub. The normalized susceptance B can be found by dividing Equation 6 by the characteristic admittance Y of line 20.

- J'n 1;L ]BN Y0 tan [3Z2 Y0 001,51 I and since 1/01- &2 'YOIC1-YOICZ k then tan cot fil Since the tapping ratio a l /l, then l =(1a)l, and

B =k[taIl aBZ-COt (1oz)Bl] 1+tan Bl tan aBZ l: am tan BZtan 0431 B tan aBZ+l tan aBl-tan s2 9) Solving for tan 0:51,

It can further be shown that at the half power points the normalized susceptance B of the stub is equal to two. Substituting this value of B in Equation 10 gives where 5,, is the phase constant of the stub at the cutoff or half power points. Since l= /4,

where f and f are the frequencies corresponding to midband and cutoff respectively.

Substituting and solving for or 1/2 l[1k 2k tan tan 1 As shown hereinabove, the loaded Q or each stub section can be increased by increasing the ratio k while keeping the other variables constant. The selectivity of a bandpass filter in accordance with the present invention, therefore, can be improved by making the ratio k larger than unity. However, since the relationship between k and the loaded Q is a smooth function this improvement is gradual. As will be discussed in greater detail hereinbelow, it is generally desirable to make k as large as possible within limits imposed by the physical dimensions of the filter structure; although in a practical filter, a significant improvement is obtained when the ratio k is larger than two.

FIG. 3 is a graphical representation of the tapping I ratio a plotted as a function of f /f for values of k equal to 1.0, 2,5, 3.33, and 5.0. The curves were obtained by inserting the appropriates values of k and f /f into Equations 13 and 14 and solving for a. From a physical standpoint, the graph of FIG. 3 indicates that, for a given bandwith, a filter can be realized by choosing either a relatively high tapping ratio, a, and low k, or a relatively low a and high k.

As an illustration of the design procedure, the following example is given. It is the objective of the design to obtain a two-section maximally-flat narrow-band band- :pass filter having a center frequency of 2220 megacycles per second and an attenuation of at least 10 decibels at afrequency of 2100 megacycles per second. The filter is a) to have an insertion loss as small as possible at the center frequency.

The loss function of a maximally-flat filter is given where again n is the number of stubs. By inserting the design values into Equation 15, the total Q, Q of the filter can be obtained, Thus,

n 1 minim g P i QT 2220 2100 (16) In other words, a two-section filter having a total Q of 16 will satisfy the specified design requirements. In order to provide a certain design margin, a convenient bandwith of 100 megacycles per second shall be chosen for the filter. At a center frequency of 2220 Inegacycles per second, this bandwith corresponds to a Q of 22.2.

The schematic diagram of the illustrative filter corre sponds to that shown in FIG. 2 and the nomenclature and numbers of the various elements are therefore carried over to this example.

Substituting into Equation 2, the Q of each section is,

(2 :22.2 sin 7r/4=l57 (17) Q =22.2 sin 31r/4=15.7 (18) These values are then substituted into Equation 1 to determine the Qs of the stubs.

= 0.065 normalized bandwidth In this example, it is assumed that k and k equal 2.5. Therefore, from FIG. 3 the tapping ratio a is approxi mately 0.83. If a more exact ratio of a tapping ratio is desired, it can be obtained from Equations 13 and 14.

At this point, it should be noted that the curves of FIG. 3 are not quite symmetrical about the center frequency f For this reason, a particular value of a will not yield a value of f which corresponds exactly with the arithmetic mean of the upper and lower cutoff frequency. It is seen, however, that for stubs of high Q this difference between the desired and the actual values of t is negligible.

Once the values of k and a have been determined, it is a relatively simple matter to realize the desired filter. The type of transmission line structure (for example, coaxial line, waveguide, strip transmission line, etc.) should be chosen to meet the application for which the filter is intended. The characteristic admittance of uniform line is generally dictated by the characteristic admittance of the connecting lines of the utilization circuit.

Because of the many advantages enjoyed by strip trans mission line structures, further illustrative embodiments of the present invention will be shown with reference to such structure.

A strip transmission line filter, constructed in accordance with the teachings of the present invention, is illustrated in the pictorial view of FIG. 4. In FIG. 4 a uni form strip transmission line 40 having tapped stubs 41 and 42 is bonded, deposited or etched on a dielectric sheet 43. The dielectric sheet 43 can comprise any suitable low loss material. For example, the materials known commercially as Teflon or Tellite have been successfully employed in these structures. A conductive ground plane 44 is bonded to the bottom side of sheet 43. A second sheet of dielectric material 45, having conductive ground plane bonded to its upper surface, is positioned above sheet 43. For the purpose of clarity in FIG. 4, the two dielectric sheets are shown separated. In practice, however, it is understood that the two sheets would be clamped together by cover plates or other securing means well known in the art.

Shorting eyelets or rivets 47 are utilized to short circuit the bottom portion of stubs 41 and 42 to ground plane 44. A similar set of eyelets 48% extend from the upper ground plane 46 through dielectric sheet 45 to contact eyelets 47 and thereby short stubs. 41 and 42 to the upper ground plane.

The entire filter structure of FIG. 4 therefore resembles a sandwich wherein the center conductor and the stubs are sandwiched between two conductive ground planes separated by insulating sheets. In such a structure, the characteristic admittances of the center conductor and stubs are functions of their widths and thicknesses as well as the dielectric constant of the insulating sheets and the spacing between the ground planes. The relation between these variables is given in Reference Data for Radio Engineers, 4th edition, International Telephone & Telegraph Corporation, New York, 1956, pages 598600.

In the embodiment of FIG. 4, the thickness of the dielectric sheets 43 and 45 is inch. The dielectric constant of the material of sheets 43 and 45 is 2.25 over the frequencies of operation. Although, for the sake of clarity, the thicknesses of the ground planes and conductive strips are shown in exaggerated proportion in the drawings, their actual thicknesses are negligible compared to the thicknesses of the dielectric insulating sheets. These dimensions are primarily determined by the commercial availability of the conductively clad dielectric insulating sheets. With the dielectric constant specified, the length of stubs 41 and 42 was determined as 0.88 inch. This corresponds to one-quarter wavelength at the center frequency of 2220 megacycles per second. Since then . l =0.732 inch (22) and l =0.148 inch (23) The spacing d between stubs 41 and 42 likewise corresponds to one-quarter wavelength at frequency Due to the fringing elfects associated with the finite width of the stubs, however, it was determined experimentally that the spacing should be increased a few percent for optimum results. The spacing finally employed was 0.905 inch. The characteristic admittance of the uniform line 40 was chosen as 1 mho which corresponds to a line width of 0.10 inch. Since the characteristic admittance of stubs 41 and 42 equals k times that of the line, or mho, their width are 0.33 inch.

As shown above, high Q is associated with a high ratio k of stub-to-line characteristic admittance. With a filter having the physical structure of the embodiment of FIG. 4, however, the Q can only be increased to a certain point. This is so because as Q increases, k must increase, and with a transmission line of a given characteristic admittance, a greater k corresponds to a greater stub width. When the width of the stubs becomes significant compared to the one-quarter wavelength spacing between them, undesirable fringing effects and interaction between the two stubs occur.

One solution of this problem would be to increase the spacing between the stubs to a higher odd multiple of one-quarter wavelength (for example, three-quarter wavelength). This, however, has the obvious disadvantage of making the overall length of the filter much longer. In

addition, such a structure has higher losses and a spuri- 'ous resonance point corresponding to that frequency at which the spacing equals one-quarter wavelength.

Another solution is to construct the filter in accordance with'the embodiment illustrated in the pictorial view of FIG. 5. The principal features of the filter of FIG. are its lower insertion loss and its greater selective capabilities. In the embodiment of FIG. 5 there is shown a conductive ground plane 50 bonded to the lower surface of a dielectric insulating sheet 51. A uniform transmission line 52 having a given characteristic admittance is bonded, deposited or etched on the upper surface of insulating sheet 51. At junction 53 along its length, line 52 branches into two separate but substantially identical lines 54 and 55 which again merge into line 52 at junction 58. The dimensions of lines 54 and 55 are proportioned so that their characteristic admittances are substantially equal to one-half the characteristic admittance of line 52.

A pair of quarter-wave tapped stubs 56 are shunted across line 54 and spaced apart one-quarter wavelength at the center frequency. A substantially identical pair of stubs 57 are similarly connected across line 55. The characteristic admittances and tapping ratios on of stubs 56 and 57 are determined in accordance with the design procedure given above except that the uniform line used as a reference is line 52 rather than branching lines 54 and 55.

Although in the embodiment of FIG. 5, the upper ground plane and insulating sheet are not shown, it is understood that in order to prevent radiation and accompanying losses, it is generally advantageous to utilize these elements in a manner similar to that shown in connection with the filter structure of FIG. 4.

Although only one embodiment is illustrated in FIG. 5, it is understood that it is susceptible to many modifications. For example, more than two branching transmission lines may connect the two sections of uniform line 52. In such cases, the characteristic admittance of each of the branching lines is substantially equal to that of the uniform line divided by the number of branching lines. Likewise the frequency selective portions of each branching line may take a form other than that of the tapped stubs shown in FIG. 5. Thus, tapped stubs 56 and 57 can be replaced by untapped stubs or other frequency selective microwave circuits known in the art.

Furthermore, if the electrical length of each of the branching lines is made equal to an integral multiple of one-half wavelength at the midband of the band of frequencies to be passed, their characteristic admittances canassume a value other than the described fraction of the characteristic admittance of the uniform line.

In the embodiment of FIGS. 4 and 5, the shorted ends of the tapped stubs are shown connected to the respective ground planes by means of small rivets or eyelets. FIG. 6 shows in pictorial view an alternative device for use in conductively shorting a portion of a strip transmission line to the upper and lower ground planes. Element 60 consists of a thin rectangular piece of conductive material, such as copper, bent into the form of an E. The width of this element is preferably equal to the width of the stub to be shorted.

FIG. 7 is an exploded pictorial view of a portion of the strip transmission line filter of FIG. 4 wherein shorting element 60 has replaced eyelets 47 and 48. Like numbers have been carried over from FIG. 4 to FIG. 7 to correspond with like structural elements. Blocks 70 and 71 have been cut from the upper and lower dielectric sheets respectively to facilitate insertion of element 60.

FIG. 8 is a cross-sectional view of the resulting structure with element 60 and blocks 70 and 71 in place. As

element 69 and the ground planes 44 and 46 have been greatly exaggerated. In practice, these thicknesses are on the order of a few thousands of an inch. As a consequence, there is little if any gap between the strip transmission line and the lower surface of dielectric sheet 45.

Although certain specific embodiments of the invention have been shown in the drawings and described in the foregoing specification, it is understood that the invention is not limited to those specific embodiments, but is capable of modification and rearrangement by those skilled in the art without departing from the spirit and scope of the invention.

What is claimed is:

1. A filter capable of passing microwave energy over a given band of frequencies comprising, in combination, a uniform transmission line having a given characteristic admittance, a plurality of tapped transmission line stubs shunted across said line at intervals substantially equal to an odd multiple of one-quarter wavelength at the midfrequency of said given band of frequencies, each of said stubs having a length substantially equal to an odd multiple of one-quarter wavelength at said midfrequency, each of said stubs having one open-circuited and one short-circuited end, the connection between said uniform line and each of said stubs being intermediate said ends, and wherein the characteristic admittance of each of said stubs is substantially greater than that of said uniform line.

2. A microwave filter comprising, in combination, a first and second uniform transmission line section having a given characteristic admittance, a plurality of branching transmission lines connected between one end of said first transmission line section and one end of said second transmission line section, the characteristic admittance of each of said branching lines being substantially equal to said given characteristic admittance divided by the number of said branching lines, and at least one separate frequency selective means associated with each of said branching lines.

3. The filter according to claim 2 wherein the length of each of said branching lines is substantially an integral multiple of one-half wavelength at said midfrequency.

4. A balanced strip transmission line filter capable of passing microwave energy extending over a given band of frequencies comprising, in combination, a pair of extended conductive surfaces in spaced parallel relationship, a thin strip of conductive material having a substantially uniform width disposed between said surfaces in parallel relation thereto and conductively insulated therefrom, a plurality of conductive strip stubs conductively connected to said strip at intervals substantially equal to an odd multiple of one-quarter wavelength at the midfrequency of said band of frequencies, said stubs having lengths substantially equal to an odd multiple of one-quarter wavelength at said midfrequency and widths substantially greater than that of said strip, said stubs being connected to said strip at a region along their lengths intermediate their ends, and means for conductively shorting one end of each of said stubs to said conductive surfaces.

5. A balanced strip transmission line filter for electromagnetic wave energy comprising, in combination, a pair of extended conducting surfaces in spaced parallel relationship, first and second longitudinally spaced thin strips of conducting material disposed between said surfaces in parallel relation thereto and conductively insulated therefrom, said first and second strips being dimensionally proportioned to offer a given characteristic admittance to said energy, a plurality of branching strips conductively connecting adjacent ends of said first and second strips, each of said branching strips being dimensionally proportioned to offer a characteristic admittance to said energy which is substantially equal to said given characteristic admittance divided by the number of said branching strips, and separate frequency selective means associated with each of said branching strips.

6. The filter according to claim 5 wherein two branching strips are utilized and wherein said frequency selective means comprises at least one tapped stub element.

7. A filter capable of passing microwave energy over a given band of frequencies comprising, in combination, a uniform transmission line having a given characteristic admittance, a plurality of tapped transmission line stubs shunted across said line at intervals substantially equal to an odd multiple of one-quarter wavelength at the midfrequency of said given band of frequencies, each of said stubs having a length substantially equal to an odd multiple of one-quarter Wavelength at said midfrequency, each of said stubs having one open-circuited and one short-circuited end, the connection between said uniform line and each of said stubs being intermediate said ends, and wherein the characteristic admittance of each of 10 said stubs is at least twice the characteristic admittance of said uniform line.

References Cited UNITED STATES PATENTS 2,819,452 1/1958 Arditi 333-84 2,859,417 11/1958 Arditi 333-84 2,915,716 12/1959 Hattersley 333 73 2,945,195 7/1960 Matthaei 333-73 2,964,718 12/1960 Packard 333-73 10 2,984,802 5/1961 Dyer 333-43 FOREIGN PATENTS 798,629 7/1958 Great Britain.

15 HERMAN KARL SAALBACH, Primary Examiner.

C. BARAFF, Assistant Examiner.

Disclaimer 3,345,589.Gemld 0. Di Piazza Lake Hiawatha, N.J. TRANSMISSION LINE TYPE MICROWAVE FILTER. Patent dated Oct. 3, 1967.

Disclaimer filed June 5, 1972, by the assignee, Bell Telephone Laboratories Incorporated.

Hereby enters this disclaimer to claims 1, 4 and 7 of said patent.

[Oyficz'al Gazette January 2, 1.973.]

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Referenced by
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US3428918 *May 26, 1966Feb 18, 1969Us ArmyMultiplexer channel units
US3497835 *Dec 10, 1965Feb 24, 1970Hughes Aircraft CoMicrowave filter
US3530411 *Feb 10, 1969Sep 22, 1970Bunker RamoHigh frequency electronic circuit structure employing planar transmission lines
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US4999596 *Nov 28, 1989Mar 12, 1991Fujitsu LimitedSecond-harmonic-wave chocking filter
US5097236 *May 1, 1990Mar 17, 1992Murata Manufacturing Co., Ltd.Parallel connection multi-stage band-pass filter
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US7057481 *Mar 9, 2004Jun 6, 2006Alpha Networks Inc.PCB based band-pass filter for cutting out harmonic high frequency
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US20110081873 *Aug 19, 2010Apr 7, 2011Fujitsu LimitedFilter, transmitter-receiver, and amplifying circuit
US20110102111 *Dec 15, 2009May 5, 2011Hon Hai Precision Industry Co., Ltd.Band-pass filter circuit with transmission lines
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Classifications
U.S. Classification333/204, 333/246
International ClassificationH01P1/201, H01P1/20
Cooperative ClassificationH01P1/201
European ClassificationH01P1/201