US 3489930 A
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Jam 13, 1970 A. SHOH APPARATUS FOR CONTROLLING THE POWER SUPPLIED TO AN ULTRASONIC TRANSDUCER Original Filed June 21, 1967 5 Sheets-Sheet 1 M 1 (IL 0 R 1 .rv Ji D a INVENTOR. fladrew .571 all BY MAL KJMZfA/.
Jan. 13, 1970 A. SHOH 3,489,930 APPARATUS FOR CONTROLLING THE POWER SUPPLIED TO AN ULTRASONIC TRANSDUCER Original Filed June 21, i967 5 Sheets-Sheet 2 Jan. 13, 1970 A. SHOH APPARATUS ,FOR CONTROLLING THE POWER SUPPLIED TO AN ULTRASONIC TRANSDUCER 1.967
5 Sheets-Sheet 5 Original Filed June 21,
Jan. 13, 1970 A. SHOH 3,489,930 APPARATUS FOR CONTROLLING THE POWER SUPPLIED TO AN ULTRASONIC TRANSDUCER Original Filed June 21, 1967 5 Sheets-Sheet 4 Jan. 13, 1970 A. SHOH APPARATUS FOR CONTROLLING THE POWER SUPPLIED TO AN ULTRASONIC TRANSDUCER 1967 5 Sheets-Sheet 5 Original Filed June 21,
NWN @W United States Patent O 3,489,930 APPARATUS FOR CONTROLLING THE POWER SUPPLIED TO AN ULTRASONIC TRANSDUCER Andrew Shoh, Ridgefield, Conn., assignor to Branson Instruments, Incorporated, Stamford, Conn., a corporation of Delaware Continuation of application Ser. No. 647,701, June 21, 1967. This application July 29, 1968, Ser. No. 751,683 Int. Cl. H04r 17/00; G01]: 11/00 US. Cl. 3108.1 7 Claims ABSTRACT OF THE DISCLOSURE The disclosed apparatus includes a source supplying power to a piezoelectric, ultrasonic transducer. An appropriately valued capacitor is utilized to derive an electrical current proportional to the clamped capacity of the transducer; this current being subtracted from the total current through the transducer to derive a current 1 proportional to the motional current and thus the amplitude of mechanical vibration of the transducer. A feedback circuit responsive to the current I operates to control the source accordingly to thus supply energizing power to the transducer calculated to maintain the amplitude of mechanical vibration thereof constant or at least below a predetermined maximum.
REFERENCE TO RELATED APPLICATIONS This application is a continuation of application 647,- 701, filed June 21, 1967 now abandoned; which is a divisional application of my copending application Ser. No. 416,816, filed Dec. 8, 1964, entitled Sonics; which is in turn a continuation-in-part of my application Ser. No. 265,751, filed Mar. 18, 1963, entitled Ultrasonic Cleaning Apparatus, now issued as Patent No. 3,293,456.
BACKGROUND AND OBJECTS OF THE INVENTION In recent years sonic energy has found wide use in science and industry for cleaning, soldering, welding, material treatment, homogenizing, dispersing, microbiological cell disruption and the like. For many years both magnetostrictive and electrostrictive (piezoelectric) transducers have been used in sonic cleaning systems. Magnetostrictive transducers utilize metal laminated core structures that change in length when subjected to a magnetic field provided by a coil. Electrostrictive or piezoelectric transducers utilize crystalline or ceramic elements that change their length under the influence of an applied electric field. Except for the above-mentioned use in cleaning tanks, up until a few years ago, the vast majority of sonic transducer systems for providing high sonic power levels were of the magnetostrictive type. However, the development of piezoelectric ceramics, such as lead titanite lead zirconate, capable of withstanding the high temperatures developed in high powered transducer systems have led to their use in high power general purpose systems. Such a general purpose transducer system or sonic converter is disclosed in the copending application of Stanley E. Jacke and Henry Biagini, Ser. No. 384,025 filed July 13, 1964 entitled Sonics.
In the above-identified application the sonic vibrations developed electrostrictively in a pair of ceramic wafers are concentrated and increased in amplitude by a concentrating horn or acoustic impedance transformer to provide at the end thereof extremely high power densities per unit area. This sonic converter is being used in many of the ultrasonic process applications listed above, for example, for homogenizing, dispersing and disrupting bio- 'ice logical cells, in ultrasonic welding, soldering and treatment of materials and the like.
As discussed in the above-identified copending application, one of the problems encountered in operating such a high power sonic converter is that when the device is energized and the tip of the concentrating horn is not coupled to any medium other than air the tip vibrates violently at very large amplitudes. Very little power is transferred to the air and substantially all of the power supplied to the converter must be dissipated therein. In contrast therewith, when the tip of the concentrating horn is coupled to a less compliant medium, such as a liquid, energy transfer occurs from the horn to such a medium, thus leaving a smaller amount of energy to be dissipated in the converter. Depending upon the efliciency of the converter, the medium which receives the acoustic power and the degree of coupling achieved, the power to be dissipated in the converter may vary over a ratio of 10 to 1 from the condition of no power transfer to the load, to the other condition when good power transfer from the horn to the load is achieved.
In order to protect the transducers from the possibility of destruction by excessive power dissipation, the prior art employed principally two approaches. One method is to limit the power supplied to the converter to the amount of power which the converter safely can dissipate under the condition of no power transfer. As is readily apparent, this approach seriously limits the power which is available to the converter when good power transfer between the horn and the load is achieved. The other approach concerns the provision of manually adjustable power control means, for instance, selectable voltage levels in order to vary the power supplied to the converter. This latter method, as is readily apparent, is not satisfactory as the operator must judge the degree of power transfer and, in the event that too much power is applied to the converter, or if the converter is uncoupled from the load and inadvertently left to operate into air without power reduction, the converter may destroy itself.
As a consequence of these problems, converters of this type have been generally limited to low power levels. For extmple, the converter disclosed in the above-identified copending application has been limited to a maximum of approximately 100 watts. This being the greatest power the converter can internally dissipate when operating in air. Generally speaking the greater the impedance of the acoustic load presented to the converter the smaller the power supplied to the medium.
I have discovered methods, apparatus and systems for controlling the power supplied to a converter, specifically as disclosed in the above-identified copending application. In order to better understand my invention I have made the following observation utilizing an S converter manufactured by Branson Sonic Power Division of Branson Instruments, Inc. in Danbury, Conn. in accordance to the above-identified copending application.
The Branson S-75 converter was connected to a power amplifier to form an oscillator in the manner described in the above-identified copending application, Ser. No. 265,751. As explained therein, it thus operated at the frequency of maximum conversion elficiency. The converter using a solid step horn acoustic transformer was operated first in air. Various voltages were supplied to the converter. The amplitude of the motion of the tip of the concentrating horn and the motional current supplied to the converter were measured. The motional current was derived in the manner described in the aboveidentified copending application. That is, a current equal to the clamped capacitance current of the converter was subtracted from the total current supplied to the converter. This produces a current equal to the motional current in the series branch equivalent circuit of the converter.
It was found that the motional current supplied to the converter and the amplitude of the motion of the tip of the concentrating horn were directly proportional throughout the useful operating range of the converter. It was further found that this proportionality was independent of the acoustic load presented to the transducer when the transducer was operated in air, water or heavy oil.
It was further found by driving the converter with a variable frequency oscillator that this proportionality was substantially independent of frequency at or near the natural resonance of the transducer.
It was further found that the internal dielectric losses of the converter were small and may be neglected. Furthermore it was found that the internal losses in the converter appear to the driving generator or amplifier as a constant impedance in series with a varying impedance directly proportional to the acoustic load presented to the converter.
It is therefore an object of the present invention to provide methods, apparatus and systems for keeping the heat dissipation in a sonic converter constant or below a predetermined maximum.
Another object of the invention is to provide methods, apparatus and systems for keeping the mechanical deformation in a sonic converter constant or below a predetermined maximum.
A further object of the invention is to provide methods, apparatus and systems for keeping the amplitude of the motion of the transducing elements of a sonic converter constant or below a predetermined maximum.
Still another object of the invention is to provide methods, apparatus and systems for keeping the power dissipated in a sonic converter constant or below a pre determined maximum.
A still further object of the invention is to provide methods, apparatus and systems for keeping an electrical quantity supplied to a sonic converter proportional to the quantities of the above objects constant or below a predetermined maximum.
Still another object of the invention is to provide methods, apparatus and systems for keeping motional current supplied to an electrostrictive sonic converter constant or below a predetermined maximum.
Another object of the invention is to provide automatic methods, apparatus and systems for increasing the total power supplied to a sonic converter as the acoustic impedance into which the converter is working increases.
Still another object of the invention is to provide methods, apparatus and systems for keeping the total current supplied to a sonic converter employing electrostrictive elements substantially constant or below a predetermined maximum.
Yet another object of the invention is to provide methods, apparatus and systems for driving a high power sonic converter from an apparently high impedance source.
A further object of the invention is to control the power delivered to an ultrasonic transducer.
Another object of the invention is to compensate for the varying acoustic impedances into which an ultrasonic power transducer operates.
A further object of the invention is to provide methods, apparatus and systems whereby a conventional ultrasonic power transducer is able to deliver more power into a load than heretofore possible.
Other objects of the invention will in part be obvious and will in part appear hereinafter.
The invention accordingly comprises the several steps and the relation of one or more of such steps with respect to each of the others and the apparatus and systems employing features of construction, combinations of elements and functions, and arrangements of parts which are adapted to effect such steps, all as exemplified in the following detailed disclosure. The scope of the invention will be indicated in the claims.
For a fuller understanding of the nature and objects of the invention reference should be had to the following detailed disclosure taken in connection with the accompanying drawings in which:
FIGURE 1 is a front view, partially cut away, of an ultrasonic power converter and electronic driver in accordance with the present invention.
FIGURE 2 is an equivalent circuit of the ultrasonic power converter shown in FIGURE 1.
FIGURE 3 is a block circuit diagram of one embodiment of the present invention.
FIGURE 4 is a block circuit diagram of another embodiment of the invention.
FIGURE 5 is a block circuit diagram of another embodiment of the invention.
FIGURE 6 is a block circuit diagram of a preferred embodiment of the invention.
FIGURE 7 is a detailed circuit diagram of the embodiment of the invention shown in FIGURE 6.
FIGURE 8 is a block circuit diagram of another embodiment of the invention.
FIGURES 8a and 8b are diagrams showing the method of operation of the embodiment of the invention of FIG- URE 8.
The same reference characters refer to the same elements throughout the several views of the drawings.
GENERAL DESCRIPTION The ultimate objects of the present invention are to limit the dissipation in a sonic converter to a safe, maximum value for all loading conditions and to obtain a loading characteristic wherein the mechanical power output from the converter increases when the converter operates into a load of increased acoustical impedance.
The invention may be realized in its most perfect form by the use of feedback techniques to hold an electrical quantity proportional to acoustic deformation or motion in the converter constant.
In the case of converters employing electrostrictive elements such an electrical quantity is the motional current I This current may be derived from the electrical power supplied to the converter by compensating for the clamped capacitance of the converter in the manner described in the above-identified copending application Ser. No. 265,75 1.
Alternatively the electrical quantity held constant may be derived from a second pickup or slave transducer driven by the converter. Such techniques are also described in the above article.
According to this invention, the electrical quantity proportional to motion in the converter is subtracted from a constant reference electrical quantity of the same kind. The resultant error or negative feedback signal is used to control the source of electrical power driving the converter. Thus, the motion in the converter is held constant.
Since the heat dissipated in the converter is a function of the motion therein, this is held constant accomplishing the first major object of the invention. Furthermore, the motion in the converter may be chosen below that which will cause undue fatique or strain therein or produce objectionable audible noise or subharmonics of the operating frequency.
When the converter operates into a load of low acoustic impedance such as air, it is poorly coupled and will deliver little power thereto. Yet, when it operates into a stiff load of high acousitc impedance such as thick oil or solids and the acoustic coupling improves, because the motion is constant, great amounts of power are delivered into the load. Thus, the second major object of the invention is accomplished.
The required negative feedback control power generator for a converter employing electrostrictive elements may be accomplished in a great ,variety of ways.
FIGURE 6, the feedback error signal is utilized to control phase shift when the converter is connected in loop circuit with a power amplifier to form a self-running oscillator. This causes the converter to operate off its mechanical resonance when it is demanding a motional current greater than the desired maximum. A detailed circuit for doing this is shown in FIGURE 7.
In another embodiment of the invention the motional current error signal may be used to control the duty cycle of a power amplifier connected to drive the converter. This embodiment is shown in FIGURE 8. In this embodiment the motional current is not kept instantaneously constant. The time integral of the motional current is kept constant by adjusting the duty cycle of the system.
SPECIFIC DESCRIPTION Now referring to FIGURE 1, a high power density ultrasonic converter is generally indicated at 20. This is constructed in the manner detailed in the above-identified application of Jacke et al., Ser. No. 384,025. The converter 20 comprises a metal casing 22 supporting a perforated metal vent plate 24 integral with the transducer system generally indicated at 26. The transducer system is bolted together and comprises a pair of ceramic discs 28 separated by a metal plate 30. The ceramic discs 28 are backed by a massive metal back plate 32 and operate into a sonic energy concentrating step horn 34. Electrical energy is supplied to the transducer system 26 by means of a wire 36 connected to the metal plate 30 and the metal casing 22 connected to the vent plate 24. The converter also comprises a fan 38 for cooling'the transducer system 26.
Power for the transducer system 26 and the fan 38 is supplied via an insulated cable 40 from a power generator generally indicated at 42. The power generator 42 is connected via line cable 44 to a supply of 115 volts, 60 cycle power.
The converter 20 may be turned on by means of an on-off switch 46 or by means of a foot switch (not shown) which alternatively may be in series with the on-olf switch 46. A jewel light 48 may be provided to indicate that the power generator 42 is energized. A power level control 50 may also be provided.
The experiments discussed in the introduction above indicate that at a given operating frequency the converter 20 may be represented by the equivalent circuit shown in FIGURE 2. This circuit comprises a clamped capacitance C representing the frequency independent capacitance of the converter. The clamp capacitance C is in parallel with a resistance R representing the dielectric or voltage dependent losses in the converter.
A third parallel arm of the equivalent circuit comprises a frequency dependent reactive impedance X which may be represented by the series connected capacitance C and inductance L. The resistance R represents the internal motional dependent losses in the converter. The resistance R represents the transformed acoustical impedance of the load into which the converter 20 is operated. The resistance R has been found to be directly proportional to the acoustical impedance of the load over the useful operating range of the converter 20. The resistance R that is the motionally dependent internal losses, has been found to be substantially constant over when the converter operates into air is substantially zero. The total Q of the equivalent circuit is found to be approximately 800. When the converter is loaded, R may be as high as several thousand ohms. R is so large that it has not been accurately measured but appears to be greater than 100,000 ohms. It can therefore be neglected. C is approximately .0038 microfarad.
Thus, when operating at resonance, the current I through the clamped capacitance of C is constant. The current I through the dielectric resistance R is substantially zero. The motional current I is substantially dependent on R since at a constant frequency, the reactive impendance of X is zero and R is constant.
Thus, the total current I supplied to the converter may be vectorially added to the current passing through a capacitor having a value of .0038 microfarad connected across the source of the voltage V supplying the converter to derive the motional current I Methods for doing this are disclosed in detail in the above-identified application of Shoh. Ser. No. 265,751.
As disclosed in that application, the derived motional current I supplied to the converter may be utilized to provide a positive feedback signal for a power amplifier in the power generator 42 to cause the entire circuit to oscillate and thus the converter to operate at the natural mechanical resonance of the converter.
In many embodiments of the present invention the derived motional current I is utilized to provide a negative feedback signal for controlling the power supplied to the converter. Thus, the value of I is kept constant.
The invention further contemplates using positive feedback and negative feedback simultaneously to attain the two objects of operating the converter at its frequency of maximum conversion efficiency and keeping the motional current 1 constant.
When the motional current I is kept constant the amount of power delivered by the converter into a load will depend solely on the acoustic impedance of the load. That is, as the acoustic impedance of the load increases, R increases and the load dissipated power I R will increase. At the same time the internal motional losses in the converter I R will remain constant.
NEGATIVE FEEDBACK TECHNIQUES Gain controlled amplifier Now referring to FIGURE 3, in perhaps the simplest form of the invention to understand, a reference feedback error signal V is derived from I and used to control the gain of a linear amplifier 52. The linear amplifier produces a signal V This is supplied as the input to a power amplifier 54. The output of amplifier 54 is supplied to the converter 20 and to a compensating capacitor C equal in value to the clamped capacitance C of the equivalent circuit of the converter 20 shown in FIG- URE '2.
The current I passing through the capacitor C is equal to the current I in the clamped capacitance C of the equivalent circuit. The current passing through the converter 20 is equal to I plus I the current in the motional branch of the equivalent circuit. These currents are vectorially subtracted from one another at the summing point or summing network 56 to produce current I I is supplied to amplifier 58 and transformed into control signal V proportional thereto. V is subtracted from a fixed reference signal V at the summing point 60 to produce the negative feedback error signal V supplied to amplifier 52.
Thus, the signal V supplied to the power amplifier 54 has a magnitude inversely proportional to the deviation of I from a fixed value as determined by the reference signal V The frequency of operation is determined by the signal V which may be supplied by an oscillator or by a positive feedback signal derived from I by circuits disclosed in my above-identified application, Ser. No. 265,751.
The transfer functions of the amplifiers 52, S4, 58, are shown in FIGURE 3. As is well known in the art, the
' transfer function K of an amplifier is equal to the output signal divided by the input signal. Thus, the transfer functions of two amplifiers in series, such as amplifiers 52 and 54, multiply together. Therefore, the output V of the power amplifier 54 may be expressed by the equation:
It will be understood by those skilled in the art that if A, the loop gain of the system of FIGURE 3, is very much greater than the maximum total load R -I-R on the converter that I will be held substantially constant by automatic changes in the voltage V Power amplifier 54 is powered from a conventional power supply 62 connected to power line.
VOLTAGE CONTROLLED POWER SUPPLY In the embodiment of the invention shown in FIGURE 4, the voltage V supplied to converter 20 is controlled by means of a voltage controlled power supply 64. That is, as indicated by the transfer characteristic of the conventional power amplifier 54, the output voltage V thereof is proportional to the voltage V supplied by the voltage control power supply 64. This in turn is proportional to the error signal V derived from the motional current by means of a clamped capacitor C, summing point 56, amplifier 58, reference voltage V and summing point 60 in the same manner as that described with reference to FIGURE 3.
Again the frequency of operation of the circuit is determined by the frequency of the signal V from an oscillator or a feedback circuit. It is assumed that this signal is suificiently large to saturate the power amplifier 54 at all times. The system of FIGURE 4 can also be characterized by the equation V =A (BI where A is the loop gain K K K and is much greater than the maximum of R +R Now referring to FIGURE 5, alternatively, the power amplifier 54 may be linearly responsive to an input signal V supplied by a driver amplifier 68 under control of the voltage controlled power supply 64. In this case the driver amplifier 68 is supplied with a signal V from an oscillator or feedback circuit sufiicient to saturate the driver under all conditions and its output V is proportional to the control voltage supplied by the power supply 64.
Again the system may be characterized by the equation V =A (BI where A is the loop gain K K K K and is very much greater than the maximum of R +R Negative feedback controlled phase in a positive feedback loop In a preferred embodiment of the invention shown in FIGURE 6, the power amplifier 54 is connected in a positive feedback circuit with the converter 20, clamped capacitance C and summing network 56 so that it operates at its natural resonant frequency in the manner fully disclosed in the above-identified copending application, Ser. No. 265,751.
Additional elements provided by the present invention are a voltage control phase shifter 72. This controls the net phase shift in the loop circuit and thus the frequency of operation of the converter 20, the converter operates at its natural mechanical resonance when the total phase shift is zero. It operates off resonance if the total phase shift is different from zero. Amplifier 58 converts the motional current I to a control voltage V which is subtracted from a reference voltage at summing point 70 to produce an error signal V in the manner described with reference to FIGURES 3, 4, and 5. The error signal V controls the phase shift of the voltage control phase shifter 72.
Summing point 70 has the property of producing a negative error signal V only when the control signal V is greater than V When the control signal V is less than V indicating less than the maximum permissible motional current I no control voltage is produced from the summing network 70.
Thus, if the motional current I is less than the preselected maximum value, the converter 20 will operate at its natural mechanical resonance since the total phase shift in the loop circuit, including the voltage control phase shifter 72 and power amplifier 54, will be zero. If the motional current attempts to increase beyond the preselected value an error signal V will be created causing the phase shifter 72 to add a phase shift The output of the voltage control phase shifter I is equal to I shifted by the phase angle When V is less than or equal to V the angle o is zero. When V is greater than V the angle p is different from zero.
The phase shifter 72 can be chosen to add a positive or a negative phase shift depending on whether it is de sired to shift the operating frequency of the converter up or down in frequency from its natural mechanical resonance. In terms of the system shown in FIGURE 6, this choice is arbitrary but in practical situations it may be desired to shift the operating frequency in one direction or another because of the presence of nearby undesirable resonances of the converter.
The output of the power amplifier 54 is a constant voltage V of varying frequency to insure that the mo tional current I in the converter is substantially constant. t
The magnitude of voltage V may be chosen to be just below the maximum voltage permissible across the piezoelectric elements of the converter 20. 7
As indicated in the drawings, the frequency E of the voltage V is a function of the angle 11). Therefore by substitution:
Since the function of a constant is a constant, this equation characterizing the system may be reduced to:
That is, the frequency of operation is a function of the motional current I the function being dependent upon the Q of the converter 20. A=K K the loop gain of the system and is chosen to be very much greater than the maximum value of R +R A detailed circuit diagram of a system according to FIGURE 6 is shown in FIGURE 7.
Referring to FIGURE 7, the ultrasonic power generator generally indicated at 42 is connected to a source of 115 volts, 60 cycle A.C. potential by line plug 74. When on-olf switch 46 is closed, current is supplied through a fuse 76, switch 77 and the primary of a power supply transformer 78. The secondary of the transformer 78 produces 40 volts A.C. applied across a conventional pair of solid state rectifiers 80 to produce at midpoint between them a positive DC. voltage of approximately 15 volts.
The power supply includes a one millihenry inductor 82 and a series connected 8,000 microfarad capacitor 84 connected between +V and V The filtered positive 20 volt B+ terminal 86 is connected in the emitter circuits of ten parallel connected transistors 88 of type 2N2076 (only one of which is shown) through ten .2 ohm resistors 90 (only one of which is'shown).
Terminal 86 is similarly connected to drive ten more type 2N2076 parallel connected transistors 92 (only one of which is shown) through ten emitter resistors 93 (only one of which is shown). The bases of transistors 88 and 92 are connected to opposite sides of the secondary of an input transformer 94. The collectors of tran sisto-rs 88 and 92 are connected to opposite sides of the primary coil of an output transformer 96. The negative supply terminal 98 of the power supply is connected, as shown, through a network comprised of resistors 100 and 102 and capacitor 104 into the ganged push-pull amplifier circuit of transistors 88 and 92.
Resistor 100 is 47 ohms, resistor 102 is 1 ohm and capacitor 104 is 200 microfarads. The primary and secondary turns ratio of transformer 94 is 20 to 1. The primary to secondary turns ratio of transformer 96 is 12 to 200. Both coils of transformer 94 and the primary of transformer 96 are center tapped, as shown. The amplifier is conventional.
The secondary of the output transformer 96 is connected in circuit with the converter 20 through primary coil 106 of transformer 108 and the upper half 110 of the primary of transformer 94.
A compensating capacitor 112 is connected in parallel with converter 20 through a second primary 114 of transformer 108 and the lower half 116 of the primary of transformer 94.
Thus the total current through the converter I passes through the upper half 110 of the primary of transformer 94 and the clamped capacitance current I passes in series bucking relation through the lower half 116 of the primary of transformer 94. This produces a current in the secondary of transformer 94 proportional to the motional current I in the converter 20. Capacitor 112 is .0038 microfarad, the same as the clamped capacitance C shown in FIGURE 2.
The reactance of the load comprised of the converter 20 and compensating capacitor 112 is compensated for by connecting a .04 microfarad capacitor 118 across the secondary of the transformer 96 and opening the core of transformer 96 to produce a net inductance in the secondary thereof. The net inductance of the secondary of transformer 96 and the capacitor 118 are tuned to the operating frequency of the converter 20.
Thus, amplifier 42 is provided with a narrow bandpass characteristic so that operation of the system far off the desired operating frequency is prohibited.
Connected across the upper half 110 of the primary transformer 94 is a capacitor 120 and a saturable core reactor 122. A 16 microfarad D.C. blocking capacitor 124 is connected in this circuit for purposes that will become apparent hereinafter.
Reactor 122 is conventional employing an Allen-Bradley ferrite core type W04. When no direct current is passing through it, it has an inductance of .5 millihenry. Capacitor 120 has a value of .068 microfarad.
The circuit of inductor 122, capacitor 120 and transformer 94 is tuned to 27 kilocycles. This compensates for the net phase shift in the transistor circuits and causes converter 20 to operate at its natural resonance near 20 kilocycles.
When a DC. current passes through saturable reactor 122, its inductance decreases thus producing a net phase shift in the feedback circuit to cause the converter 20 to operate up to several hundred cycles below its natural resonance.
For the purpose of deriving the motional current I the total current through the converter 20 passes through primary 106 and the compensating capacitor current passes through primary 114 of transformer 108. Primaries 106 and 114 are connected in bucking relation to produce a current in secondary 126 directly proportional to I This is converted to a voltage signal by means of the series connected resistor 128 and variable resistor 130 connected across secondary 126. Resistor 128 is ohms and variable resistor has a maximum value of 500 ohms. Resistor 130 may be connected to power level control 50 shown in FIGURE 1 to set the desired motional current level.
The AC. motional voltage across resistors 128 and 130 is rectified in a peak detector comprising diode 132, resistor 134 and capacitor 136.
A 5 volt reference voltage is produced across Zener diode 138 from +V terminal of the power supply. This plus 15 volts is supplied through a second peak detector comprised of diode 140, resistor 142 and capacitor 144. Thus the voltage appearing at the base of transistor 146 is plus 5 volts when there is no motional current and less than 5 volts depending on the motional current I which is converted to a voltage subtracted from that across the Zener.
The diodes 132 and 140 are both types IN2071, resistor 134 is 1 kilohm, capacitor 136 is 10 microfarads, resistor 142 is 200 ohms and capacitor 144 is 50 microfarads. Transistor 146 is a type 2N350A and is connected in compound transistor configuration with transistor 148 which is type 2N2076. This configuration provides high current gain.
As long as the base of transistor 146 is positive I will be zero and the converter will continue to operate at its natural resonance. When the motional current I exceeds a preset value producing negative 5 volts D.C. across resistor 134, compound transistors 146 and 148 will conduct. A DC. current I will then pass through saturable reactor 122 reducing the inductance thereof. This changes the net phase shift in the positive feedback loop circuit of the amplifier and the converter. The converter will then operate at a lower frequency bringing I down to the desired preset value.
The inductance of inductor 122 can, in the circuit of FIGURE 7, change by a factor of 3 to cause converter 20 to operate up to several hundred cycles below its natural resonance. The circuit of FIGURE 7 is capable of delivering up to 400 watts of power into a loaded converter formerly limited to approximately 100 watts.
DUTY CYCLE MODULATOR Now referring to FIGURE 8, in another embodiment of the invention the power amplifier 54 is supplied by an input voltage V sufficient to saturate it. The voltage V is modulated by a duty cycle modulator 150 at a fixed frequency. The on time T of the duty cycle modulator is proportional to the voltage error signal V The frequency of the input V to the modulator 150 is the same as the natural resonance of the converter 20.
Thus, as the modulator alternatively turns on and off, the oscillations in the converter 20 will build up and decay and I will similarly vary. In this embodiment of the invention the time integral of I over one cycle or period P of the modulator 150 is kept constant. To this end summing network 56 produces I by subtracting current through capacitor C from the total current passing through the converter 20. I is supplied as the input to an amplifier 152. The output V thereof is equal to a constant K multiplied by the integral of I over one period of the duty cycle modulator 150.
V is compared with reference voltage V at summing network 60 to produce a voltage error signal V for controlling the duty cycle modulator 150. The on time T of the duty cycle modulator 150 is equal to its transfer function K times the voltage error signal V Thus the on time T of the voltage V may be expressed by the function P T =A (B -f 1M) where A is the loop gain KqKg and B is V /K The integral as indicated is taken over one complete period or cycle P of the duty cycle modulator.
Again the system may be characterized by the equation D =A[Bf(l where D0 is the controlled electrical quantity T the on time of modulator 150.
When the converter 20 is short circuited during the off time the operation of the system is illustrated in FIGURES 8a and 812. When the total motional resistive impedance of the converter 20 is low and the oscillations and thus the envelope if I build up toward and decay from a high steady state I as indicated at 154 the duty cycle modulator turns on V for but short periods.
Conversely, as illustrated in FIGURE 8b, when the total impedance of the converter 20 is high, that is, it is operating into a stiff load, and the oscillations build up toward and decay from a low steady state I the on time of the supply voltage V is increased so that the total area or time integral of I at 158 in FIGURE 8b remains the same as that of the envelope 154 of I shown in FIGURE 8a.
As will be apparent to those skilled in the art, the shape of the envelope of I 154 in FIGURE 8a and 158 in FIGURE 8b will vary if the converter is open circuited during the off time or if amplifier 54 has a significant internal impedance. Furthermore, since the power dissipated in the converter 20 is proportional to the RMS of 1 perfect control thereof requires that the transfer function K of amplifier 152 be expressed by the equation:
V I0 1M2 SUMMARY OF THE INVENTION It will thus be seen that I have provided methods, apparatus and systems for keeping the dissipation in a high power density sonic converter, due to motion therein, substantially constant by keeping an electrical quantity derived from the power supplied to the converter constant. In the case of sonic converters employing electrostrictive elements, this electrical quantity may conveniently be the motional current in the converter.
The systems that I have provided that are most efficient in keeping the motional current constant employ negative feedback derived from the motional current in a sonic converter employing electrostrictive elements.
It will be obvious to those skilled in the art that the systems, circuits, methods and apparatus disclosed herein may be modified in many ways for use in practical sonic converter systems specifically designed for particular working conditions. It will also be obvious to those skilled in the art that the methods, apparatus and systems disclosed herein may be modified for use with sonic converters employing magnetostrictive elements.
It will further be noted that the detailed systems disclosed herein all employ positive feedback derived from the motional current in sonic converters employing electrostrietive elements for controlling the frequency of op- 12 eration of the converter in the manner disclosed in my above-identified application Ser. No. 265,751, of which this application is a continuation-impart.
It will further be seen that in the present application I teach the use of an amplifier system connected in such positive feedback circuits having a narrow band pass characteristic centered about the desired operating frequency of the converter with Which it is connected. Furthermore, I provide novel means for causing the converter, when initially turned on, to operate off its normal operating frequency so that as oscillations build up in the converter it approaches its operating frequency from either a lower or a higher initial frequency of operation in order to avoid operation of the converter at an undesired nearby mechanical resonance.
It will thus be seen that the objects set forth above, among those made apparent from the preceding description, are efliciently attained and, since certain changes may be made in the above methods, systems, apparatus and circuits Without departing from the scope of the invention it is intended that all matter contained in the above description or shown in the accompany drawings shall be interpreted as illustrative and not in a limiting sense.
It is also to be understood that the following claims are intended to cover all of the generic and specific features of the invention herein described, and all statements of the scope of the invention, which, as a matter of language, might be said to fall therebetween.
Having described my invention what I claim as new and desire to secure by Letters Patent is:
1. Apparatus of the class described comprising, in combination:
(a) an electroaconstic sonic converter having first and second power input terminals;
(b) electrical power means connected to said first terminal;
(c) a reactive element having one side connected to said first sonic converter terminal;
(d) summing means connected to the other side of said reactive element and to said second sonic converter terminal for deriving in response to the currents through said reactive element and said sonic converter a control signal proportional to the motional current component in said sonic converter; and
(e) controlling means connected between said summing means and said power means and responsive to said control signal for controlling the electrical energy supplied to said sonic converter accordingly such as to maintain the power dissipated in said sonic converter substantially. constant.
2. The combination defined in claim 1 wherein said controlling means includes:
(1) means for comparing said control signal with a reference signal to produce an error signal,
(a) said error signal being supplied to control said power means.
3. The combination defined in claim 1 which further includes:
(f) feedback circuit means supplying said control signal to said power means such as to control the frequency of the energy supplied to said converter.
4. Apparatus of the class described comprising, in
(a) a piezoelectric sonic converter having first and second power input terminals;
(b) electrical power supply means including a power amplifier connected to said first converter terminal;
(c) a capacitor having one side connected to said first converter terminal;
(d) summing means connected to the other side of said capacitor and to said second converter terminal for deriving in response to the currents through said capacitor and said converter a control signal pro- 13 portional to the motional current component in said converter; and a (e) controlling means connected between said summing means and said power amplifier and responsive to said control signal for controlling said power amplifier such as to maintain the power dissipated in said converter substantially constant. 5. Apparatus of the class described comprising in (a) said control signal being supplied to said power amplifier as a positive feedback signal, and
(2) said controlling means comprises a signal controlled phase shifter responsive to said error signal for controlling the phase of said control signal supplied to said power amplifier.
7. The combination defined in claim 6 wherein said phase shifter includes a saturable core inductor connected 10 in a positive feedback circuit for said control signal; said comparing means includes a Zener diode connected in circuit to provide a fixed reference voltage; said summing means comprises a current transformer; and said controlling means further includes electronic valve means supplying a direct current to said saturable core inductor proportional to said error signal.
(a) an electroacoustic sonic converter having first and second power input terminals;
(b) electrical power supply means connected to said first terminal;
(0) a reactive element having one side connected to said first sonic converter terminal; 15
(d) summing means connected to the other side of said reactive element and to said second sonic converter terminal for deriving in response to the currents through said reactive element and said sonic References Cited UNITED STATES PATENTS converter a control signal proportional to the 2 motional current component in said sonic convert r; 0 2,799 787 7/1957 Guttl'lel 310-8.1 (e) means connected to said summing means for eom- 2,752,512 1/1956 sarfatt 3l08.] paring said control signal with a, reference signal to 2,812,485 11/1957 Schlebfil' 318448 produce an error signal; and 2,852,076 9/1958 Y 25036 (f) means connected between said comparing means 25 2,917,691 12/ 1959 Deprl co 318-1l8 and said power supply means and responsive t id 3,121,534 2/1964 W SOII 239102 error signal f0! accordingly controlling the frequency 2 1/1967 De burgt 3 l8116 f electrical energy supplied to said converter such 3,311,842 3/1967 Beck 33166 as to cause said converter to operate 01f its resonant frequency when the power dissipated in said con- 30 verter reaches a predetermined maximum.
6. The combination defined in claim 5 wherein:
(1) said power supply means includes a power amplifier,
J D MILLER, Primary Examiner US. Cl. X.R.