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Publication numberUS3560976 A
Publication typeGrant
Publication dateFeb 2, 1971
Filing dateAug 21, 1968
Priority dateAug 21, 1968
Also published asDE1942678A1, DE1942678B2, DE1942678C3
Publication numberUS 3560976 A, US 3560976A, US-A-3560976, US3560976 A, US3560976A
InventorsFoldes Peter
Original AssigneeRca Corp
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Feed system
US 3560976 A
Abstract  available in
Images(4)
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Claims  available in
Description  (OCR text may contain errors)

P. FOLD ES FEED SYSTEM Feb. 2, 197i A TORNA Y Feb. 2, 1971 P. FOLDES 3,560,976

FEED SYSTEM Filed Aug. 2l, 1968 4 Sheets-SheetI 2 ab( f/ TO QECEIVEL TERMINAL.

/ -f To BLEvnT/ou maw :um bl INVENTOR PETEIL FoLDt-:5

BY M) Mm ATTO/WIEV P. FOLDES FEED SYSTEM Feb. 2, 1971 4 Sheets-Sheet 3 Filed Aug. 21, 1968 Feb. 2, 1971 P, FOLDES 3,560,976

FEED SYSTEM Filed Aug. 21, 1968 4 Sheets-Sheet 4 INVENTOH PETEL FoLoes y bel/)w06 .United States Patent Oce Patented Feb. 2, 1971 3,560,976 FEED SYSTEM Peter Foldes, Montreal, Quebec, Canada, assignor to RCA Corporation, a corporation of Delaware Filed Aug. 21, 1968, Ser. No. 754,373 Int. Cl. H04b 7/10; H01q 13/02, 15/24 U.S. Cl. 343-100 17 Claims ABSTRACT OF THE DISCLOSURE A feed system suitable for satellite communication earth station antennas and which includes a three port polarizer assembly, multimode horn assembly and tracking difference mode circuitry is described. The three port polarizer assembly is coupled to the horn assembly for providing full polarization mode fiexibility for the sum modes. The multimode horn assembly provides excitation and control of both the sum and difference modes and inhibits the undesirable modes. The tracking difference mode circuitry is coupled to the multimode horn to process the difference modes and provide the azimuth and elevation information for monopulse tracking.

BACKGROUND OF THE INVENTION This invention relates to microwave communication systems and more particularly to a wideband multimode monopulse antenna feed system with high gain to noise temperature ratio which provides the simultaneous functions of receiving, transmitting and automatic tracking, wherein the first two are accomplished by the use of sum signals and the third is accomplished by using difference mode signals.

Feed systems capable of generating and receiving microwave energy in a plurality of modes have been developed and are known as multimode feed systems. 'Such multimode feed systems are often used in monopulse tracking wherein the energy transmitted and received by the feed systems is combined in such a manner that sum and difference radiation patterns are produced during transmission and/or reception. These patterns are analyzed to determine the position of the object which may be either an aircraft, a missile, or a satellite or to provide automatic tracking of that aircraft, missile, or satellite. Monopulse tracking systems are discussed in Introduction to Radar Systems by Merrill L. Skolnick, published 1962 by Mc- Graw-Hill Book Co. and Introduction to Monopulse by D. R. Rhodes, published in 1959 by McGraw-Hill Book Co.

The typical tracking feed system may include several horns or apertures. When only a small number of horns, such as in the four horn system, are used, the radiation patterns have undesirable characteristics which lower the efciency of the system and have increased noise level. Some attempts have been made to produce an efficient low noise multimode feed system with a single aperture. The prior art single aperture devices although operative have a relatively low gain to noise temperature ratio when they are used as feed systems for reflector type antennas and when operated over a wide range of frequencies. In addition, the prior art multimode feed systems do not have full polarization mode exibility in either the sum (communication) channel or the difference (tracking) channel.

Although many feed systems have been built in the past in which one or more of the three functions of transmit, receive and tracking are handled independently by separate parts of the feed, efficient systems with operational simplicity are required with a single aperture in the region of the focus of the primary or secondary reectors serving simultaneously for all three functions over a relatively wide range of frequencies. Moreover, to obtain the best field distribution efficiency, an undivided radiating source aperture is desirable as it is achieved by a single horn. Also, since the polarization (linear or circular) may vary from satellite to satellite, and the attitude of the linear polarization of the satellite may vary, a full polarization mode flexibility is therefore desirable for optimum gain in the sum mode and for tracking. Also, the receive and transmit polarization in practical systems must be maintained orthogonal (right vs. left circular or orthogonal linear polarization) in order to match the polarization modes of the feed to the employed polarization modes of the satellite.

It is an object of this invention to provide an improved multimode feed system operable over a relatively wide range of frequencies.

BRIEF DESCRIPTION OF INVENTION In accordance with the teachings of the present invention, a full polarization mode flexibility monopulse multimode feed system is provided wherein the feed system includes a mode coupler having a sum mode coupling port aperture and a separate plurality of difference mode port apertures symmetrically located about the sum port. A sum or communication chanel propagation network is coupled to the sum port and is responsive to the energy supplied to the sum port for providing the first plurality of sum modes. Likewise a separate monopulse tracking network including monopulse bridge circuitry is coupled to the plurality of difference mode ports and is responsive to the energy supplied to these ports for providing circularly polarized difference mode signals for the second plurality of modes. The novel feed system also includes a horn and mode filter for controlling the amplitudes and phase relationships between the first and second plurality of modes in a single aperture.

DESCRIPTION OF A PREFERRED EMBODIMENT A more detailed description follows in conjunction with the following drawings wherein:

FIG. 1 is a block diagram of the antenna feed system in accordance with one preferred embodiment of the present invention,

FIG. 2 is a sketch of the mode coupler used in accordance with the present invention,

FIG. 3 is a diagram of the wanted modes in the multimode horn,

FIG. 4 is a block diagram illustrating phase relationships for right circular polarization,

FIG. 5 is a perspective view of the mode filter,

FIG. 6 is a sketch of the side view of the mode lter,

FIG. 7 is a perspective view of the horn,

FIG. 8 is a perspective View of an orthogonal coupler,

FIG. 9 is a sketch of the side view of a 90 differential phase shifter in accordance with the present invention,

FIG. l0 is a sketch of the end view of the differential phase shifter of F-IG. 9,

FIG. 11 is a sketch of the rotary polarizer assembly, and

FIG. 12 is a fragmentary view in section of the rotary joint taken in the general direction of arrow A.

Referring to FIG. 1, a wideband tracking `feed system is provided including a multimode horn assembly 11, a three port polarizer assembly 12 and tracking mode circuitry 13. The composite feed assembly -for a satellite communication network is required, for example, to cover a wideband 5925 to 6425 rnHz. frequency band for transmit and a 3700 to 4200 mHz. frequency band for receive. The three port polarizer assembly 12 maintains propagation of the transmit or receive sum communication signals in orthogonal relationship while the monopulse difference signals are processed at the tracking mode circuitry 13. The multimode horn assembly 11 includes a mode coupler 16, a mode filter 17 and a single aperture horn 18.

FIG. 2 is an end view of the mode coupler 16 looking from the mode filter 17 toward the coupler 16. The mode coupler 16 is constructed of a large conically shaped port at the center of a square waveguide section and includes eight symmetrically located identical waveguide ports 26 through 33. The ports 26 through 33 in corner pairs form four circularly polarized exciting subapertures `with each port being orthogonally polarized with respect to its corner paired port.

The center conically-shaped transducer port 25 is, for example, for the above stated frequency band about 2.12 inches in diameter at the receiver equipment end 34 of the mode coupler and increases to about 3 inches in diameter at the antenna horn end 35. A step 36 is formed at the end of the conical port 25 between the 3 inch diameter conical at the end and a 4 inch square waveguide section at the mode lter end of the mode coupler 16. The step 36 is formed by the edge surface of the coupler 1-6 extending in front of and in a plane forward of the plane including the openings for the ports 26 through 33 and the end opening 35 for the conical coupler 25. The length of the conical, center transducer port 25 in this example is about 4.5 inches. The step 36 excites in the horn in addition to the basic TElo mode applied to the mode coupler 16, the TE30 mode as well as a small amount of T1312, TM12, TE21 and TM21 modes. Refer to FIG. 3 for a diagram of the wanted TEN and TEBO modes.

The eight symmetrically located identical waveguide ports 26 through 33 which are used for monopluse tracking terminate at the step 36 at which the TE30 mode excitation occurs. For an understanding of the monopulse tracking, the total step aperture 36 of the mode coupler 16 is considered for discussion as divided into four subapertures labeled a, b, c and d as shown in FIG. 4. Each corner of subaperture a, b, c and d of the mode coupler 16 has a respective horizontally polarized port 28, 27, 32, 31 and a respective vertically polarized port 29, 26, 33 and 30. Also, the power in each subaperture is equal as provided by the short-slot hybrids 44 through 47 as illustrated in the arrangement shown iu FIG. 1 and 4.

To provide circular polarization in the feed Ifor monopulse tracking which is adaptable to receive or transmit right or left circular polarization or linear polarization, the phase relationship between the -four subapertures a, b, c and d of the step aperture 36 should remain the same. However, each aperture a, b, c and d should support the TEM mode (horizontal polarization) and the TEM, mode (vertical polarization) and these should be either delayed or advanced relative to` each other to excite a circular polarized electromagnetic signal for each subaperture.

The monopulse tracking circuitry for providing the difference modes for either linear polarization or right or left circular polarization is illustrated in FIGS. l and 4. The circuit consists of six short-slot hybrids 44 through 47, 50 and 53 and two magic tee hybrids 51 and 52 and various bends in the waveguides and waveguide sections. The sizes and dimensions of these waveguides and `waveguide sections are optimized at the receiver tracking frequency bands. Referring to FIGS. l, 2 and 4, each of the eight paired ports, 26 through 33, is coupled to its orthogonal paired port at one of the yfour shortslot hybrids 44 through 47. One terminal of each of the hybrids 44 through 47 is terminated at terminals 55 through 58 respectively, and the other terminal is combined for example, with the output of another one of the hybrids 44 through 47 at hybrid 50 or 53. The proper ports of the hybrids 44 to 47 to be terminated depends upon the polarization, right or left circular polarization. For right circular polarized signals, waveguide section 48 is used with output terminals 14 and 15 coupled to hybrid 50 and waveguide section 49 is used with output terminals 19 and 20 coupled to hybrid 53. For right circular polarized signals as shown in FIG. l, output terminals 1, 3, 5 and 7 of hybrids 44 through 47 are terminated and are unused, and terminals 2, 4, 6 and 8 are used and are connected to like numbered terminals in waveguide sections 48 and 49. For left circular polarized signals, terminals 2, 4, 6 and 8 are terminated and terminals 1, 3, 5 and 7 and waveguide sections 59 and 60 are used. The input terminals 1 and 3 of waveguide section 59 are connected to the like numbered terminal in hybrids 44 and 45 and output terminals 14 and 15 of waveguide section 59 are connected to hybrid 50. The input terminals 5 and 7 of waveguide section 60 are connected to the like numbered terminal of hybrids 46 and 47 and the output terminals 19 and 20 of waveguide section 60 are connected to hybrid 53. Also, the roles of the output terminals 61 and 62 change from elevation difference modes to azimuth difference modes and vise versa as indicated in FIG. l with respect to the use of right or left circular polarized signals. Each of the four hybrids 44 through `47 which are coupled to the corner paired ports 26 through 33 provides a power division and provides a 90 relative phase shift between paired ports. The combined system of the four subapertures each with the 90 relative phasing of the vertically polarized T1310 mode and the horizontally polarized TEM mode with proper phasing between subapertures a, b, c and d provides the vertically polarized TE20 mode and horizontally polarized hybrid HE11=TEH+TM11 modes in the mode filter 17, `which modes, when they are superimposed provide the circularly polarized azimuth difference mode system. See FIG. 3 showing these wanted modes.

Referring to FIG. 4 and considering, for example, for right circular polarized azimuth difference mode, the short-slot hybrids 50 and 53 and magic tees 51 and 52 are arranged so that the phase relationships at each port of the subapertures are as shown. The phase marked after the symbol A in FIG. 4 shows the relative phases in the azimuth difference mode system. The properly phased (a and d in effective 0 phase, b and c in effective 180 phase) and vertically polarized TEM, modes in the subapertures a, b, c and d excite the vertically polarized TE20 mode. The similarly phased and horizontally polarized TEM, modes in the subapertures a, b, C and d wherein a and d effective phase is 0 and b and c is effective 180 phase excite the horizontally polarized hybrid mode. Since the physical polarization or the direction of the electric field in subaperture b is opposite that of a and the physical polarization direction of subaperture c is opposite that of subaperture d and the ports a. and d and c and d are physically located on opposed ends of that field, the subapertures a and d in effect are cophased at 180 and subapertures b and c in effect are cophased at phase. The simultaneous presence and the 90 phasing of the vertically polarized T1320 mode and the horizontally polarized hybrid HE11=TE11+TM11 mode provides the circularly polarized difference mode with zero radiated field in the vertical plane (horizontal or azimuth difference mode).

The sarne combined system of the four subapertures with an orthogonal set of proper phasings provides the vertically polarized hybrid HE11=TMH+TE11 mode and horizontally polarized TEOZ mode in the mode filter, which modes, when they are superimposed provide the circularly polarized elevation difference mode system. The phase relation between the circularly polarized subapertures a, b, c and d is determined by the remaining four hybrids of the system which are two short-slot hybrids 50 and 53 and the two magic tees 35 and 52.

Referring to FIG. 4 and considering, for example, the right circularly polarized vertical difference mode, the hybrids 50 and 53 and magic tees 35 and 52 are arranged so that the pshae relationships at each port of the subapertures a, b, c and d are as shown. The phase marked after the symbol E or elevation shows the relative phases in elevation difference mode of the subapertures for the combined circularly polarized difference mode comprising the combined horizontally polarized TE02 mode and vertically polarized T Elfi-T M11 modes. See FIG. 3 showing these wanted modes. In this example at subaperture a the horizontal polarization E or elevation is 0 phase, at subapertures b and d the horizontal polarization is 180 phase and at subaperture c the horizontal polarization is 0 phase relative to the elevation terminal 61.

Since the physical polarization direction or the direction of the electric field in subaperture b is opposite that of subaperture a and the physical polarization direction of subaperture c is opposite that of subaperture d, therefore the effective electrical phase at subaperture b is 0 phase and at subaperture c is 180 phase. By means of subapertures a and b being 0 effective relative phase and subapertures c and d being 180 effective phase between the subapertures for E or elevation system, the vertically polarized hybrid HEn mode and the horizontally polarized T E02 modes are excited. The simultaneous presence y and 90 phasing of the vertically polarized HEM mode and the horizontally polarized TE02 mode provides the circularly polarized elevation difference mode. Each triangle in the FIG. 4 indicates the phase between the hybrids, `whereas the phase in the subapertures is indicated by the number following E or elevation and A or azimuth phase. The phase with respect to either the azimuth marked A or elevation marked E indicates the phase of the corresponding terminal at the input of the eight ports of the four subapertures a, b, c and d for right circular polarization. Operation for left circular polarization is similar. The roles, however, of the feed terminals 61 and 62 are interehanged when the polarization in the difference mode changes from right circular polarization as shown in FIG. 4 to left circular polarization and thus therefore the azimuth difference mode channel becomes the elevation difference mode.

The tracking mode circuitry at this point on in the direction of the receiver and processing circuitry network for combining the difference mode signals conforms to standard monopulse bridge arrangements such as found on p. 178 of Introduction of Radar Systems mentioned previously.

Referring back to FIG. 2, the center conical sum mode transducer port is specially dirnensioned and formed having an inner ring 37 which is in the example described above about 1/s inch thick and about 3 inches from the horn end of the port 25 in order to cut off the propagation of the generated difference modes (hybrid HE11=TE11+TM11 and TE20) back into the sum system and thus avoid the deterioration of the difference mode axial ratio. Each of the eight difference mode ports has tuning stubs 40 therein for suppressing the unwanted TEN, TMm, TEM and TMm modes in the throat of the horn and to reduce the coupling between the sum and difference ports and between the various difference mode ports. The higher order difference modes generated by theeight element ring array after being refiected and modified by the conical sum mode transition in the mode coupler are finally filterd and phased by the mode filter 17 and horn 18 to produce the desired difference mode aperture distribution and primary patterns.

With just the four subapertures a, b, c and d providing circularly polarized waves radiating directly into space with the phasing between the ports of the suby apertures, the feed system would degenerate to a circularly polarized four horn monopulse system. However, since the mode coupler radiates into a single horn instead of space, the system becomes a single horn multimode system which has a more efficient monopulse aperture distribution than a four horn system.

A wideband tracking feed system required for satellite communication networks shall cover, for example, the 5925 to 6425 mHz. band for transmit and the 3700 to 4200 mHz. band for receive operation. The proper widet band phasing and control of the described modes for such cases is provided by mode filter 17 and horn 18. The mode filter 17 is placed between the mode coupler 16 and the horn 18 wherein the communication (sum) and tracking (difference) mode signals are combined in a single aperture. Referring to FIGS. 5 and 6, the mode filter 17 is a square semi-exponential waveguide section having a ridge 41 on each of the four sides 42 to excite a wideband of frequencies. The mode filter 17 shown in FIGS. 5 and 6 has approximately a half cosine function profile along the axis of propagation, a square cross section and symmetrically located ridges 41 in its two main planes. This shape or any such appropriately tapered variation such as the half cosine function profile is arranged so that it results in a minimum square cross section between the two terminating flanges of the mode filter 17. The half cosine function profile is only one example. Any one of many appropriately tapered variations along the axis of propagation may be selected wherein the selection of the length l of the mode filter and the rate of change in width w or shape of the profile (such as the amplitude of the abovedescribed cosine function) is arranged so the frequency dependence of the phase between the TEZO, TEOZ and hybrid HE11=TE11+TM11 difference modes on one hand and the frequency dependence of the phase between the TEm and TESU sum modes on the other hand is virtually eliminated, and thus a very wide band multimode operation can be achieved. The square cross section of the mode filter 17 has a minimum width w of typically 2x and a minimum length l of 5A where A is the geometrical means of the limits of the nearly one octave frequency band represented by the lower receive and higher transmit frequencies mentioned in the above example. By the dimensions of the filter being so selected, only the desired higher order sum and difference modes can propagate through this filter section and the phase between these wanted modes assures low side lobes in the E and H planes in the primary pattern and leaves the axial ratio for the circularly polarized sum and difference modes unchanged. The exact dimensions of the above-described mode filter for each particular case using the above-given rules is easily achieved by anyone skilled in the microwave antenna art. The output cross section of the mode filter is made large enough so that only a small amount of a differential phase shift takes place between the modes in the multimode horn itself. The basic TEW and the higher order T1330 sum modes generated by the step in the mode coupler 16 to the four inch cross section as filtered and phased by the mode filter and horn finally produce the sum mode aperture distribution and patterns. The mode lter 17 and the square semiexponential waveguide section 18 follow in such a manner that the generated higher order modes arrive at the aperture of the horn in the proper phase. The term proper phase means that the total aperture field of the horn is more or less axially symmetrically tapered in the complete, for example, 500 mMz. receive frequency band.

Referring to FIG. 7, the final shaping and the frequency stabilizing of the aperture field of the horn is achieved by the shape of the horn 18 which has an increase in the fiare angle at the end of the horn. This increased flare angle produces a relatively large quadric phase error in the transmit frequency band and a smaller quadratic phase error in the receive band thus bringing the transmitter band beamwidth even closer to the receiver band beamwidth. The horn 18 also has fins 21 along the fiared portion on each of the four sides to approximate even more a Gaussian shaped output pattern. The horn 18 also has a square waveguide section 22 in each of the four corners to increase the speed of the traveling waves in the extra fiare section at the low end of the frequency band and doing so to provide a more uniform phase front at the aperture for low frequencies.

Thus far, we have concerned ourselves with the processing of the tracking mode signals and not the communication signal. Referring back to FIG. 1, the sum mode signals at the mode coupler 16 are fed to a three port polarizer section 12 through the conical center port 25. The three port polarizer includes an orthogonal coupler 71, a fixed 90 differential phase shifter 73 and a rotating 90 differential phase shifter 75.

Referring to FIG. 8, the orthogonal coupler 71 is constructed of a square waveguide section 80 having a ridge 81 on the inside surface of each of the sides 82 to excite a wideband of frequencies. The coupler 71 acts to combine the transmit signals from transmit terminal 70 and the receive signals toward receive terminal 72 to a common square waveguide dimensioned and ridge loaded to propagate both signals. The transmitted signals are, for example, in the TEN, mode and the receive signals are, for example, in the orthogonal TEM mode. Three internal irises 83 or narrow metal strips are provided Within the orthogonal coupler 71 at the transmit end of the coupler. These irises are placed orthogonally to the electric field of the TEN mode of the transmitter. The TEW mode signals from the transmitted are coupled toward the first fixed phase shifter 73 with negligible reflection at the irises 83 and the orthogonal receive signals from the first fixed phase shifter 73 in the TEM mode are reflected at irises 83 in the side arm of the orthogonal coupler 71 to the receiver terminal 72. In this manner, these irises do not interfere with the transmitted wave, but act as a 90 elbow for the waves coming to receiver terminal 72 through the side arm 84.

The first fixed phase shifter 73 illustrated in FIGS. 9 and 10 is a square waveguide section having a ridge 86 on each of the four sides to excite wideband of frequencies and is the next element in the direction toward the horn 18 from the orthogonal coupler 71. This square waveguide section is capable of propagating signals in both the TEN mode, for example, for the transmitter signals, and the orthogonal TEM mode for the receiver signal. The first phase shifter 73 has fifteen irises 88 in this above-mentioned example in the lower right-hand corner of the waveguide when viewing the waveguide in the direction of propagation and has fifteen irises 88 in the upper left-hand corner of the waveguide. The series of corner irises 88 act in such a manner that an incoming wave with polarization parallel to the side wall of the square waveguide such as the TEN or T1201 mode is split into two equal power orthogonal components and a nominally 90 differential phase shift takes place between the two components by the time these components are propagated through the phase shifter 73. The end result of the power split and differential phase shift is that the unit converts the incoming linearly polarized wave into a left circularly polarized wave or right circularly polarized wave for a vertically or horizontally (transmit or receive respectively) polarized wave respectively. For this combination of polarization, the irises 88 in the phase shifter 73 are in the upper leftand lower righthand corners, when one views the three port polarizer from the transmit input terminal. For the proper wide band operation of these irises y88 a post 90 has to be placed in the middle of each iris. These posts 90 resonate in the upper part of the frequency band for instance at 6.2 gHz. in the above example and the irises 88 resonate in the lower part of the frequency band for instance at 4 gHz. in the above example. In this manner their combined impedance remains nearly constant as a function of frequency and wide band operation is obtained. The residual mismatch effects of these strips or irises are cornpensated by tuning screws 89 which can be used also to further optimize the axial ratio produced by the phase shifter in the transmit and receive frequency band in the above example.

Since the orthogonal coupler 73 assures orthogonality Ibetween the transmitted and received waves, the corresponding received polarization is right circularly polarized or left circularly polarized.

The next unit in the three port polarizer towards the antenna is the rotating polarizer assembly 75, see FIG. 1l. The assembly consists of a rotary joint 74 with a square to circular waveguide transition 96 and then a circular to square waveguide transition 97, a rotating 90 differential phase shifter 75 and a second rotary joint 76 with a square to circular waveguide transition 105 and a short section of a circular waveguide 106 coupled to the center port 25 of mode coupler 16.

Referring to FIGS. ll and 12, the first rotary joint 74 has a square to circular waveguide transition section 96, a circular to square waveguide transition 97 and a pair of coaxial rotary flanges 98 and 99l wherein the gap 109 between the rotating flanges of the transitions is about 0.025 inch. The coaxial fianges 98 and 99 are located coaxially with both transitions and cover the gap between the transitions. Each of the combined rotary joint coaxial flanges 98 and 99 has two grooves 100 and 10.1 within the inner circumference of the rotary joint coaxial fiange cover wherein the distance from the end of the groove `100 or 101 in the rotary joint to the end edge of the transitions is half a wavelength (M2) at the transmitter frequency for one groove and for the other groove is half a wavelength (M2) at the receiver frequency. The grooves 100 and 101 therefore act to provide two chokes, one resonant at the receiver frequency to refiect back into the waveguide the received signal and the other to reflect back the transmitted signal to thusly reduce radiation losses at the rotary joint.

The first rotary joint 74 acts with respect to the circularly polarized transmitted wave as a double transformer with a square to circular waveguide transition and a circular to square waveguide transition connected back to back. The output terminals of the first rotary joint 74 have a square waveguide cross section supporting the properly phased TEM, and TEM modes. The center of the rotary fiange 98 is circular. The diameter of the circular section of the waveguide is about the same as that of a side of a square so as to assure propagation of the circular TEH mode. The output of the first rotary joint 74 is coupled to a rotating 90 differential phase shifter 75 which is electrically identical to the first fixed 9 differential phase shifter 73 in FIGS. 9 and l0, having therein the 30 irises and posts with l5 of each in the appropriate two corners.

The rotating polarizer assembly accepts the circularly polarized waves from the fixed phase shifter 73 transforms these waves back to linear polarized waves at the outputs of the rotary joint 76. The attitude of the output polarization is now however an orientation function of the rotating polarizer and the orthogonal relationship between the receive and transmit band waves remains at all attitudes. The output of the 90 differential phase shifter 75 toward the horn is fed to a second rotary joint 76 similarly built as shown and described in connection with FIG. l2 with the chokes resonating at the receive and transmit ybands respectively. The output of the second rotary joint 76 is applied to the 2.125 inch diameter center circular waveguide port which is part of the mode coupler 116. Since the output polarization is always parallel to one side of the square waveguide, the output terminal of the fixed 90 polarizer and the position of the rotating polarizer determines the orientation of the linearly polarized transmitted waves. The rotatable 90 polarizer may be rotated using a motor 110 and gears 111 and `112 to align the sum mode circuitry with the linear polarization of the satellite. The attitude of the linear polarization is a function of the angle between the two phase Shifters. The original orthogonal relationship however between the transmit and receive frequency band waves remains in all the attitudes. By removing the second phase shifter 75 and the two rotary joints 74, 76 from the system, right or left circularly polarized signal operation is obtained for use with the transmission and reception of circular polarized signals at the horn.

What is claimed is: 1. A monopulse multimode feed system for propagating energy supplied thereto over a wide band of frequencies and comprising:

a mode coupler having a sum mode coupling port and a separate plurality of difference mode ports,

first means coupled to said sum port and responsive to said energy supplied thereto for providing a first plurality of modes having at least one state of polarization,

second means coupled to said separate plurality of difference mode ports and responsive to said energy supplied thereto for providing a second plurality of modes which are circularly polarized,

and third means coupled to said mode coupler including au energy propagating horn having a single aperture for controlling the amplitudes and phase relationships over said wide band of frequencies between the energy in said first and second pluralities of modes for providing radiation patterns Iwith low side lobes in the E and H planes in the primary pattern and for maintaining the axial ratio for the circularly polarized difference modes.

2. The monopulse feed system as claimed in claim 1 wherein said sum mode coupling port is conically shaped.

3. The monopulse feed system as claimed in claim 1 wherein said mode coupler has a square waveguide cross section at the end of the coupler in the direction of said horn.

4. The monopulse feed system as claimed in claim 3 wherein said mode coupler includes means at said sum mode coupling port and within said square waveguide cross section whereby said first plurality of modes including the symmetrical TEN and TE modes and TEU and TM21 and TEM and TMm modes are excited into said third means.

5. The monopulse feed system as claimed in claim 4 wherein said separate plurality of difference mode ports are orthogonally paired and are located symmetrically about said sum mode port.

6. The monopulse feed system as claimed `in claim S 10 wherein said difference mode ports are eight in number and wherein a pair of said difference mode ports is orthogonally positioned at each of the four corners of said square waveguide cross section.

7. The monopulse feed system as claimed in claim 6 wherein said second means includes four differential phase shifter and power splitters with a different one of said phase shifter and power splitters coupled to each pair of said orthogonally paired waveguide ports to provide circular polarization in said feed.

8. The monopulse feed system as claimed in claim 7 wherein each of said 90 differential phase shifted and power splitters is a short-slot hybrid.

9. The monopulse feed system as claimed in claim 8 wherein said second means includes means for properly phasing said paired waveguide ports relative to each other to excite TE20, T1502, and hybrids T Ell-j-TMH modes in the output of the mode coupler and to provide simultaneous presence and 90 phasing of the vertically polarized TE20 mode and horizontally polarized hybrid T Ell-f-TMH mode whereby a circularly polarized difference mode with a null plane in one plane is obtained and to provide the simultaneous presence and 90 phasing of the horizontally polarized TE02 mode and vertically polarized hybrid TEll-i-TMH mode whereby a circularly polarized difference mode with a null plane in a plane orthogonal to said one plane is obtained.

10. The monopulse feed system as claimed in claim 9 wherein said means for properly phasing said paired waveguide ports includes a fifth and sixth hybrid with the output of a first one of said hybrids combined with the output of a second one of said hybrids at said fifth hybrid and with the output of a third one of said hybrids combined with the output of a fourth one of said hybrids at said sixth hybrid.

`11. The monopulse system as claimed in claim 1 wherein said third means includes a mode filter having a single square aperture cross section and wherein the length and the change in width of said square aperture cross section of said mode filter is determined so as to control the frequency dependency of the phasing of the TElo and TEBO modes over said wide band of frequencies to assure low side lobes in the primary pattern and to control the frequency dependency of the phasing of the T1520, TEM and hybrid HE11=TEH+TMH modes over said wide band of frequencies to maintain the axial ratio for the difference modes.

12. The monopulse feed system as claimed in claim 11 wherein the minimum width is on the order of 2A and its length is on the order of 5k, where A is the geometrical means of the limits of an operating band of said wide band of frequencies.

13. The monopulse feed system as claimed in claim 12 wherein said square waveguide section has a half cosine function profile.

14. The combination as claimed in claim 1 wherein said horn is a square pyra-midal horn having a flare angle close to the aperture of the horn that increases substantially so as to produce a large quadratic phase error at the higher frequencies of said wide frequency band to bring the higher frequency band beamwidth close to the lower frequency band beamwidth.

15. The combination as claimed in claim 6 wherein said conically shaped port has an inner iris ring in order to cut off the propagation of the generated difference mode back into the sum system.

16. The combination as claimed in claim 6 wherein each of the eight difference mode ports has tuning stubs therein for suppressing the unwanted TE12, TM21 and TEzl-l-TMIZ modes and to reduce coupling between the sum and difference ports and between the difference ports.

17. The combination as claimed in claim 1 wherein said first means includes:

a first 90 differential phase shifter responsive to linearly polarized signals applied thereto and converts the linearly polarized waves into opposite sense cir- References Cited Cularly polarized waves, UNITED STATES PATENTS a second 90 differential phase shifter coupled between 3 277 485 10/1966 Howard 343 16(SD)X the first phase shifter and the sum ports responsive 311516 3/1967 Packard 343 100(.3) to said circularly polarized waves present at its input 5 3,353,183 11/1967 Giger 343 16(SD)UX to convert the circularly polarized waves present at its input into linearly polarized waves, and RICHARD A- PARI-EY, Primary Examiner means for rotating one of said phase Shifters relative T H TUBBESING, Assistant Examiner to the other where the attitude of the linear polarization is a function of the angle between the two posi- 10 US C1' XR' tions of the two phase Shifters. 343-16, 756, 772, 783, 786

Referenced by
Citing PatentFiling datePublication dateApplicantTitle
US3696434 *Jan 15, 1971Oct 3, 1972Radiation IncIndependent mode antenna feed system
US4507665 *Jun 9, 1982Mar 26, 1985Thomson-CsfPrimary source with frequency re-utilization
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US4885593 *Sep 18, 1986Dec 5, 1989Scientific-Atlanta, Inc.Feeds for compact ranges
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US20050200548 *Mar 3, 2005Sep 15, 2005Fusco Vincent F.Single aperture monopulse antenna
US20100238086 *Feb 22, 2010Sep 23, 2010Electronics And Telecommunications Research InstituteDouble-ridged horn antenna having higher-order mode suppressor
CN101841081A *Mar 17, 2010Sep 22, 2010韩国电子通信研究院Double-ridged horn antenna having higher-order mode suppressor
EP0068940A1 *Jun 4, 1982Jan 5, 1983ALCATEL THOMSON FAISCEAUX HERTZIENS Société Anonyme diteFrequency reuse primary active antenna
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Classifications
U.S. Classification342/352, 343/783, 342/188, 343/786, 343/756, 342/153, 343/772, 342/365
International ClassificationG01S13/00, H01Q25/00, H01P1/17, G01S13/44, H01Q13/00, G01S3/14, H01P1/165, H01P5/02, H01Q25/04, H01Q13/02
Cooperative ClassificationH01Q25/04, G01S13/4409, G01S3/146
European ClassificationH01Q25/04, G01S3/14C, G01S13/44B