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Publication numberUS3577147 A
Publication typeGrant
Publication dateMay 4, 1971
Filing dateSep 8, 1969
Priority dateSep 8, 1969
Publication numberUS 3577147 A, US 3577147A, US-A-3577147, US3577147 A, US3577147A
InventorsHannan Peter W
Original AssigneeHazeltine Corp
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Phased array antenna having a wave speeding ground plane
US 3577147 A
Abstract  available in
Images(4)
Previous page
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Claims  available in
Description  (OCR text may contain errors)

[5 6] References Cited [2]] App No zrsnsnhmtgwn, N.Y. UNITED STATES PATENTS 221 Filed 3,259,902 7/1966 3,277,488 10/1966 Hofiman Haultine Corporation Primary E.xamr'nerEli Lieberman Attorney-Kenneth P. Robinson Peter W. Harman Sept. 8, 1969 Assignee VII-UV vmwwu l l.

[72] Inventor [45] Patented May 4,1971

ABSTRACT: A phased array antenna consisting of a plurality of radiating elements associated with a ground plane and including a reactive impedance for reducing the catastrophic ef- 343/778, fect produced by end-fire grating lobes and surface waves. A 343/785, 343/846, 343/91 1 series capacitance or shunt inductance is placed in the trans- Int. mission path formed by the ground plane and free space in 343/753, order to increase the phase velocity of any TM wave that 853, 854, might propagate along the ground plane so that it is at least 909, 91 1 equal to the phase velocity of a free space wave [54] PHASED ARRAY ANTENNA HAVING A WAVE SPEEDING GROUND PLANE 13 Chins, 18 Drawing Figs. [52] US. [5 l] ...............H0lq 13/00 [50] Field of 770, 771, 785, 778, 739, 846,847, 868,

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PATENTED MAY 41971 SHEET 2 OF 4 1 CONDUCTNE GROUND PLANE cououcnve GROUND PLANE Fl G. 30

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CONDUCT'VE GROUND PLANE FIG. 60

PATENTEU HAY l97| MEI t Of 4 9022.. mmzmmadm FIG. 9b

. FIG. 9d

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PHASED ARRAY ANTENNA HAVING A WAVE SPEEDING GROUND PLANE BACKGROUND OF THE INVENTION A phmed array antenna consisting of a plurality of radiating elements is capable of radiating a narrow beam of electromagnetic energy which is steerable in space. The beam is radiated in the desired direction by adjusting the phase of the power coupled to each radiating element in a manner that causes the contribution from each element to be in phase along a phase front which is perpendicular to the desired direction of radiation. However, in addition to being in phase in the main beam the contributions from each element may also be in phase at, at least, one other angle thereby tending to form a spurious beam referred to as a grating lobe.

The angular relationship between the main beam and the grating lobe is dependent on the spacing between the radiating elements, specified in wavelengths. For an array substantially larger than a wavelength, if the spacing between the elements is small enough, less than 0.5A where A is the wavelength of the propagating energy, the grating lobe will not exist in real space. However, there are usually other design criteria which require that the spacing between the elements be greater than 0.5a and the existence of grating lobes is usually a factor which must be considered in the design of phased array antennas.

While grating lobes are generally undesirable since they reduce the power radiated in the main beam and can produce spurious responses, these disadvantagesare of a minor nature as compared to the catastrophic effect which grating lobes can produce in phased array antennas which consist of slots or holes in a metal ground plane or otherwise consist of radiating elements asociated with a ground plane. Although this type of array is very desirable from a mechanical and environmental standpoint, it has a basic defect in electrical performance. When the main beam is steered sufficiently far from broadside, the grating lobe appears in the end-fire direction, the direction'of radiation coplanar with the ground plane, producing a drastic reduction in radiated power. Substantially all the power is reflected from the array and there is no significant power radiated.

There are many instances when the ground plane departs from a simple flush configuration such as when the ground plane is covered with dielectric material in order to provide environmental protection for the radiating elements. The departure from a flush configuration usually results in the propagation of surface waves along the ground plane. At certain scan angles surface wave propagation can cause an almost total reflection of radiated power.

The scan angle at which this complete reflection due to sur-, face wave propagation occurs is often less than the scan angle in which propagation of the grating lobes in the end-fire direction causes almost complete reflection of the energy. The scan angle is the angle between the direction of the main beam and the broadside direction.

The catastrophic reflection of power, whether caused by end-fire grating lobes or surface waves is a limitation on the maximum scan angle of the array and accordingly a serious limitation on the performance of the array.

It is, therefore, an object of the present invention to provide new and improved phased array antennas which substantially reduce or eliminate the problems of end-fire grating lobes and surface waves.

It is a further object of this invention to provide new and improved phased array antennas which include a reactive impedance in the transmission line formed by the ground plane and free space in order to eliminate the drastic effect produced by surface waves and end-fire grating lobes.

In accordance with the present invention, there is provided in a phased array antenna wherein at certain scan angles there may be produced a spurious TM wave along the ground plane which can cause substantially all the power to be undesirably reflected by the array resulting in a drastic reduction of radiated power, the combination comprising: a plurality of radiating elements for cooperatively producing a beam of electromagnetic energy which is capable of being steered in space by the coupling of signals of differing phase to the radiating elements; a ground plane associated with the plurality of radiating elements for preventing substantial radiation behind the array; and means for providing a reactive impedance in the transmision path formed by the ground plane and free space for increasing the phase velocity of a TM wave which tends to propagate along the ground plane to a phase velocity at least equal to the phase velocity of a free space wave; whereby the range of scan angles over which there is no substantial reflection of power resulting from propagation along the ground plane is increased.

BRIEF DESCRIPTION OF THE DRAWINGS For a better understanding of the present invention together with other and further objects thereof, reference is had to the following description taken in conjunction with the accompanying drawings, and its scope will be pointed out in the appended claims:

FIGS. la and lb illustrate one embodiment of the present invention;

FIG. 2 is a graphical representation of the radiation characteristics of a typical phased array;

FIGS. 3a and 3!; illustrate the radiation characteristics of a single element over a ground plane;

FIG. 4a illustrates the equivalent circuit of a transmission line formed by the ground plane of a phased array and free space for a wave traveling along the array surface;

FIG. 4b illustrates the equivalent circuit of such a transmission line having a series capacitance;

FIG. 5 illustrates the effect of a series capacitance on the equivalent transmission line; FIGS. 60 and 6b illustrate another embodiment of the present invention;

FIGS. 7a and 7b illustrate another embodiment of the present invention;

FIGS. 8a and 8b illustrate another embodiment of the present invention;

FIGS. 9a and 9b illustrate a further embodiment of the present invention which introduces shunt inductance into the transmission path formed by the ground plane and free space, and

FIG. 10 is another embodiment of the present invention which introduces shunt inductance into the transmission path formed by the ground plane and free space.

DESCRIPTION OF THE INVENTION FIGS. la and lb illustrate one embodiment of the phased array antenna constructed in accordance with the present invention. FIG. la is a partial sectional view looking in at an edge of the antenna. FIG. lb is a partial front view of the antenna. The antenna may be considered to be an infinite array extending in all directions. The only difference between such an array and a practical array are the boundary conditions which are not relevant to the present invention.

The FIG. I antenna includes a plurality of radiating elements illustrated as loop radiating elements 10 for cooperatively producing a beam of electromagnetic energy which is capable of being steered in space by the coupling of the signals of different phase from signal generator 11 to each of the radiating elements 10. The manner in which signals of variable phase, coupled from signal generator ll, cooperate to produce a narrow steerable beam is well known in the art.

The antenna also includes a ground plane including a planar conductive surface illustrated as conductive sheet 12 associated with the plurality of radiating elements 10 for preventing substantial radiation behind the array. Beside providing environmental protection, the ground plane directs the power from what would be an omnidirectional radiating element in the absence of the ground plane so that the element only radiates in front of the array thereby at least doubling the amount of available power in the hemisphere in front of the array. In those configurations where the gound plane is a simple flush configuration, such as where the radiating elements consist of holes or slots in a gound plane, the conductive sheet 12 is the ground plane. However. in those configurations where the ground plane departs from the simple flush configuration. such as when the metal sheet is covered by a dielectric layer, the ground plane is a complex structure and the conductive sheet 12. although it is an esential element, does not constitute the complete ground plane structure. The impedance of the ground plane is affected by the structure which causes it to depart from the simple flush configuration and such structure is therefore a part of the ground plane.

The antenna further includes means for providing a reactive impedance in the transmission path formed by the ground plane and free space for increasing the phase velocity of a TM wave which tends to propagate along the ground plane to a phase velocity which is at least equal to the phase velocity of a free space wave. This impedance is provided by dielectric slab 13 which is coextensive with conductive sheet 12, and thc plurality of conductive pins 14, which are inserted in holes which are regularly spaced along the surface of the dielectric slab l3, perpendicular to conductive sheet 12. Pins I4 are in a physical contact with conductive sheet 12. The combination of conductive sheet 12, dielectric slab 13 and conductive pins 14 constitute a ground plane having a series capacitance in the transmission path formed by the ground plane and free space which increases the phmc velocity of a TM wave which tends to propagate along the ground plane so that it is at least equal to the phase velocity of a free space wave without reducing the phase velocity of a TE wave which tends to propagate along the ground plane to less than that of a free space wave. This capacitive ground plane increases the range of scan angles over which there is no substantial reflection of power resulting from propagation along the ground plane.

THEORY AND OPERATION OF THE INVENTION A typical prior art phased array antenna consists of a plurality of slots or holes in a metal ground plane. As previously explained gating lobes customarily accompany the radiation of the main beam and when the grating lobe is in the endfire" direction. there is a substantial reduction of radiated power. This phenomenon is illustrated in FIG. 2 which illustrates the typical relationship between the transmission coefficient plotted along the ordinant with respect to the scan angle plotted along the abscissa. While FIG. 2 is based on values calculated for an infinite phased array it is equally applicable to a finite array having a large number of radiating elements. Curve A of FIG. 2 represents the radiation characteristics of a simple flush mounted array and illustrates the catastrophic reduction in radiated power that occurs when the gating lobe is in the end-fire direction at scan angle 0,. There is almost complete reflection of the power at that scan angle.

Curve 8 in FIG. 2 illustrates the effects of surface wave propagation caused by placing a dielectric sheet over the gound plane. At a scan angle which is less than the end-fire grating lobe angle there is a complete reflection of energy as indicated by the absence of radiation at that angle. It can be seen that either the propagation of surface waves or the endfire gating lobe limits the range of scan angles over which the antenna can be utilized with the effect of surface waves being potentially even more catastrophic than end-fire gating lobes.

'lhe drastic reduction in radiated power caused by either end-fire gating lobes or surface waves is a phenomenon associated with propagation along the E plane, that is to say the condition will only exist when the radiated beam has a componentlyingirrtheEplaneofthearrayandcurvesAandBof FIG. 2 are representative of scanning in the E plane. Scanning in the H plane does not ordinarily produce the catastrophic reduction of power due to either end-fire grating lobes or surface waves as is illustrated by curve C.

The reason why the end-fire grating lobe does not ordinarily produce the catastrophic effect of total reflection in the H plane is illustrated by FIGS. 30 and 3b which are a representation of the H plane and E plane radiation patterns, respectively, of a single element over a ground plane. As is shown in FIG. 30, there is no radiation along the ground plane for H plane scan while the element will radiate very strongly along the ground plane in the E plane. As a result the element does not couple to an end-fire grating lobe in the H plane but strongly couples to said end-fire grating lobe in the E plane.

Further where the ground plane departs from the simple flush configuration, the structure is also susceptible to the propagation of surface waves. Typically, prior art departures from a flush configuration resulted in the propagation of TM surface waves whereas the departure usually did not have sufficient effect in the H plane to produce TE surface waves.

In the E plane of scan, a wave propagating along the gound plane is excited only in the TM mode. TM wave propagation along the ground plane therefore corresponds to E plane propagation. Similarly, in the H plane of scan, a wave propagating along the ground plane is excited only in the TE mode. TE wave propagation along the ground plane therefore corresponds to H plane propagation. Therefore as illustrate by curve C in FIG. 2 there are no catastrophic reductions in radiated power in the H plane.

The FIG. I embodiment of this invention substantially reduces the drastic effect produced by end-fire grating lobes and surface waves by introducing a series capacitance into the transmission path formed by the ground plane and free space which increases the phase velocity of the TM wave which tends to propagate along the gound plane to a phase velocity at least equal to the phase velocity of a free space wave.

FIG. 4a illustrates the equivalent circuit of a transmission line formed by the ground plane and free space for a wave traveling along the array surface for an ordinary metal ground plane and is shown to consist of a series inductor L land shunt capacitor C. FIG. 4b illustrates the addition of a series capacitor C into this transmission line. the effect thereof being to increase the phase velocity of a wave which would tend to propagate along this transmission path. The increased wave velocity along the ground plane may be thought as causing a refracting effect which bends the wave away from the ground plane as illustrated in FIG. 5a. This results in a progressively weaker wave amplitude at the ground plane as the wave travels away from the radiating element. At a large distance from the element there is little or no radiation near the ground plane for any plane of polarization. The resulting radiation pattern for the slot is as illustrated in FIG. 5b which represents the E plane radiation pattern of a single element over a ground plane wherein series capacitance has been added to the transmission path formed by the gound plane and free space It will be noted that the radiation pattern in the E plane now is similar to the radiation pattern in the H plane and accordingly the element will not radiate along the ground plane in either plane of scan. in order to obtain the E plane as illustrated in FIG. 5b, sufficient series capacitance must be added so as to increase the phase velocity of a wave traveling in the E plane so that it is greater than the wave velocity of a free space wave.

Besides the elimination of the drastic effect of the end-fire grating lobe, increasing the phase velocity along the ground plane in the E plane also tends to eliminate the problems associated with the propagation of surface waves. Increasing the phase velocity along the ground plane in the E plane of scan so that it is just equal to the velocity of a free space wave will provide the substantial benefit that the angle at which total reflection due to surface wave propagation occurs. 6,, is increased to the angle at which the end fire grating lobe occurs, 9 thereby increaing the usable range of scan angles. Further increasing the phase velocity so that it is greater than the velocity of a free space wave completely eliminates the catastrophic effect of total reflection caused by either the endfire gating lobe or surface wave propagation.

Although the addition of a series capacitance to the transmission path formed by the ground plane and free space may eliminate the catastrophic effects due to end-fire grating lobes or TM surface waves in the E plane of scan it is possible that the addition of series capacitance to the transmission path 5 may cause effects in or near the H plane of scan similar to those eliminated in the E plane. For waves traveling in the H plane the series capacitance produced in a practical structure is likely to vary with scan angle in such a way as to become an open circuit at some angle thereby preventing radiation from the array at that angle. Furthermore, in a practical structure the dielectric or other loading is likely to have an effective thickness sufficiently great that a TE surface wave may exist in or near the H plane. Such a surface wave would create the same kind of drastic reduction of radiated power that the introduction of the series capacitance eliminated in the E plane of scan by preventing the propagation of TM surface waves.

The FIG. 1 embodiment provides the reactive impedance in the form of a series capacitance in the transmission path formed by the ground plane and free space that increases the phase velocity of a TM wave tending to propagate along the ground plane so that it is at least equal to the velocity of a free space wave without reducing equal to the velocity of a TM wave which tends to propagate along the ground plane to less than that of a free space wave thereby increasing the range of usable scan angles in the E plane without undesirably effecting the range of usable scan angles in the H plane. A TM wave incident on the face of the dielectric at any angle is constrained by pins 14 and made to travel in the direction perpendicular to conductive sheet 12. Pins 14 in effect provide a plurality of transmission lines which are short circuited by conductive sheet 12. As is well known a shorted transmission line provides a reactive impedance. At a distance between one-quarter and one-half wavelength from the shon circuit, the impedance is capacitive. Therefore, if the thickness of dielectric slab l3 and accordingly the length of pins 14 is between one-quarter and one-half wavelength the front surface 13' of dielectric slab 13 will present a capacitive impedance to a TM wave.

A TE wave incident on the face of the array has an electric field vector perpendicular to pins 14. If the diameter of pins 14 is small enough, they appear to be invisible to an electrical field vector oriented perpendicular to them. TE pins 14 have no substantial effect on the propagation of a the wave.

The dielectric constant K of the slab and thickness of the slab and accordingly the length of the pins L are determined by first determining the capacitive reactance X,., relative to a free space impedance Z that is required. As previously stated the amount of capacitance is related to the amount of wave speeding required. The phase velocity of the wave must be increased so that it is at least equal to the velocity of a free space wave and preferably so that it is greater than a free space wave. The amount of series capacitance to be introduced by the combination of dielectric slab 13 and metal pins 14 is accordingly determined by the reactive impedance of the transmision path that would exist absent this combination.

Having determined=the amount of capacitive reactance, X required, the values of L and K are determined by solving the following transcendental equation for gl mined and the freespace wavelength )t The actual values for K and L should be slightly less than those derived in accordance with the above equation in order to guarantee that a the surface wave is not created by the structure for waves in or near the H plane. There will still be an angle of radiation in the H plane where the structure places an open circuit in series with the equivalent transmission line, thereby preventing radiation. However, by using the values described above, this angle of nonradiation will be very close to the direction parallel to the array face, the end-fire direction, and planar phased arrays are not ordinarily operated near the end-fire direction.

The principle by which the FIG. 1 embodiment avoids the harmful effects possible for waves radiated in or near the H plane is that the structure appears to be an open circuit to an incident wave traveling in the H plane at an incidence angle very close to grazing incidence, the end-fire direction. For a certain value of K there is only one value of L which achieves this and vice versa as determined by equation (2). If K or L is too large, a the surface wave in and near the H plane can exist. If K or L is too small the array impedance will become an open circuit for H plane radiation at an angle not very close to grazing, and therefore closer to the operating scan angles of the array.

Metal pins 14 maintain the capacitive reactance of the ground plane substantially constant for all angles of radiation in the E plane. Since an incident wave in the E plane is guided by pins 14 into a path nearly perpendicular to the face of the structure, substantially constant path length is maintained for an incidence angle. As previously stated, this path length in dielectric material 13 is some value between a quarter wavelength and half wavelength depending on the desired capacitive reactance X,.

Although the FIG. 1 embodiment provides this series capacitance in the transmission path formed by the ground plane and free space so a to eliminate the end-fire grating lobe and surface wave problems in the E plane without affecting the H plane, the resultant antenna has an inherent limitation in that wire antennas tend to become impractical for radiating energy having a wavelength less than an inch. FIGS. 6a and 6!: illustrate a portion of a phased array antenna constructed in substantially the same manner as the FIG. I antenna but having narrow diameter waveguides as the radiating elements.

As in FIG. 1, selected ones of the holes in dielectric slab 13, positioned at regularly spaced intervals, have a larger diameter than the remainder of the holes. Within each of the larger holes is positioned a hollow conductive cylinder 15 which is in contact with conductive sheet 12. Although having a larger diameter than the diameter of pins 14 each cylinder 15 has an internal diameter less than that required to propagate the energy diameter from source 11 as an unloaded waveguide. However, as is well known a narrow diameter waveguide can be caused to propagate by "loading" the waveguide. Internally modifying the waveguide by filling the inside with high-k dielectric material 16 as illustrated in FIG. 6 or placing metal ridges inside the waveguide parallel to the electric field vector lowers the cutoff frequency of the waveguide so that it will propagate the lowest mode. This configuration provides a wave speeding effect for a TM wave without any deleterious effect on a TE wave.

The radiating elements in the FIGS. 1 and 6 embodiments have a narrow bandwidth. FIG. 7 illustrates another embodiment of a phased array antenna constructed in accordance with the present invention having a broader bandwidth. FIG. 7a is a partial sectional view looking in at the edge of the antenna while FIG. 7b is a front view of FIG. 7a. The antenna includes a plurality of radiating elements illustrated as waveguides 17 for producing a steerable beam as a result of the coupling of signals of variable phase from signal generator 11. The antenna also includes a ground plane including conductive sheet 12 associated with the plurality of radiating elements 17 for preventing substantial radiation behind the array.

The FIG. 7 antenna provides series capacitance in the transmission path formed by the ground plane and free space in substantially the same manner as the FIG. I and FIG. 6 cmbodiments with the principal difference that there are no pins between the radiating elements 17. In FIG. 7 the radiating elements, waveguides 17, provide an action similar to that of pins 14 in FIGS. 1 and 6. In this embodiment the waveguides have a larger diameter than the wageguides of FIG. 6 thereby permitting a better impedance match to frequency bandwidth. However, the larger waveguides have a loading effect on the ground plane similar to that ofan artificial dielectric. Therefore, the K of the real dielectric 13 between waveguides I7 must be reduced with respect to the value calculated in conjunction with the description of FIG. 1 in order to maintain the condition ofan open circuit to an incident wave traveling in the H plane at an incidence angle very close to grazing incidence.

FIG. 8 shows a still further embodiment of the present invention which provides a series capacitance in the transmis sion path formed by the ground plane and free space for increasing the phase velocity of a TM wave which tends to propagate along the ground plane so that it is at least equal to the phase velocity of a free space wave without reducing the phase velocity ofa TE wave to less than the velocity of a free space wave. In FIG. 8 the diameter of waveguides 18 has been enlarged with respect to the FIG. 8 embodiment so that their artificial dielectric loading effect provides the desired open circuit condition without any real dielectric material between waveguides l8.

Waveguides l8 serve a function analogous to that of pins 14 in the FIG. 1 embodiment, namely, that an incident TM wave is constrained by waveguides l8 and made to travel in the direction perpendicular to conductive sheet I2.

lfa series capacitance is to be presented to an incident TM wave as a result of a constraining effect produced by waveguides 18, the length L of each waveguide 18 above conductive sheet 12 must be between M4 and M2 where A is the freespace wavelength of the radiated energy. However, as will be more fully explained, flanges l9 permit the length L to be low than H4 and still introduce a series capacitance in the transmission path fonned by the ground plane and free space.

As indicated, each of waveguides 18 in FIG. 8 include a metal flange 19. As is more fully illustrated in FIG. 8b, flanges 19 are constructed so that a space exists between flanges 19 associated with adjacent waveguides. Any adjacent pair of flanges 19 therefore comprises a pair of conductors separated by a dielectric (air) and it is therefore apparent that flanges 19 provide a lumped capacitance to a TM wave incident on the ground plane.

Since flanges 19 provide a lumped capacitance which is in parallel with the reactive impedance produced by the constraining cffect provided by waveguides 18 the length L' of waveguides 18 may be somewhat less than M4. Any inductive reactance provided by the constraining effect of waveguides 18 would be overcome by the capacitance provided by flanges 19 to provide a resultant series capacitance in the transmission path formed by the ground plane and free space.

FIG. 9 is another embodiment of a phased array antenna constructed in accordance with the present invention which includes a plurality of radiating elements associated with a conductive sheet 12, each radiating element includes a waveguide ending in a hole in the conductive sheet 12. As previously explained, conductive sheet 12, absent any structure which disturbs the flush configuration, would constitute the ground plane. However, disturbance of the ground plane by the addition of the structure described below makes the ground plane a complex structure of which conductive sheet 12 is an essential part.

The antenna further includes a plurality of thin conductive wires 20 each attached at one end to conductive sheet 12 and arranged perpendicular to it at regular intervals for providing a shunt inductance in the transmission path formed by the ground plane and free space for increasing the phase velocity of a TM wave which tends to propagate along the ground plane to a phase velocity at least equal to the phase velocity of a free space wave without reducing the phase velocity of a TE wave which tends to propagate along the ground plane to less than that of a free space wave. As previously explained, increasing the phase velocity of a TM wave that tends to travel along the ground plane provides improved performance by reducing or eliminating the effect of grating lobes and surface waves. For the same reason that a series capacitance achieves this result the shunt inductance introduced by wires 20 provides the desired increase in phase velocity.

Since wires 20 are located across the transmission path formed by the ground plane and free space and are grounded by conductive sheet 12 they provide a shunt reactance to a TM wave traveling in that transmission path. If the length of wires 20 in front of conductive sheet 12 is between M4 and M2, the shunt reactance provided is inductive. The dimension 2] is therefore between M4 and M2. In order to provide the desired inductance the wires must be very thin with respect to a wavelength; i.e., in the order to 0.001). The spacing between wires 20 is dependent on the amount of inductance required.

Since the electric field vector of a TE wave traveling in the transmission path formed by the ground plane and free space is perpendicular to wires 20, they appear to be invisible to that vector and have no substantial effect on the corresponding TE wave.

It should be noted that the thin conductive wires 20, although in appearance similar in structure to pins 14 of the FIG. 1 embodiment, provide a shunt inductance since they are in front of the radiating elements and therefore in the trans mission path formed by the ground plane and free space while the pins 14 are behind the radiating elements.

FIG. 10 is another embodiment of the present invention which provides shunt inductance in the transmission path formed by the ground plane and free space in substantially the same manner as the FIG. 9 embodiment. The FIG. 10 embodiment consists of thin inductive wires 22 which are encapsulated in the dielectric material 23 and top loaded by the metal discs 24. Metal discs 24 collectively provide wide angle impedance matching as is fully described in the copending application of Peter W. Harman, Ser. No. 815,566, filed Apr. l4, I969 and entitled PHASED ARRAY ANTENNA INCLUD- ING IMPEDANCE MATCHING APPARATUS." Whereas in the above referenced application the metal discs were mounted on thin dielectric supporting discs in order to avoid the propagation of surface waves, discs 24 are mounted on solid cylinders of dielectric material 23 which contribute to the impedance matching efiect provided by discs 21. Solid cylinders of dielectric material 20 can be utilized since the thin conductive wires 22 increase the phase velocity so as to avoid surface wave propagation as described above.

In the present embodiment, besides providing the desired impedance matching, metal discs 24 serve to top load thin wires 22. Wires 22 can therefore have an actual length less than a quarter wavelength while having an eflective length between H4 and M2 thereby presenting a shunt inductance to a TM wave.

Although the inductive wires 22 in FIG. 10 are not directly connected to conductive sheet 12, as in FIG. 9, the insertion of the wire into the radiating element by approximately M4 makes it appear that the wire is connected to conductive sheet 12. The portion ofwire 22 in the waveguide, which ends at opening 10, forms a transmission path which is open circuited at end 25 of wire 23. A point a quarter wavelength from an open circuit in a transmission line appears to be a short circuit. Therefore, if point 26, being the point where the plane of conductive sheet 12 intersects wire 22, is a quarter wavelength from point 24, wire 22 will appear to be shorted to conductive sheet 12 or at least have a low impedance connection thereto, at that point. Wires 22 therefore provide a shunt inductance in substantially the same manner as wires 18 in FIG. 9.

While there have been described what are at present considered to be the preferred embodiments of this invention, it will be obvious to those skilled in the art that various changes and modifications may be made therein without departing from the invention and it is, therefore, aimed to cover all such changes and modifications as fall within the true spirit and scope of the invention.

l claim:

1. In a phased array antenna wherein at certain scan angles there may be produced a spurious TM wave along the ground plane which can cause substantially all the power to be undesirably reflected by the array resulting in a drastic reduction of radiated power, the combination comprising:

a plurality of radiating elements for cooperatively producing a beam of electromagnetic energy which is capable of being steered in space by the coupling of signals of differing phase to the radiating elements;

a ground plane associated with said plurality of radiating elements for preventing substantial radiation behind the array;

and means for providing a reactive impedance in the transmission path formed by said ground plane and free space for increasing the phase velocity of a TM wave which tends to propagate along the ground plane to a phase velocity at least equal to the phase velocity of a free space wave;

whereby the range of scan angles over which there is no substantial reflection of power resulting from propagation along the ground plane is increased.

2. in a phased array antenna wherein at certain scan angles there may be produced a spurious TM wave along the ground plane which can cause substantially all the energy to be undesirably refiected by the array resulting in a drastic reduction of radiated power, the combination comprising:

a plurality of radiating elements for cooperatively producing a beam of electromagnetic energy which is capable of being steered in space by the coupling of signals of differing phase to the radiating elements;

a ground plane including a planar conductive surface as sociated with said plurality of radiating elements for preventing substantial radiation behind the array;

and means for providing a series capacitance in the transmission path fomied by said ground plane and free space for increasing the phase velocity of a TM wave which tends to propagate along the ground plane to a phase velocity at least equal to the phase velocity of a free space wave without reducing the phase velocity of a TE wave which tends to propagate along the ground plane to less than that of a free space wave;

whereby the range of scan angles over which there is no substantial reflection of power resulting from propagation along the ground plane is increased.

3. In a phased array antenna the combination as claimed in claim 2 in which the means for providing a series capacitance includes a dielectric slab coextensive with said planar conductive surface having a plurality of regularly spaced holes perpendicular to said conductive surface and a plurality of thin conductive pins each positioned in one of said holes and in contact with said conductive surface at one end thereof, the thickness of said slab and the length of said pins being greater than ).,,/4 and les than AJZ, where A, is the wavelength of said electromagnetic energy in said dielectric.

4. in a phased array antenna the combination as claimed in claim 3 in which selected ones of said holes positioned at regularly spaced intervals have a larger diameter than the remainder of said holes and which additionally includes a plurality of hollow conductive cylinders each positioned in one of said selected holes and in contact with said conductive surface at one end thereof and in which said radiating elements consists of thin wire loops positioned in front of said dielectric slab and connected to a signal source through said hollow conductive cylinders.

5. In a phased array antenna the combination as claimed in claim 3 in which selected ones of said holes positioned at regularly spaced intervals have a larger diameter than the remainder of said holes and which additionally includes a plurality of hollow conductive cylinders each positioned in one of said selected holes in contact with said conductive surface at one end thereof, each cylinder having an internal diameter less than the diameter required to propagate said electromagnetic energy as an unloaded waveguide and in which each radiating element consists of a loaded waveguide including said conductive cylinders internally modified to propagate said electromagnetic energy.

6. In a phased array antenna the combination as claimed in claim 2 which includes a plurality of loaded metal wall waveguides comprising said plurality of radiating elements, each of said waveguides protruding from said planar conductive surface, the length of each waveguide above said conductive surface being between M4 and M2 where )t is the free space wavelength of the radiated energy.

7. in a phased array antenna the combination as claimed in claim 2 which includes a plurality of loaded metal wall waveguides protruding from said planar conductive surface comprising said plurality of radiating elements, each of said waveguides having a flange positioned between said conductive surface and free space, flanges of adjacent waveguides being in spatial relationship so as to present a lumped capacitance to an incident TM wave.

8. In a phased array antenna wherein at certain scan angles there may be produced a spun'ous TM wave along the ground plane which can cause substantially all the energy to be undesirably reflected by the array resulting in a drastic reduction of radiated power, the combination comprising:

a plurality of radiating elements for cooperatively producing a beam of electromagnetic energy which is capable of being steered in space by the coupling of signals of differing phase to the radiating elements;

a ground plane including a planar conductive surface associated with said plurality of radiating elements for preventing substantial radiation behind the array;

and means for providing a shunt inductance in the transmission path formed by said ground plane and free space for increasing the phase velocity of a TM wave which tends to propagate along the ground plane to a phase velocity at least equal to the phase velocity of a freespace wave without reducing the phase velocity of a TE wave which tends to propagate along the ground plane to less than that of a freespace wave;

whereby the range of scan angles over which there is no substantial reflection of power resulting from propagation along the ground plane is increased.

9. In a phased array antenna the combination as claimed in claim 8 in which the means for providing a shunt inductance includes a plurality of thin conductive wires oriented in the transmission path formed by said ground plane and free space substantially perpendicular to said planar conductive surface, connected at one end thereto and having an effective length between M4 and All, where A is the freespace wavelength of said electromagnetic energy.

10. [n a phased array antenna the combination as claimed in claim 9 which additionally includes dielectric material encapsulating and supporting said thin conductive wires.

11. In a phased array antenna the combination as claimed in claim 8 in which the means for providing a shunt inductance includes a plurality of thin conductive wires oriented in the transmission path formed by said ground plane and free space substantially perpendicular to said planar conductive surface, said radiating elements include metal wall waveguides terminating in openings in said conductive surface and which additionally includes a plurality of solid cylinders of dielectric material each positioned within one of said waveguides and having a low impedance connection at one end to said thin metal discs and at the opposite end to said conductive surface 13. In a phased array antenna the combination as claimed in claim 12 in which said thin wires project into said waveguides by a distance equal to M4.

Patent Citations
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US3277488 *Jul 27, 1964Oct 4, 1966Hoffman John GAntenna decoupling by means of a lossy dielectric slab
Referenced by
Citing PatentFiling datePublication dateApplicantTitle
US4536767 *Mar 17, 1983Aug 20, 1985Licentia Patent-Verwaltungs-GmbhMicrowave directional antenna employing surface wave mode
US5231409 *Jan 16, 1990Jul 27, 1993Societe Europeenne De PropulsionMicrowave antenna capable of operating at high temperature, in particular for a space-going aircraft
US6788957 *Sep 23, 1999Sep 7, 2004Sagem SaTwo-band transmitter-receiver with dual radiation device
US7889127 *Sep 22, 2008Feb 15, 2011The Boeing CompanyWide angle impedance matching using metamaterials in a phased array antenna system
US20100073232 *Sep 22, 2008Mar 25, 2010Soji SajuyigbeWide Angle Impedance Matching Using Metamaterials in a Phased Array Antenna System
Classifications
U.S. Classification343/778, 343/785, 343/846, 343/911.00R
International ClassificationH01Q3/30
Cooperative ClassificationH01Q3/30
European ClassificationH01Q3/30