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Publication numberUS3593138 A
Publication typeGrant
Publication dateJul 13, 1971
Filing dateJul 31, 1968
Priority dateJul 31, 1968
Publication numberUS 3593138 A, US 3593138A, US-A-3593138, US3593138 A, US3593138A
InventorsAllen Walter K, Dunn James G, Fletcher Irving Lane, Krause Irving A, Miller John E
Original AssigneeNasa
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Satellite interlace synchronization system
US 3593138 A
Abstract  available in
Images(14)
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Claims  available in
Description  (OCR text may contain errors)

United States Patent [72] Inventors James G. Dunn 3,351,858 [1/1967 Jowett et 325/15 Montclair; 3,363,180 H1968 Geissler 325/4 Irving A. Krause, Nutley; Irving Lane 3,418,579 12/1969 Hultberg 325/58 Fletcher, East Rutherford, NJ.; Walter K. 3.430237 2/1969 343/5 X fig i g John Mluer Primary Examiner-Benedict V. Safoure d R. F. K f pp Na 749,121 Attorneys G T McCoy an omp [22] Filed July 31, 1968 [45] Patented July 13, I971 [73] Ass'gnee The p? as ABSTRACT: A master station and a plurality of slave stations was A by ,Admmmraor of the include synchronization equipment to enable each of the stam and tions to have access to a common repeater in a different time slot of a time division multiplex format at the repeater, there being motion between the stations and the repeater. The [54] SATELLITE INTERLACE SYNCHRONIZATION master station propagates a sync burst through the repeater. SYSTEM Each of the stations receives this sync burst from the repeater 14 Claims 20 Drawing Figs and adjusts the frequency of the timing signals therein to compensate for the doppler shift experienced in the propagation U.S. path from each of the stations to the repeater so that the 325/39' 1525/58: 343/5 DP, 343/75 343/179 desired frequency of the timing signals is present in the re- [5 l 1 i peaten Each of the slave stations also propagates diffe ent low SMH: DP, 7, power level psuedo noise code ranging signal through the re. 7 1005A? 325/4, 15, USA1513 38, 39; peater back to itself which is used to adjust the phase of the 179/15 A; 8 6 X timing signals digitally and in an analog manner by means of a motor-driven phase shifter to account for the changing range [56] References Cmd between the repeater and each of the slave stations so that a UNITED STATES PATENTS data burst of each of the slave stations appears in the proper 3,320,61 l 5/1967 Sekimoto et a]. 34317.5 time slot of the time division multiplex format at the repeater.

COMMON REFEATER (Moe/L 6 0R Flxea) i I; i MASTER S(.AV SLAVE s m 7/ ON srq now 1 STAT/0M *a (Moan. 5 (M05 IL (M O8/LE on F1 X60) on FIXED) on nxio) I 5 L A V5 51. A VE sm no $734 7/0 "7 (M08 IL 5 ,(Moa/z a on nxso on FIXED) PATENTED JUL] 3191:

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PATENTED JUL 1 3 Ian sum on or 14 U s xmxouwy m PATENTEU JUL 1 3 um SHEET 07 0F 14 PATENTEU JUL 1 3 I971 SHEET 09 1 14' SHEET 12 0F 14 PATENIED Jun 31% PATENTED JUL] 31971 SHEET 130F 14 SATELLITE INTERLACE SYNCHRONIZATION SYSTEM BACKGROUND )F THE INVENTION This invention relates to communication systems and more particularly to communications systems wherein a plurality of stations gain access to and communicate through a common propagation media, such as a common repeater.

Multiple access communication systems have been utilized for many years to achieve multiple access to long-distance telephone trunk systems. In addition, this multiple access technique is applicable to other communication systems including, but not necessarily restricted thereto, (1) supervisory control systems to enable supervision, from a fixed common repeater, or from a central station through the common repeater, of the activities of a plurality of mobile stations, (2) remote control systems to enable control, from a fixed common repeater, or from a central station through the common repeater, of various responsive devices contained in a plurality of mobile stations, (3) communication systems to establish, maintain and/or enable communication between a fixed common repeater, or a communication center coupled to the fixed common repeater, and a plurality of mobile stations, such as is necessary between an airport control tower and a plurality of airliners, and between a dispatcher communication center and a fleet of taxicabs, emergency vehicles and cargo carrying trucks, and (4) a communication satellite system to enable a plurality of fixed ground stations to utilize a common repeater carried by an orbiting satellite.

In providing the multiple access for the various systems above set forth, different techniques have been employed in the past. One such technique is the so called random access technique to enable a plurality of stations to have access to and communicate through a common repeater on an undefined basis, namely, a random basis. Another such technique to permit achieving of multiple access is in the employment of frequency division multiplex techniques wherein each of the plurality of stations employs a different carrier signal and wherein the common repeater has the bandwidth to handle all of the different frequency carriers and the intelligence carried thereon. Still another technique enabling multiple access to a common repeater has been by the employment of time division multiplex techniques wherein each of the plurality of stations are assigned to, or are capable of selecting, a time slot in time division multiplex frame or format at the common repeater to thereby permit communication through the common repeater in a noninterferring relationship.

In multiple access systems employing time division multiplex techniques it is mandatory that there be a strict time synchronization so that each of the plurality of stations transmits its intelligence in a different one of a plurality of time slots of a time division multiplex format and be so confined to that time slot selected for a particular station that its communication will not interfere with communications of other stations in adjacent time slots of the format.

The multiple access systems employing time division multiplex techniques have used both analog modulation, such as pulse amplitude modulation and pulse position modulation, and digital modulation, such as pulse code modulation. The general trend is toward pulse code modulation systems because of simplicity of radio equipment and efficiency of transmission in a power limited environment, such as may be encountered in satellite communication systems.

In time division multiplex multiple access systems, it has in the past been the practice for a common repeater to receive a number of independent carrier signals and by commutation equipment carried in the repeater interleave the independent carrier signals bit by bit in a continuous sequence. This arrangement requires considerable equipment in the repeater. If the repeater is mobile, such as in satellite communication systems and the like, there could result a weight problem for the vehicle carrying the repeater equipment and with respect to a satellite carrying the repeater equipment an increase in the cost of the launch vehicle to place the satellite in a desired orbit.

In a prior art time division multiplex multiple access system, such as described in US. Pat. No. 3,320,61 l and Belgium Pat. No. 669,3l8, there is described an ar'angement enabling a reduction in the hardware required in the repeater and, hence, a reduction in the problem of providing a vehicle to carry this repeater. By removing the time division multiplex equipment from the repeater itself it is possible to use a simple clipper/amplifier or hard limiting repeater.

It has been found, in addition, that the pulse or bit-by-bit interleaving imposes considerable equipment problems in the plurality of stations requiring access to the common repeater. This complexity can be overcome, or at least materially reduced where the interleaving at the repeater is performed on bursts of pulses from each station.

Where there is relative movement between the common repeater and the plurality of stations, whether it is the repeater that is moving, or the stations that are moving, or both the repeater and stations moving relative to each other, it is necessary, where time division multiplex techniques for multiple access to the common repeater are employed, to provide in some manner the range information between the station considered and the common repeater on a continuous basis. In the above-cited prior art patents, this range information was obtained from a computer or li.;: ic. contained in each of the plurality of stations which provide information of the relative location of and range between the common station and the considered one of the plurality of stations with the programming of the computer being based upon predicted relative movement between the common repeater and the considered station. The total inaccuracy of the range prediction with elementary equipment has been determined to be in the order of one microsecond. Hence, the system timing format was developed having a one microsecond guard band between transmission from each station and the next adjacent station in the format. To realize reasonable efficiency of utilization of the common repeater, each station burst interval must be long in comparison to this guard band, hence, a burst length of microseconds was established. Thus, each station must have equinment to store communication traffic for a short period of time and transmit this in a 125 microsecond burst. The repeti- Jon interval and, consequently, the required storage time is the product of burst length and the number of simultaneous users for which the multiple access system has been designed.

In a known prior art arrangement, the continuous range information is provided by means of a pseudo noise code signal transmitted from each of the stations through the repeater back to itself with the equipment responding to this pseudo nol code signal to adjust the timing signals to account for changing range between the station and the repeater with the control of the timing signals being performed in a digital manner.

SUMMARY OF THE INVENTION An object of the present invention is to provide a time division multiplex multiple access system of the types described above employing time division multiplex techniques of an improved nature relative to the previously employed time division multiplex multiple access system.

Another object of the present invention is to provide a synchronizing system for a time division multiplex multiple access system wherein the synchronization equipment is located in each of a plurality of slave stations and a master station with the common repeater being of the hard limiter repeater.

Still another object of this invention is to provide a synchronizing system for a time division multiplex multiple access system that does not require knowing or predicting the position of the considered station and the common repeater and the range between the considered station and the common repeater.

A further object of this invention is to provide a synchronizing system for a time division multiplex multiple access system wherein the range information is continuously obtained by employing a pseudo noise code ranging signal.

Still a further object of this invention is to provide a combined digital and analog synchronizing system for a time division multiplex multiple access system incorporating therein an arrangement to compensate for the doppler shift of the transmission path from the considered station to the common repeater.

A feature of this invention is the provision ofa synchronization system to control signals transmitted from each of a master station and a plurality of slave stations to be propagated through a common repeater in a different one ofa plurality of time slots of time division multiplex format at the repeater, the stations and the repeater having relative motion therebetween, comprising first means disposed in the master station to transmit a sync burst through the repeater in one of the time slots; second means disposed in each of the slave stations responsive to the sync burst from the repeater to control the production of timing signals employed to control the time of transmission of the transmitted signals from the associated one of the slave stations; third means disposed in each of the slave stations to transmit a ranging signal through the repeater in its associated one of the time slots; and combined digital and analog means disposed in each of the slave stations coupled to the second means responsive to the ranging signal received from the repeater to adjust the phase of the timing signals so that the time of transmission of the transmitted signals from the associated one of the slave stations is such that the transmitted signals occur in the proper one of the time slots at the repeater.

Another feature of this invention is the provision of means in each of the above-mentioned first means and second means to adjust the frequency of the synch burst and the frequency of the transmitted signal to compensate for the doppler effect in the transmission path from the master station to the repeater and in the transmission path from the associated one of the slave stations to the repeater to provide the sync burst and the transmitted signal received at the repeater with the desired frequency.

BRIEF DESCRIPTION OF THE DRAWING The above mentioned and other features and objects of this invention will become more apparent by reference to the following description taken in conjunction with the accompanying drawings, in which:

FIG. 1 is a block diagram illustrating the multiple access system in accordance with the principles of this invention;

FIG. 2 is a timing diagram illustrating the frame format of the time division multiplex frame at the common repeater of FIG. 1;

FIG. 3 illustrates the manner in which the sheets containing FIGS. 4, 5, 6, 7, 8, and 9 should be arranged;

FIGS. 4, 5, 6, 7, 8 and 9, when arranged as illustrated in FIG. 3, is a block diagram of the equipment contained in each of the stations of FIG. 1 in accordance with the principles of this invention;

FIG. 10 is a timing diagram illustrating the signals identified in the receive clock phase locked loop of FIG. 5;

FIG. 11 is a timing diagram illustrating the signals present in the pseudo noise ranging phase locked loop of FIG. 8;

FIG. 12 illustrates the characteristic of the lock detector and error detector of the pseudo noise ranging phase lock loop of FIG. 8;

FIG. 13 illustrates the manner in which the sheets of drawings containing FIGS. 14 and 15 should be arranged;

FIGS. 14 and 15, when arranged as illustrated in FIG. 13, is a block diagram of the receive timer and control equipment therefore in accordance with the principles of this invention;

FIG. 16 is a timing diagram illustrating the timing signal present in both the receive and transmit timers of the station equipment illustrated in FIGS. 4, 5, 6, 7, 8 and 9;

FIG. 17 illustrates the manner in which the sheets containing FIGS. 18, 119 and 20 should be arranged; and

FIGS. 18, I9 and 20, when arranged as illustrated in FIG. 17, is a block diagram of the transmit timer and the control circuitry therefore in accordance with the principles ofthis invention;

DESCRIPTION OF THE PREFERRED EMBODIMENT Referring to FIG. 1, there is illustrated therein a block diagram of a generalized multiple access communication system wherein a plurality of stations, which for purposes of illustration and explanation are assumed to be master station 111 and slave stations 11 to 10, are shown in two-way communication with common repeater 12. As indicated, common repeater 12 can be fixed or mobile and each of stations 1 through 1111 can be fixed or mobile. It should be noted that the multiple access system of FIG. 1 can be used in any of the applications outlined hereinabove under the heading Background of the Invention." While this multiple access system can employ frequency division multiplex or random access techniques for multiple access, in accordance with the principles of this invention it is intended that multiple access be provided to repeater 12 by employing time division multiplex techniques.

In the multiple access communication system each of the stations ll through 11 transmits bursts which are timed to ar rive at the repeater 112 on a noninterferring relation relative to bursts from other stations. The phasing of the transmitted signals must then be adjusted to compensate for the difference in range between repeater 12 and the various stations 11 to 111. Where there is motion between repeater 112 and stations 11 and II, the range is continuously changing and the continuous phase adjustment results in a frequency offset, namely, the doppler effect which is proportional to the rate of change of range.

The time division multiplex format for the system of this invention is shown in curve C, FIG. 2, and is based on the real time transmission of information. A single sync burst, as illustrated in Curve A, FIG. 2, is transmitted from the master station and 10 data bursts of 10 microseconds duration are transmitted from slave stations ll through 110. Curve B, FIG. 2, illustrates the data bursts transmitted from slave station 11. As il1ustrated in Curve C, FIG. 2, the frame format at repeater 12 includes the sync burst and data bursts each having a 10 microsecond duration with a 1.25 microsecond guard time therebetween resulting in a frame period of 123.75 microseconds.

Synchronization is maintained by doppler tracking in which master station 1 1 tracks the doppler actively and slave stations 11 to 10 track the doppler passively. The sync burst carrier is modulated by 800 kHz. (kilohertz) which is harmonically related to the frame rate and is used for all time divisions in the format. Master station 11 transmits the sync burst consisting of 800 kHz. minus its own doppler and receives the sync burst as 800 kHz. plus its own doppler. The transmitted frequency is adjusted so that sum of the transmitted and received frequencies is a constant. This method of doppler tracking assures that the sync signal at the repeater 112 is true 800 kHz.

A slave station receives the pulsed sync burst from master station 111 via repeater 112. Since the signal was a true 800 kHz. at repeater 1.2, it will be received with the true doppler of the considered slave station. The slave station uses this signal to preset its transmit frame rate by a technique identical to that used by master station 1111 for doppler tracking. The slave stations used in pseudo noise code ranging technique to determine the correct initial phase for their transmit format.

The equipment employed in each of the master station 1111 and slave stations I through is illustrated in block diagram form in FIGS. 4, 5, 6, 7, 8 and I), when arranged as illustrated in FIG. 3. By proper manipulation of switches A tlSEiSl, A4853, A4856, AdSltS, and AiSdI-I (FIG. 9), it is possible to have the equipment operate as a master station wherein the pseudo noise and subcarrier operation is inhibited and the sync burst and carrier operation are activated. To provide operation as a slave station, the sync burst modulation and carrier are inhibited and the pseudo noise modulation and subcarrier are activated.

in a master terminal, the master sync burst is generated in generator 13 and is used to phase modulate the carrier signal from oscillator 14 in phase modulator 15. The output of the phase modulator 15 is gated in gate 16 to form the sync burst. The gate control signal is generated in transmit timer 17 by dividing down the transmit clock provided by voltage-controlled oscillator 18 (FIG. 6) of the transmit clock phase locked loop 19.

The received [F (intermediate frequency) signal is demodulated by the carrier tracking phase locked loop 20 (FIG. 4). The resulting 800 kHz. sync burst is used to lock the receive clock phase locked loop 21 (FIG. 5) thereby generating a continuous 800 kHz. clock which drives receive timer 22. Timer 22, in turn, generates the gating ,.ulse for the carrier tracking phase lock loop 20 and the reference signals for the receive clock phase locked loop 21. initially, neither of these loops is locked. To aid acquisition, the output of the carrier-tracking phase lock loop 20 drives the sync burst detector 23. This circuit which contains an 800 kHz. filter, envelope detector and threshold detector. This circuit will detect the time of occurrence of the sync burst even before carrier tracking phase locked loop 20 acquires. The occurrence of the sync burst resets the phase of receive timer 22 to within a fraction of a burst. This is sufficient for the carrier tracking phase locked loop 20 and receive clock phase locked 100p 21 to lock.

Receive clock phase lock loop 21 may lock in such a manner that receive timer 22 is an integral number of clock cycles out of correct phase. Receive clock phase locked loop 21 includes an error detector to measure relative timing between the received burst and the locally generated reference signal. If an error exists, receive timer 22 is stepped in the proper direction until it is operated in correct phase.

The transmit clock generator, identified as transmit clock phase locked loop 19, contains the doppler compensation circuit which adjusts the transmitted clock frequency to maintain the repeater clock frequencies at exactly 800 kHz. independent of first order doppler effects. This is implemented by passing the receive and transmit clock frequencies through a mixer arrangement which extracts their sum frequency, then compares The sum frequency with a 1600 kHz. reference frequency and adjusts the phase of the voltage controlled oscillator 18 in such a manner to keep the sum exactly equal to 1600 kHz.

Letting fR and/T denote the receive and transmit frequencies, respectively, this may be written as fR-l-fT=l600. Assuming that fT is decreased by the doppler coefficient d in transmission, the repeater frequency will be frep. =fT-d. Ignoring second order effects, the receive frequency will be equal to the repeater frequency reduced by d, namely,fR frep. d. Substituting the last two equations into the first equation will demonstrate that the repeater frequency is 800 kHz.

Now let us consider briefly the operation of the equipment of FIG. 4, 5, 6, 7, 8 and 9 when operating as a slave terminal. The slave terminal has the same circuits used in the master terminal for receiving the master sync burst and generating the received clock and doppler compensated transmit clock. In the case of the slave terminal, however, this doppler compensation is an open loop operation. it cannot correct for differences between the 1600 kHz. reference frequency at the master and slave stations, or for second order frequency errors in the repeater clock frequency. In addition, the initial phase of the transmit timer must be adjusted according to the range of the repeater from the slave stations.

The open loop doppler compensation provided by transmit clock phase locked loop 19 (FIG. 6) is supplemented during acquisition by pseudo noise ranging phase locked loop 24 (FIG. 8). The slave terminal transmits a subcarrier signal derived from oscillator 25 (FIG. 9) which is phase modulated by the pseudo noise code in modulator 26. The subcarrier frequency is 2 MHz (MegaHertz) from the master carrier as produced by oscillator 14 and is at low power level level so as to cause no interference with the sync burst or any data bursts.

Subcarrier tracking phase lock loop 27 is used to demodulate the received pseudo noise code ranging signal. The demodulated pseudo noise signal is compared with the demodulated sync burst as follows. Ret eive timer 22, which is locked to the demodulated sync burst, generates a reference sequence in generator 28 (FIG. 8). This reference sequence is compared with the demodulated pseuno noise code in the pseudo noise ranging phase lock loop 24 including therein cross correlators. The phase of the transmit timer 17 is stepped until a lock to within a fraction of a burst is obtained. Once this approximate lock has been obtained, the timing error between the received and reference pseudo noise code sequences is measured. The timing error signal is used to correct the phase of the timing signals of transmit timer l7 digitally and then by a motor driven phase shifter 29 (FIG. 6). Once the pseudo noise ranging phase locked loop 24 has slewed in, it tracks the residual frequency error of the doppler compensation system described above.

The foregoing has been a brief summary of the operation of the equipment shown in FIGS. 4, 5, 6, 7, 8 and 9 for both a master station and a slave station. The description will now proceed with a more detailed description of the various functional blocks of this equipment. As pointed out herein above the equipment of FIGS. 4, 5, 6, 7, 8 and 9 include all the equipment necessary to mal" the station operation as a master or slave station. In the master station the transmission of the master sync is enabled and the pseudo noise ranging signal is disabled. In the slave station the transmission of master sync is disabled and the pseudo noise ranging system is enabled.

The operation of the various components in a terminal are related to one another. The received IF enters the station at the 70 MHz. lF distributor 30 which consists ofa set of buffer amplifiers arranged to provide demodulator 1F, carrier 1F and subcarrier lF. The demodulator [F provides an input to unit 31 wherein the communication data is demultiplexed, demodulated and otherwise processed for utilization. The carrier [F is the input to carrier-tracking phase locked loop 20 where the master sync signal is extracted. The subcarrier [F is the input to ubcarrier-tracking phase locked loop 27 where the pseudo no code is extracted.

The carrier-tracking phase locked loop 20 is used as a phase demodulator as will be described in further details hereinbelow.

The demodulated signal is used to look a 6.4 MHz. voltage controlled oscillator 32 in the receive clock tracking phase locked loop 21. Since the 800 kHz. in the received sync burst contains the repeater to ground doppler, a clock is obtained which is proportional to the receive frame interval. The phaselocxed loop 21 also serves as a narrow band filter for the incoming 800 kHz. sync signal. Included in receive clock tracking phase locked loop 21 is a set of correlators which measure the phase integrity of local receive frame referenced with the incoming sync and local frame clock. Phase-locked loop 21 will be described in further detail hereinbelow.

The local receive frame interval and other necessary timing signals are generated by receive timer 22. Receive timer 22 consists of a set of binary counters clocked from the 6.4 MHz. voltage control oscillator 32 in the receive tracking local phase-locked loop 21 which are arranged to generate the timing signals necessary in the operation of the equipment. Since the clock driving these counters is receive doppler compensated, the receive frame is correct in frequency. The receive frame is phase corrected by receive slew control and receive mode control based on correlator outputs from phase-locked loop 21 which will be described in greater detail hereinbelow.

All time references for the receive frame interval are provided by receive timer 22. These include a signal to gate carrier tracking phase-locked loop 20, reference signals to receive clock-tracking phase-locked loop 21, sync reference signals to drive the sync early-late correlator, pseudo noise reference signals to drive the pseudo noise ranging phase locked loop correlators, burst reference signals to the received channel gates which are used by unit 31 to demultiplex data channels, and 800 kHz. receive signal to transmit clock phase-locked loop 19.

In phase locked loop 19, the received 800 kHz. is added to locally generated 800 kHz. signals the resultant is then compared to a 1600 kHz. frequency reference. The resulting error in turn drives the 6.4 MHZ voltage controlled crystal oscillator 18 from which the local 800 kHz. was derived by dividing the 6.4 MHz. by 8 in binary counter 33. This arrangement forms a phase-locked loop which in conjunction with phase-locked loop 21 forms the equipment to obtain doppler compensation.

To compensate for secondary effect of doppler and to correct for the other factors, such as error in standards, mechanical phase shifter 29 driven by a servomotor 3d adjusts the phase of the output signal from oscillator 18 for use as the clock for the transmit timer R7.

The transmit timer I7 is structurally identical to receive timer 22. It consists of a set of three binary counters arranged to count out the transmit frame interval. All time references of the transmit frame interval are derived from transmit timer 17. When the receive and transmit phase locked loops are locked and the proper drive is provided to servomotor 34, the transmit frame is correct in frequency.

The transmit frame is phase corrected through logic in transmit slew control and transmit mode control based on the correlator outputs from pseudo noise-ranging phase-locked loop 24 as will be described in greater detail hereinbelow. The servo and phase correction logic for transmit timer 17 are only used in a slave terminal. The transmit timer in a master terminal is considered the reference. Therefore, under normal conditions, the servomotor and the slew control are made inoperative in a master terminal.

All timing signals for the transmit frame interval are derived from transmit timer 17. These signals include a gate signal to generate the master sync burst, signal to modulate the master sync burst, burst signals to derive the fill burst selector switches, burst signals to derive the pseudo code generator, burst clock signals to unit 35 (FIG. 9) to control the modulation and multiplexing and other processing of data from the sources of data for transmission purposes, and burst reference signals to drive the transmit channel gates which are used by unit 35 to time the multiplexing of data channels.

The master sync signal from generator 13 is used to phase modulate the output of the 68 MHz. oscillator 14 in modulator 15 whose output is gated by gate 16 into summer amplifier 36. The unmodulated 68 MHz. carrier is also gated into summer 36 to select the fill bursts. As mentioned before the master sync transmission is only operative when the station is a master.

For a slave station, the subcarrier is enabled. The output of the pseudo noise generator 37 modulates the output of 70 MHz. subcarrier oscillator in modulator 26 whose output is gated into summer 36 by transmission gate 38. The summer 36 mixes the carrier, subcarrier and output of modulator unit 35 and presents these signals to the station transmitter for transmission.

In the pseudo noise-ranging circuit the subcarrier IF is coupled through a 70 MHz. band pass filter and limiter 39 and buffer amplifier 40. The output of amplifier d0 drives the subcarrier tracking phase locked loop 27 which acts as a phase demodulator. The received pseudo noise code at the baseband output of phase locked loop 27 is correlated with the reference pseudo noise code supplied by generator 28 activated by receive timer 22 at the pseudo noise ranging correlators. The output of the correlators provide the basis for adjusting the transmit timer phase.

The station equipment of FIGS. 4, 5, 6, 7, 8, 9 will now be described more specifically. Referring to FIG. 43, the received IF signal coupled to distributor 30 which includes therein buffer amplifiers 4] and 42 to couple the IF signal to locked loop 20, buffer amplifier 43 to couple the IF signal to unit 31 which is not part of the synchronization system, and buffer amplifier M to couple the IFsignaltothe phase-locked loop 27 (FIG. 7). Phase-locked loop 20 locks to the sync burst received from the master station via the repeater by employing phase detector 55 coupled to the output of amplifier M and to the output of voltage controlled oscillator &6 through buffer amplifier 67. The error signal at the output of detector 35 is gated by sampling gate 48 during the 10 microsecond period that the sync burst is received, the gate signal being received from receive timer 22. This gate signal is normally a +3 volt signal which holds gate 38 open. During the sync burst, this gate signal goes to 0 and closes gate 48.

The reasons for gating the error signal in phase-locked loop 20 are to reduce the effect of noise, which is present during the entire frame, and to prevent signals other than the sync burst from pulling the loop frequency.

The output from gate 48 passes through rise time network 49 before applying it to loop filter 50 to prevent the step voltage at the output of gate 68 from overloading the operational amplifier in the loop filter circuit. Network l9 also has the advantage of providing a low source impedance for the loop filter during the time gate dis is opened. This is significant because otherwise stray capacitance on this line would effect the loop filter gain by acting as a holding capacitor.

The incoming carrier frequency can be $400 kHz. from its normal value due to doppler and local oscillator frequency tolerances. Locked loop 20 tracks over this range with a maximum phase error of 9.

A lock detector is provided by supplying phase detector 51 with the output signal of oscillator 66 shifted by in phase shifter 52. This phase shift can be provided by the proper length of coaxial cable. The output of phase detector 51 is also gated by sampling gate 53 to reduce the effect of noise and to drive lock detector 54. The output of detector 54 is normally +3 volts and goes to 0 when lock is detected.

The demodulated 800 kHz. sync signal appears at the output of detector 45. However, a separate phase detector 55 is coupled to the output of oscillator 46 and amplifier 62 to extract the baseband to avoid loading effects on the sampling gate 48 following phase detector 35. Phase detector 55 has its output coupled to baseband amplifier 56 which provides voltage gain, buffering and also inversion if required. Phaselocked loop 20 will lock in a stable manner at integral multiples of the sampling rate of approximately 800 Hertz from the incoming carrier frequency. This is due to the pulse nature of the incoming sync burst and occurs whether the loop is gated or not.

Referring to FIG. 5, receive clock phase-locked loop 211 is in effect a gated loop due to the reference signals applied to multipliers 57, 58 and 59 from receive timer 22. Loop 2i operates on the received master sync burst supplied from amplifier 56 (FIG. 4). Loop 21 tracks the doppler on the 800 kHz. tone in the demodulated sync burst and provides a continuous 800 kHz. clock to operate transmit timer 17.

The input to loop M is the demodulated signal from amplifier 56 of loop 20 identified as 8(2) and illustrated in Curve A, FIG. 10. This signal drives multiplier 47 which is the phase error detector for the loop having applied thereto reference signal X (t) as shown in Curve B, FIG. It). The baseband signal also is applied to multiplier 58 together with a reference signal X t) as illustrated in Curve C, FIG. It) which is used for the lock detector. Since loop 21 can lock an integral number of clock cycles from the correct time, a means for detecting the leading and trailing edges of the sync burst is provided by multiplier 59 coupled to the baseband signal input and to the reference signal X (1) supplied from receive timer 22 as illustrated in Curve D, FIG. 10.

When the input and reference signals are aligned in time, as illustrated in FIG. It), the output from multiplier 57 will be a burst of I600 kI-Iz. square wave and will have an average value of zero. Loop filter 60 passes only this DC term which then corrects the phase of the receive clock provided by oscillator 32. For a timing error less than one-quarter a clock period, the DC value is proportional to the timing error.

The output of multiplier 58 has a maximum DC value when its two inputs are correctly aligned in time and reaches zero when the timing error equals one-quarter clock period. The DC component of the output of multiplier 58 is applied to threshold detector 61 through low pass filter 62. Detector 61 has the threshold adjusted for one-half the maximum and corresponds to a timing error of one-eighth clock period or 157 nanoseconds.

Multiplier 58 has false peaks for timing errors of 11.25 microseconds, $2.5 microseconds, etc. In addition, these timing errors correspond to stable nulls in the loop. The output of multiplier 59, the early-late detector, has a zero DC value for the correct lock position. If the master sync burst is early, that is, S (t) leads X (t) (Curves A and C, Fig. 10) then multiplier 59 has an output which is a maximum positive DC value when the timing error is +1.25 microseconds, etc. The positive threshold as established in thres Id device 63 which receives its input from low pass filter 64 is adjusted for one-half this maximum DC value. Thus, for a false lock, in which the sync burst is early by an integral number of clock periods, threshold device 63 generates an advanced signal for receive timer 22.

Upon receipt of this signal and in the automatic mode, receive timer 22 advances one clock period, that is, 1.25 microseconds. If the sync burst is still early after waiting a period of approximately 0.1 second, timer 22 advances again. This process repeats until the loop reaches the correct lock.

A similar procedure is followed when the sync burst is late by an integral number of clock periods employing negative threshold device 65.

Details of receive timer 22 and the manner of controlling the same from the output signals of locked loop 21 will be described hereinafter with reference to FIGS. 14 and 15.

Referring to FIG. 6, the 800 kHz. +d output of receive timer 22 is coupled to transmit clock phase locked loop 19. Locked loop 19 compensates the transmit clock for first order doppler effects and the servo driven phase shifter 29 compensates for secondary clock errors. The doppler compensation circuit consists of a source of reference frequency 66, a mixer arrangement including mixers and filters 67, 68 and 69 to form a 1600 kHz. resultant signal at the output of mixer and filter 69 for application to phase detector 70 which receives its other input from source 66. It is necessary to avoid mixing the transmit and receive 800 kHz. clocks directly, otherwise, the sum frequency could not be distinguished from the second harmonies of the two input frequencies. The receive 800 kHz. is mixed with a 350 kHz. offset frequency signal from source 72 in mixer and filter 67 and the 450 kHz. difference signal is extracted. The transmit 800 kHz. signal from counter 33 is mixed with the offset frequency signal from source 72 in mixer and filter 68 and the 1150 kHz. sum frequency is extracted. The output signals from mixers and filters 67 and 68 are mixed in mixer and filter 69 to form the desired 1600 kHz. sum frequency. The output from detector 70 drives loop filter 71, which in turn, controls the voltage controlled oscillator 18. The output of oscillator 18 is divided by counter 33 to provide the transmit 800 kHz. d. The receive 800 kHz. +d is from the repeater to this station only. Therefore, when the loop is locked the transmit clock will be doppler compensated.

The above description completes the description of the transmit phase locked loop 19 as it is used in master terminal. The motor-driven phase shifter 29 is inactive in the master station. However, this phase shifter 29 is used in a slave terminal as follows.

Secondary transmit clock error corrections are implemented by taking the output from oscillator 18 which is doppler compensated and phase shifting it in the servo driven phase shifter 29. The output of phase shifter 29 then provides a clock which is corrected for all other timing errors, assuming that the proper drive is applied to servo motor 34. This function is made part or the loop to generate the proper servo drive. One such loop is the pseudo noise ranging phase locked loop 24 (FIG. 8).

Phase shifter 29 is a mechanical device which introduces a phase shift by varying the dielectric between capacitor plates.

The output of phase shifter 29 is coupled through buffer amplifier 73 to transmit timer 17 which will be described in greater detail in connection with FIGS. l8 l9 and 20 herein below.

Referring to FIG. 7, there is illustrated therein the components forming subcarrier tracking phase locked loop 27. This loop is used as a phase demodulator for the pseudo noise ranging signal. The main difference bet een this loop and the carrier tracking phase locked loop is that loop 27 is not gated. The input to loop 27 is filtered at IF in band-pass filter and limiter 39 to remove the master sync signal and other data channel signals which are received on the subcarrier frequency from the repeater. The band-pass filter is followed by a limiter to remove amplitude variations caused by signal suppression effects on the low level pseudo noise-ranging signal. The output from filter and limiter 39 is coupled by means of buffer amplifier 40 to locked loop 27 which includes phase detector 74 receiving one input from amplifier 40 and another input from voltage controlled oscillator 75. The output from detector 74 which is the baseband signal is applied to baseband amplifier 76 and also to loop filter 77 to control oscillator 75. To provide the lock detector, the output from amplifier 40 is coupled to phase detector 78 which has its other input coupled to oscillator through the 90 phase shifter 79. The output from pltase detector 78 is coupled to subcarrier tracking lock detector 80 to provide indication of when loop 27 is locked.

Referring to FIGS. 6 anu v, pseudo noise ranging phase locked loop 24 is illustrated which is used to adjust the phase of the transmit timer (FIG. 9) in a slave station during initial acquisition. The correct phase is determined by comparing the time at which the slave stations pseudo noise-ranging signal is received from the repeater against time the master sync burst is received from the repeater. To facilitate this comparison, a reference pseudo noise signal is generated by generator 28 in response to receive timer 22 (FIG. 5) which itself is locked in time to the received master sync burst.

The carrier-tracking phase-locked loop 20 (FIG. 4) and receive timer 22 (FIG. 5) operate from the sync burst received via the repeater from the master terminal. The pseudo noise reference signal at the output of generator 28 can then be assumed to be fixed in time as far as the slave terminal is concemed.

1 ne purpose of locked loop 24 is to correct the initial phase Jf the slave stations transmit timer 17. This is actually accomplished in three modes. The first two modes are a digital advance or retard of transmit timer 17. The third mode is a phase correction by motor driven phase shifter 29 which operates on the 6.4 MHz. output of oscillator 18 (FIG. 6). The three modes in the acquisition procedure are: (1) coarse slew is a search in which the phase of transmit timer 17 is advanced in ste s on the order of 5 microseconds until pseudo noise lock occurs; (2) fine slew wherein transmit timer 17 is digitally advanced or retarded in steps on the order of 0.5 microseconds depending on whether the received pseudo noise signal is early or late; and (3) maintenance where a fine phase adjustment is obtained by means of phase shifter 29 according to the output of the pseudo noise error detector including multiplier 81, loop filter 82 and motor amplifier 83.

The acquisition procedure depends on the outputs of the three multipliers 81, 84 and 85 shown in FIG. 8. The demodulated pseudo noise signal coupled from baseband amplifiers 76 of phase locked loop 27 (FIG. 7) is denoted S (t) and is illustrated in Curve A, FIG. 11. The reference signals for multiplier 81, 84 and 85 are provided by generator 28 and are respectively denoted X (1) as illustrated in Curve B, FIG. ll; X (1), as illustrated in Curve C, FIG. 11; and X (t), as illustrated in Curve D, FIG. 11. The operation of multipliers 81, 84 and 85 are illustrated in the timing diagram of FIG. 11. The signal S (t) is shown correctly aligned in time with the various reference signals. Under this condition of correct timing, the receive pseudo noise signal S (t) and the locked detector reference signal X (t), have the same waveform and are in phase.

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Classifications
U.S. Classification375/211, 455/11.1, 375/285, 375/368, 342/88
International ClassificationH04B7/212
Cooperative ClassificationH04B7/2126
European ClassificationH04B7/212B1