|Publication number||US3644657 A|
|Publication date||Feb 22, 1972|
|Filing date||Oct 20, 1969|
|Priority date||Oct 20, 1969|
|Publication number||US 3644657 A, US 3644657A, US-A-3644657, US3644657 A, US3644657A|
|Inventors||Miller Francis A|
|Original Assignee||Miller Francis A|
|Export Citation||BiBTeX, EndNote, RefMan|
|Referenced by (17), Classifications (6)|
|External Links: USPTO, USPTO Assignment, Espacenet|
United States Patent [151 3,644,65 7 Miller Feb. 22, I972  ELECTRONIC AUDIOFREQUENCY lfrirr lqfy Examiner william H, Beh lr. V
MODULATION SYSTEM AND METHOD Assistant Examiner-R. Skudy Attorney-Richards, Harris & Hubbard  Inventor: Francis A. Miller, 6600 NW. 39th Expressway, Bethany, Okla. 73008  ABSTRACT  Filed: 1969 The invention is directed to modulation of the amplitude and  AppL 868,294 frequency of an audiofrequency signal by passing the signal through at least two parallel channels, with phase-shifting cir- Related US. Application Data cuitry in the channels providing signals of opposed phases. A low-frequency band is filtered from the signal in one of the  ggg g z g of 771900 channels and a high-frequency band is filtered from the signal I a an one in another of the channels and the signals shifted back into phase. The filtered signals are then combined into an output [2%] ..84/l.25,(l4/)Il:ll(;6: signaL The high and lowfiequency bands filtered from the t: d h 84/] 25 1 01 1 18 l 24 signals are selectively varied to induce vibrato, tremolo, and l e 0 84/1 1 k2 i other tonal effects into the output signal. in a preferred form of the invention, the highand low-pass filters are matched so as to form substantially a 180 phase shift stage with a flat  Referm cued frequency response. Four stages are coupled to provide 720 UNITED STATES PATENTS of phase shift. The original audiofrequency signal is then summed with the phase-shifted signal in approximately equal ,814 5/1958 Dol'f proportions to produce a coordinated effect substantially 2,845,598 1953 y duplicating that produced by a rotating speaker. The phase 3,255,297 6/1966 Long shift is started and stopped in a manner to duplicate the 3,413,418 12/1968 "34/125 dynamic effects of starting and stopping a rotating speaker. A Jacob; dual channel ystem is described for duplicating a multiple speaker system.
37 Claims, 20 Drawing Figures sos PHASE S H FT 508 MUSIC I I 6 AUDIO SOURCE NETWORK i GENERATOR AA, 5 I 2 PAIENIEBFEB 22 1912 sum 3 0F 5 .SEbo
OWN d PAIENTEUFB22 I972 SHEET [1F 5 AUDIO GENERATOR HASE SHIFT NETWORK MUSIC SOURCE FIG. IO
ELECTRONIC AUDIOFREQUENCY MODULATION SYSTEM AND METHOD This is a continuation-in-part of application Ser. No. 771,900 filed on Oct. 30, 1968, now abandoned.
This invention relates to method and apparatus for modulating an electrical signal, and more particularly to a method and apparatus for inducing vibrato and tremolo into audio frequency musical signals.
THE PRIOR ART It has long been known that the introduction of vibrato and tremolo into musical sound provides pleasing musical effects. Vibrato is generally defined as a relatively low-rate modulation of the frequency of a sound about a mean frequency, whereas tremolo refers to a relatively low-rate modulation of the amplitude or intensity of sound For simplicity of description, the word tremulant is often used in a general manner to refer to variations of tonal quality of audiofrequency signals, including both vibrato and tremolo. In practice, the tremulant modulation rate utilized in musical performances is within the range of about 1 to 15 cycles per second.
A number of electronic circuits have been heretofore developed for the introduction of tremulant effects into the electrical output of a musical instrument, such as an organ. For instance, amplifiers have been previously utilized in which the output of an oscillator is amplitude modulated by such circuitry as to control grids of a push-pull output stage. Other oscillator circuits using filtering of various'frequency bands have also been proposed, such as thesystem described in U.S. Pat. No. 3,205,294, issued Sept. 7, 1965. Systems have also been developed wherein a tremulant effect is induced by filtering a signal while varying the frequency response of the filter. Examples of such circuits are described in U.S. Pat. No. 3,388,257, issued June 11, 1968. However, such electronic modulation circuits heretofore developed have not enjoyed widespread use in the general music field, as the number and quality of tonal effects which may be created with the circuits are limited, and additionally because many of the circuits have not been adaptable for use with a variety of musical instruments.
Primarily because of the lack of an adequate electronic substitute, extensive use is presently made of mechanical tremulant devices wherein audio frequency signals are modulated by rotating loud speakers, sound channels, or sound reflectors.
An example of one such well-known mechanical tremulant device commonly termed the Leslie speaker is disclosed in U.S. Pat. Re. 23,323 issued to D. J. Leslie. Not only are rotating mechanical modulation devices useful in introducing tremolo and vibrato into musical programs, but a number of distinctive musical effects may be produced by such devices due to the starting and stopping dynamics of the mechanically rotated parts in the devices. Often during performance of certain musical compositions, such rotating mechanical devices are purposefully turned off and on a number of times to produce unusual musical effects. However, rotating mechanical devices are not only extremely bulky and expensive, but are subject to mechanical failures of the rotated parts therein.
In one embodiment of the invention, the tremulant effect produced by a rotating speaker is closely simulated by a plurality of high and low pass filter stages connected in cascade.
The amplitude of the pass bonds of the filters are slightly I mismatched so as to provide amplitude modulation in synchronism with the frequency modulation produced by the phase shift resulting from the stages. In accordance with another aspect of the invention, the vibrato rate is set at two different levels and a 'means provided for changing from one rate to the other in a'manner to duplicate the musical effects introduced by the inertial characteristics of a rotating speaker.
In accordance with another embodiment of the invention, phase shift is introduced to the musical signal without significantly changing the amplitude, then the original signal is summed with the phase-shifted signal to provide amplitude modulation by cancellation. In a preferred form of this embodiment, the original signal has an amplitude from about 0.5 to about 1.5 that of the phase-shifted signal, and has a total phase shift of at least about 720. Best results are obtained with a phase shift angle that is an even multiple of 360.
In accordance with one aspect of the invention, different frequency bands are filtered from a plurality of parallel phase shifted audiofrequency signals and then the filtered signals are combined. The frequency bands filtered from the phaseshifted signals are selectively varied in order to induce tremulant effects into the signals.
In another aspect of the invention, a plurality of parallel channels generate outputs which are added to provide a modulated audiofrequency output signal. Each of the channels comprises filters for filtering upper and lower frequency bands from a plurality of parallel signals having opposed phases. The frequency responses of the filters are selectively variable so that the crossover frequencies of the frequency responses of the filters are varied to induce modulation of the output signals. The rate of variance of the frequency responses of filters of different channels may be selectively changed to produce interesting musical effects.
For a more complete understanding of the present invention and for further objects and advantage thereof, reference is now made to the following description taken in conjunction with the accompanying drawings, in which:
FIG. 1 is a diagrammatic illustration of one utilization of the tremulant system according to the invention;
FIG. 2 is a diagrammatic illustration of another use of the tremulant circuit of the invention;
FIG. 3 is a schematic diagram of one embodiment of the present tremulant circuit;
FIG. 4 and.5 are diagrammatic waveforms illustrating different modes of operation of the circuit shown in FIG. 3;
FIG. 6 is a schematic diagram of another embodiment of the invention;
FIG. 7 and 8 are schematic diagrams of different embodiments of circuits for driving the variable light sources of the invention;
FIG. 9 is a schematic diagram of another embodiment of the present tremulant system;
FIG. 10 is a schematic block diagram of a system in according with the present invention;
FIG. 11 is a schematic diagram which assists in describing how the circuit of FIG. 10 duplicates the sound of a rotating speaker;
FIGS. l2, l3, and 14 are grafts of the phase shift, frequency modulation, and amplitude modulation produced by a preferred embodiment of the system illustrated in FIG. 10;
FIGS. I5, 16, and 17 are grafts of the phase shift frequency and amplitude of another embodiment of the system illustrated in FIG. 10;
FIG. 18 is a detailed circuit diagram of another embodiment of the tremulant system in accordance with the present invention;
FIG. 19 is a schematic circuit diagram of still another tremulant system in accordance with the present invention which permits trimming of the exact degree of phase shift and precise synchronization of the original signal with the phaseshifted signal; and
FIG. 20 is a schematic circuit diagram of still another tremulant system in accordance with the present invention.
Referring to FIG. 1, the electrical output signals from an organ 10 are fed to the tremulant modulation system 12 which is constructed in accordance with the present invention. If desired, the modulation system 12 may be packaged as a very small volume system and disposed inside the organ 10. The system 12 operates upon theaudiofrequency signals from the organ 10 in a manner to be later described and provides a modulated output signal to drive the speaker system 14. Both vibrato and tremolo effects may be introduced by the system 12 according to a preselected periodic modulation pattern, or alternatively, exterior controls on the system 12 may be operated by the musician in order to produce varied musical effects as desired. The modulation system 12 is capable of driving the speaker system 14 to closely simulate the operation of a rotating speaker of the Leslie type.
FIG. 2 illustrates another configuration of the present invention wherein an electric guitar l6 and another musical instrument such as an organ 18 are connected to the input of a preamplifier 20. It will be understood that the present invention is not limited to use with any particular type of musical instrument, but that the illustrated instruments are utilized for disclosure purposes only. The output of the preamplifier 20 is fed to a tremulant system 22 according to the invention and is also fed to the input of a nontremulant amplifying system 24. The tremulant system 22 may be operated according to a preselected periodic modulation cycle, or may be selectively varied to produce different tremulant effects by the use of suitable controls operated by the musician. The outputs of the tremulant system 22 and the nontremulant system 24 are combined and fed to an amplifying system 26 which drives a speaker system 28.
FIG. 3 is a schematic diagram of an embodiment of the basic tremulant circuit according to the invention. The input audiofrequency signal is represented by the signal source 30, it being understood that this signal may be provided by any one of a number of different musical instruments. The audiofrequency signal is fed through a transformer 32 which has a center-tapped secondary grounded via a lead 34. One terminal of the secondary of the transformer 32 is connected to a variable resistance 36. A capacitor 38 is connected between the resistor 36 and circuit ground to form a low pass filter network. The output from the low pass filter network is fed via a resistance 40 to the output of the circuit.
The other terminal of the secondary of the transformer 32 is connected to a capacitor 42, one terminal of which is connected through a variable resistance 44 to circuit ground. Capacitor 42 and variable resistance 44 comprise a high pass filter, the output of which is fed via a resistance 46 to the output of the circuit. In the preferred embodiment, the variable resistances 36 and 44 are photosensitive and are preferably, but not necessarily, of the cadmium sulfide or cadmium selenide type. A light source 48 is driven by an electrical generator 50 in a preselected manner in order to provide a variable source of illumination to the photosensitive variable resistances 36 and 44. The variable illumination impinged upon the resistances 36 and 44 varies the frequency responses of the high pass and low pass filters to induce tonal variations in the audio frequency signal in the manner to be subsequently described.
FIG. 4 diagrammatically illustrates the operation of the present tremulant circuit. The frequency response of the low pass filter comprising resistor 36 and capacitor 38 is represented by the curve 52. The frequency response of the high pass filter comprising capacitor 42 and resistor 44 is represented by the waveform 54. The waveforms 52 and 54 intersect at a crossover point 56. In this instance, the crossover point is at the SO-percent level, or midway between the high and low frequency responses of each filter bandpass.
By varying both of the variable resistors 36 and 44 with the same light source 48, the frequency response of one of the filters may be concurrently increased while the frequency response of the other filter is decreased, and vice versa. Thus, variation of the resistances in both the filters varies the rollover point 58 of the waveform 52 between points 58a and 58b, while the rollover point 60 of the waveform 54 is concurrently varied between points 60a and 60b. This variance of the rollover points of the output waveforms from the filters thus varies the position of the crossover point between the points 56a and 56b.
As an example of the operation of the tremulant circuit, assume that the value'of the resistances 36 and 44 is such that the crossover point of the waveforms 52 and 54 is at 56a. All frequencies lower than the frequency at point 560 will be 180 out of phase with the frequencies higher than the frequency at point 56a. Assuming that the light impinged upon the resistances 36 and 44 is changed such that the crossover point is moved to 56b, successively higher frequencies, are reversed in phase by 180 during the movement of the crossover point, thus producing a tremulant effect.
Interesting effects can be obtained by mixing the unmodulated signal at the input of the tremulant circuit with the modulated signal at the output of the circuit. The relative amplitudes of the unmodulated and modulated signals can be varied as desired by suitable variable networks, as will hereafter be described in detail.
The crossover point 56 of the waveforms 52 and 54 need not be at the 50-percent level. As shown in FIG. 5, the crossover point 56 may be considerably below the SO-percent level to provide an unusual amplitude modulation of the output audiofrequency signal when the crossover points are varied between 56a and 56b in the manner previously described. it will also be understood that crossover points at substantially greater levels than the SO-percent level could also a be advantageously utilized. The circuit has been described as having resistances which vary linearly to maintain the crossover point at generally a constant amplitude during variation of the frequency of the crossover point. However, it will be understood that in some instances, it may be desirable to vary the magnitude of one or more of the variable resistances in a nonlinear manner, thus providing a fluctuating amplitude to the changing crossover point to induce unusual and unique musical effects.
in order to further illustrate the operation of the tremulant circuit shown in FlG. 3, assume as a first case that the values of the variable resistances 36 and 44 are much greater than the values of the capacitive reactance of the capacitors 38 and 42 for an applied sine wave having a reference frequency. For simplicity, the values of the resistors are assumed to be equal and the values of the capacitors are also assumed equal. The low pass filter comprising the resistance 36 and the capacitor 38 will thus attenuate the portion of the input sine wave such that the amplitude of the waveform applied to the resistor 40 is negligible. Conversely, the high pass filter comprised of the capacitor 42 and the resistance 44 will not greatly affect the portion of the sine wave fed therethrough, and thus the waveform applied to the resistor 46 will be almost identical to the sine wave applied to the high pass filter. The resultant output signal from the tremulant circuit will thus have phase characteristics similar to the waveform applied to the resistor 46 from the high pass filter and will have an amplitude of approximately one-half the input sine wave. The resultant output signal will also be 180 out of phase with the portion of the input sine wave applied to the low pass filter.
Now assuming as a second case that the values of the resistances 34 and 44 are much less than the value of the capacitive reactance of capacitors 38 and 42 at a reference frequency, the waveform applied to resistor 40 will not be greatly attenuated and will thus be almost identical to the portion of the input signal applied to the low pass filter comprising the resistor 36 and the capacitor 38. However, the high pass filter comprising capacitor 42 and resistor 44 will substantially attenuate the portion of the input signal fed therethrough. Thus,
the resultant output of the circuit will have phase characv teristics similar to the signal applied to the resistor 40 from the low pass filter and will have an amplitude of approximately one-half the input signal applied from the source 30. It should be noted that the resultant output in this case is 180 out of phase with the resultant output obtained in the first case.
As the magnitudes of the variable resistances approach the capacitive reactances of the capacitors 38 and 42, the resultant output of the circuit will approach the amplitude of the resultant output provided by either the first or second case, but will be phase shifted with respect to either of the resultant outputs of the two cases. It should then apparent that the phase shift from the first case to the second case will take place smoothly upon a linear variation of the magnitudes of the resistances.
Although the operation of the tremulant circuit shown in FIG. 3 has been described with respect to an input signal having only a single frequency, it will be understood that the circuit operates on complex signals having a number of frequencies in the same manner. However, with such complex signals, each different frequency contained therein will be sequentially operated upon as the relative magnitudes of the resistances are varied.
Use of the present tremulant circuit with complex audio frequency signals causes apparent change of the frequencies of the signals to create a heterodyne change in the harmonic structure of the signals. Operation of the circuit gives the acoustical illusion of multipath effects and acoustical depth which have formerly been possible only with the use of mechanical rotating-type speakers.
It is often desirable to connect a plurality of the tremulant circuits shown in FIG. 3 in series. FIG. 6 illustrates a first tremulant circuit 70 according to the invention which is connected in series with a second like tremulant circuit 72. As the tremulant circuits 70 and 72 are identical, like numbers will be utilized for like and corresponding parts therein. Tremulant circuits 70 and 72 are similar to the basic circuit shown in FIG. 3, with the addition of amplifying circuitry. The input audiofrequency signalis fed to a transformer 74, with one terminal of the center-tapped transformer secondary being connected to a variable resistance 76. The output of the resistance 76 is fed to a grounded capacitor 78 and to a resistor 80. The lower terminal of the transformer secondary is connected to a capacitor 82 connected in series with a grounded variable resistor 84. The common terminals of the capacitor 82 and resistor 84 are connected to a resistor 86.
The common terminal of resistors 80 and 86 are connected through a feedthrough capacitor 88 to the base of a transistor 90. Resistors 92-98 are connected in a conventional amplifying configuration to the transistor 90. Capacitor 100 is connected between the emitter of the transistor 90 and ground. The output of the transistor 90 is fed from the collector thereof through capacitor 102 to the next circuit stage.
Operation of each of the tremulant circuits 70 and 72 is similar to the operation of, the circuit shown in FIG. 3, with amplification being provided to the output thereof. Each of the tremulant circuits 70 and 72 provides a frequency shift of 180 to input signals as the tremulant circuits are varied through one cycle by suitable variable light sources (not shown). The series connection of the two circuits 70 and 72 thus causes a phase shift of 360 of the input signal as the frequency responses of the filter networks therein are varied through a cycle. Similarly, three series connected tremulant circuits will provide a 540 phase shift.
The use of plurality of series connected tremulant circuits increases the slope of the frequency responses of the circuits in the area of the crossover point. For instance, whereas one tremulant circuit may be provided with a frequency response slope from the rollover point of about 6 db per octave, the slope of the frequency response of three series-connected tremulant circuits would be about 18 db per octave. The steeper slope of the frequency response in the area of the crossover point provides a much sharper phase reversal upon variance of the tremulant circuits, thereby emphasizing the tremulant effect provided by the system.
Although the use of photosensitive resistors and a variable light source have been illustrated for variance of the filters of the tremulant circuits, any other suitable variable components may be used in the invention. For instance, a mechanically variable capacitance could be utilized in place of a variable resistance. Alternatively, a filter circuit utilizing a variable inductance such as a saturable reactor could be utilized. Further, the variable resistances could be mechanically varied potentiometers.
FIG. 7 illustrates an embodiment of a circuit which has been found advantageous in driving a variable light source for control of photosensitive elements in the tremulant circuits. A DC signal is applied to the circuit via a resistance 110 to directly determine the frequency of the variable illumination of a lamp. The level of the DC signal applied via the resistor may be varied by manually operable controls (not shown) located on or near the musical instrument. A first positive reference voltage V, is applied to a potentiometer 112, while a second reference positive voltage V, is applied to a potentiometer 114. The cathode of a diode 116 is connected to the adjustable arm of the potentiometer 112, while the anode of the diode 118 is connected to the adjustable arm of the potentiometer 114.
Diodes 116 and 118 are commonly connected to a terminal 119 which is connected to a grounded capacitor 120. Terminal 119 is connected via resistor 122 and a variable resistor 124 to a unijunction transistor 126 which serves as the time base element of the circuit. Bias voltage is fed via resistor 128 to the transistor 126. A capacitor 130 is connected between ground and the input of the transistor 126. The output of the transistor 126 is applied to one terminal of a grounded resistor 132 and also to the emitter of a transistor 134.
Transistor 134 is connected in a multivibrator configuration to a transistor 136. R-C networks'138 and are connected between transistors 134 and 136 in the conventional multivibrator configuration. Bias voltage is applied to the transistor 134 and 136 via resistors 142 and 144, respectively. The output of the multivibrator is fed via a resistor 146 to the base of a transistor 148. A transistor 150 is connected across the output of the transistor 148, the emitter of the transistor 150 being connected to a grounded lamp 152. A resistor 154 is connected between the base of the transistor 150 and circuit ground-A capacitor 156 is connected between the base of the transistor 148 and ground, with a capacitor 158 being connected across capacitor 156 in series with photosensitive variable resistor 160.
The base of a transistor 162 is connected to terminal 119. The emitter of transistor 162 is connected to a grounded resistor 164 and also to the base of a transistor 166. The collector of transistor 166 is connected to agrounded resistor 168 and also to the base of a transistor 170. The collector of transistor 170 is connected to a grounded lamp 172.
In operation of the circuit shown in FIG. 7, the unijunction transistor 126 is tired when the voltage applied at terminal 119 reaches a preselected level. The output signal of the transistor 126 is a spike pulse which triggers the bistable multivibrator comprising transistors 134 and 136. 'The output of the multivibrator is fed via resistor 146 to provide a square wave having a frequency of one-half the frequency of the time base output from transistor 126. That is, for each two spike pulses received from the transistor 126, one square wave is generated by the multivibrator circuit.
The square waves from the multivibrator are fed to an R-C integrator circuit comprising resistor 146 and capacitances 156 and 158. The time constant of the RC integrator circuit is controlled by the magnitude of the voltage to terminal 119. This voltage controls the operation of transistors 162, 166 and 170. The output of the transistor 170 controls the illumination of the lamp 172, which in turn controls the magnitude of the resistance 160.
By proper selection of the component values of the circuit, the amplitude and waveshape of the signal applied to the driver transistors 148 and 150 may be substantially constant over a relatively broad range of operating frequencies. The transistors 148 and 150 drive the lamp 152 with a generally triangular response due to the R-C integrator circuit. The lamp 152 also provides an integration to the resulting illumination due to the response factors of the Wolfram lamp filament therein. Thus, the resulting illumination applied from the lamp 152 is a generally sinusoidal waveform. The period of the generator shown in FIG. 7 may be variable from about 1 cycle per second to about 15 cycles per second to provide various desired effects.
An important feature of the invention is that the rate of change of the voltage applied via the resistor 110 may be selectively varied to dynamically simulate the starting and stopping characteristics of a mechanical rotating speaker. Further, by adjustment of the potentiometers 112 and 114, the maximum and the minimum voltage that may be developed at the terminal 119 is limited to simulate the operation of a twospeed mechanical rotating speaker.
P16. 8 illustrates another embodiment of a circuit for driving a lamp to vary photosensitive elements in the tremulant circuit of the invention. A positive reference voltage is applied to the cathode of a Zener diode 180 and to the anode of a Zener diode 182. A positive bias voltage B is applied to the series connected Zener diodes via a resistance 184. Bias voltage is also applied through a resistor 186 to a unijunction transistor 188. The output of the transistor 188 is fed to theemitters of transistors 190' and 192 which are connected together in a multivibrator configuration. R-C networks 194 and 196 are connected between the transistors 190 and 192 in the conventional bistable multivibrator configuration.
The output of the multivibrator is applied via a variable resistance 198 to a grounded capacitor 200 and to a capacitor 202. A grounded photosensitive variable resistor 204 is connected in series with capacitor 202. The output from the multivibrator fed via resistor 198 is also fed to the base of a power transistor 206 which is connected in a ganged configuration with power transistors 208 and 210. The emitter of the transistor 210 is connected to a lamp 212 for energization thereof.
The input of the unijunction transistor 188 is connected via a lead 214 to a variable resistance 216 which is connected in series with a fixed resistance 218. A capacitor 220 is connected between circuit ground and the resistance 218. A resistor 222 is connected to one terminal of the capacitor 220, with a Zener diode 224 connected in series with the resistor 222 and the emitter of a transistor 226. The base of the transistor 226 is connected to a diode 228, the cathode of which is connected to a suitable switch 230. A positive bias voltage is applied via lead 232 to the transistor 226.
A terminal of the capacitor 220 is also connected to the cathode of a Zener diode 234 and to the base of a transistor 236. The anode of the diode 234 is connected to the variable arm of a potentiometer 238 connected between a supply of reference voltage and ground. Transistor 236 is connected in a ganged configuration with power transistors 240 and 242. The emitter of transistor 242 drives a lamp 244.
In operation, the circuitry of FIG. 8 operates in a similar manner to the circuit shown in FIG. 7. The minimum speed desired for the circuit is set by adjustment of the potentiometer 238. The frequency of operation of the circuit is determined by the voltage applied to the input of the unijunction transistor 188. Transistor 188 generates a stream of spiked pulses which drives the multivibrator comprising transistors 190 and 192. The square wave output of the multivibrator is fed into the integrator circuit comprising resistance 198 and capacitors 200 and 202. The resulting integrated waveform drives the power resistors 206-210 to selectively energize the lamp 212.
The variable resistance 204 is varied according to the illumination of the lamp 244, which in turn is controlled by the operation of the transistors 236, 240 and 242. The switch 230 may be manually operated in order to selectively control the time constant of the integrator network as desired.
Whereas circuits have been disclosed for providing a generally triangular 'or sinusoidal energization wave to control a light source, other waveform shapes such as trapezoidal or the like may also be advantageously utilized. The particular shape of the waveform utilized to drive the light source for variance of the filters in the tremulant circuits will be chosen according to the desired tonal qualities.
FIG. 9 shows a preferred embodiment of a system utilizing two channels of series connected tremulant circuits according to the invention. The output from a musical instrument is fed into the system via a capacitor 250 to the base of a transistor 252. The emitter of the transistor 252 is connected to a first filter network comprising capacitors 254 and resistors 256 having component values designed to filter a selected band of relatively high audiofrequencies.
For instance, the filter network comprising capacitors 254 and resistors 256 may filter frequencies up to about 800 cycles per second, while the filter network comprising resistors 258 and capacitors 260 may filter frequencies above 800 cycles per second. In some instances, a fixed shunt resistance may be desirably connected to the circuit to eliminate the modulation of frequencies under a predetermined frequency, such as about 130 cycles per second, in order to more accurately simulate the operation of a mechanical rotating speaker of the Leslie type.
The output from the high pass filter network is fed into four series connected, identical tremulant circuits 262, 264, 266 and 268. Only one of the tremulant circuits is illustrated for simplicity, with each of the circuits comprising a feedthrough capacitor 270 connected to the base of a transistor 272. Con ventional bias resistors are connected between transistor 272 and a source of bias voltage.
The collector of the transistor 272 is connected to the base of atransistor 274, while the emitter of the transistor 272 is connected to the base of a transistor 276. The signals fed to the bases of transistor 274 and 276 will thus have frequencies above the preselected cutoff frequency of the filter and will also have phases l out of phase. The emitter of the transistor 274 is fed to a high pass filter comprising capacitor 278 and variable resistor 280. The signal appearing at the emitter of the transistor 276 is fed through a capacitor 282 and a grounded resistor 284 to a variable resistor 286. A capacitor 288 in connected between ground and one terminal of the variable resistor 286 to provide a low pass filter.
The common terminal of the capacitor 278 and the resistor 280 is connected to a resistor 290, while the common terminal of the resistor 286 and the capacitor 280 is connected to a resistor 292. The combined outputs of the two filters are fed through a capacitor 294 to the base of an amplifying transistor 296. The, collector of the transistor 296 is connected to the input of the next series connected tremulant circuit 264.
A variable light source 298 is driven by a signal generator 300, which may preferably comprise the circuit illustrated in FIG. 8. The light from the light source 298 impinges directly upon the variable resistances in each of the tremulant circuits 262-268. The output from the series connected tremulant circuits is fed through a resistor 302 and a feedthrough capacitor 304 to the base of an amplifying transistor 306. The amplified output is fed through a capacitor 308 to the base of a transistor 310. The emitter of the transistor 310 generates the modulated output signal provided by the system.
The signal passed through the low pass filter comprising resistor 258 and capacitors 260 is fed tothree series connected identical tremulant circuits 320, 322, and 324. Each of the tremulant circuits comprises a feedthrough capacitor 326 which supplies the filtered audiofrequency signal to the input of the transistor 328. The collector of transistor 328 is connected to a capacitor 330 and a variable resistor 332 for filtering a selected low frequency band. The emitter of the transistor 328 is fed through a resistor 334 connected to circuit ground and also to a capacitor 336. A resistor 338 is connected between ground and a capacitor 336 and is also connected to one terminal of a variable resistor 340 accordingto the invention.
A capacitor 342 is connected to the other terminal of the resistor 340 and also to a terminal of a resistor 344. The common terminals of the capacitor 330 and the variable resistor 332 are connected to a resistor 346. The common terminals of resistors 344 and 346 are connected through a capacitor 348 to the base of an amplifying transistor 350. The collector of the transistor 350 is applied to the next tremulant circuit 322. A lamp 352 is controlled by a signal generator system 354, which in the preferred embodiment is similar to the circuit shown in FIG. 8. The output of the third tremulant circuit 324 is fed via a resistor 356 through the capacitor 304 to the output transistors 306 and 310. It is important to note that the signal generators 300 and 354 are preferably operated at different rates in order to produce unique musical effects which signal to the base of a transistor 362. A capacitor 364 is connected across the transistor 362. The emitter of transistor 362 is applied to the base of a transistor 366. The collector of the transistor 366 is fed through an R-C network 368 to the base of a transistor 370. The cathode of a diode 371 is connected between the network 368 to circuit ground. The collector of the transistor 366 is applied via a resistor 372 to the collector of the transistor 370. The collector of the transistor 370 is fed through a capacitor 374 to the primary of a loosely coupled transformer 376.
The secondary of the transformer 376 is fed via a capacitor 378 to the base of a transistor 380. The collector of the transistor 380 is fed via a capacitor 382 to the base of a transistor 384, the collector of which is fed via capacitor 386 to a potentiometer 388. The movable arm 390 of the potentiometer is fed through a capacitor 392 and a resistance 394 to the base of the transistor 306.
In the operation of the circuit shown in FIG. 9, high frequencies of the audiofrequency signal are passed through the series connected tremulant circuits 262-268, whereupon the frequency responses of the parallel filters in each of the tremulant circuits are varied according to the output of the lamp 298. The crossover frequencies of the filters in each of the tremulant circuits are concurrently increased and decreased as previously described to induce a tremulant effect into the selected high frequency band of the input signal. The series connection of a plurality of tremulant circuits provides a relatively high slope to the filter frequency responses in the area of the crossover frequency, thereby imparting relatively sharp changes of phase and amplitude to the modulated signal.
A similar tremulant effect is induced into the lower frequencies of the input signal which are fed through the three series connected tremulant circuits 320-324. When the outputs from the high and low frequency channels are combined and fed through the capacitor 304 to the amplifying output circuitry, a very unusual and pleasing musical effect is obtained. The provision of the reverberation circuitry induces additional unusual tonal qualities to the output signal. The relative rates of the signal generators 300 and 354 may be selectively operated by the musician to very accurately simulate the use of mechanical rotating Leslie-type speakers. The system shown in FIG. 9 is not limited to use with any particular musical instrument, but is unilaterally compatible with almost any source of audiofrequency signal.
As previously mentioned, it is often desirable to mix the tremulant output with the nontremulant output. Such a system is indicated generally by the reference numeral 500 in FIG.
10. The output of a music source 502 is coupled to the input of the phase shift network 506. The output of the phase shift network 506 is summed with the output from the music source 502 through a summing network comprised of resistors 508 and 510, and is then applied to an audio generator system 512. The capacitors 504 and 514 provide DC isolation from the phase shift network 506. v
The music source 502 may be any means for producing electrical signals representative of a musical sound, such as an electric organ, electric guitar, microphone pickup for voice, record playing system, or any preamplifier system for musical sounds. The audio generator 512 includes the necessary amplifier system for musical sounds. The audio generator 512 includes the necessary amplifiers to drive a speaker system as well as the speaker system for producing musical sounds from the electrical signals representative of the musical sound.
The phase shift network 506 preferably produces at least 720 of phase shift. Any phase shift network may be used for this purpose so long as the amplitude of the signal remains substantially constant as the phase is shifted. For example, stages 262, 264, 266 and 268 of FIG. 9 may be used to provide substantially 720 of phase shift without amplitude attenuation if the upper and lower fillers of each stage are properly matched. The resistors 508 and 510 in the summation network are preferably chosen so that the output of the phase shift network 506 and the output of the music source 502 are mixed in approximately equal proportions for application to the audio generator 512. For commercial applications, it is desirable to make the mixing ratio variable within a relatively narrow range and within the control of the musician so that he can vary the depth of amplitude modulation as hereafter described.
' The operation of the system 500 is illustrated in FIGS. 12-14. FIG. 12 illustrates the phase shift produced by the phase shift network 506 for each scan cycle of the lamp 298. The zero point of the cycle has been selected as a point of maximum illumination, and the 180 point of the cycle representing minimum illumination. The light dependent resistors vary in resistance inversely with illumination. The curve 520 represents the phase shift of a single frequency which may be considered a wide-range musical note. As a result of the cyclic phase shift f from 0 to a lag of 720, the frequency is modulated as represented by the curve 522 in FIG. 13. By mixing the output from the phase shift network 506 with the original signal from the music source 502 in approximately equal ratios, the amplitude is modulated as represented by the curve in FIG. 14. The combination of the frequency modulation as represented in FIG. 13 and the amplitude modulation as illustrated in 14 produces a sound substantially identical to the sound of a rotating speaker such as represented in the schematic view of FIG. 10.
Referring to FIG. 10, it will be noted that when the speaker is in the position shown in said outline and designated 0," a person positioned in front of the speaker would hear a maximum amplitude signal with no apparent change in frequency. When the speaker reaches the position, the amplitude becomes nulled because the signal emanating from the rear end of the speaker is out of phase from the signal emanating from the front of the speaker and of approximately equal amplitude, thus theoretically cancelling along an interface disposed generally 90 to the axis of the speaker. When the speaker reaches the 180 position, substantially full amplitude is again received because of the signal emanating from the rear of the speaker, and the listener is unable to detect the 180 phase difference in the signal. When the speaker reaches 270, a null once again occurs.
If the speaker is located in a square cabinet, a large number of amplitude nulls occur at various positions of the speaker, depending also upon the position of the listener. However, whenever the speaker is pointing either directly toward or directly away from the listener, the direct sound tends to override the cancellation caused by out-of-phase reflections. It has generally been known that as the speaker rotates, each frequency will decrease to a minimum point at the 90 position, will return to the nominal frequency at the 180 position, and will change to a maximum at the 270 position, and will change to a maximum at the 270 position as a result of the Doppler effect.
Now comparing the sound produced by the rotating speaker 525 to the sound depicted by the curves in FIGS. 12-14, it will be noted that the frequency of the sound produced by the system 500 decreases to a minimum at the 90 position because that is the position of greatest rate of phase shift, passes back through the norm represented by the dotted line 523 at the 180 position, reaches a maximum at the 270 position, and again returns to the norm at the 0 position. More importantly, because of phase cancellation in the summing network 508-570, the amplitude (FIG. 14) reaches a null at approximately the 75 position as a result of the 180 phase difference in the two signals, reaches a maximum at the 90 point as a result of the reinforcement at the 360 phase shift, and again is nulled at about the 105 position as a result of the out-of-phase cancellation at a phase shift of 720, which is an inphase condition. The same double null condition is produced in the vicinity of the 270 position.
In reality, the double null resulting from 720 of phase shifts is more pleasant to the ear and more closely simulates the effect of a rotating speaker contained in a cabinet than does a single null which would be produced by 360 of shift,-even when the total frequency shift is much greater than 360 using a circuit hereafter described which produces the multiple reflections and cancellation patterns. It should also be noted that lines 520, 522 and 524 represent but a single frequency.
The other frequencies in the audio range will be shifted at different points in the scan cycle as represented by the dotted lines 520l-I for the high frequencies and 520L for the low frequencies. Thus the null points actually occur at points ranging between positions S24I-I and 524L in FIG. 14, depending upon the frequency content of the musical sounds. As a result, the ear cannot detect the two distinct null positions, but instead detects a dynamically changing amplitude pattern that very closely simulates a rotating speaker in a cabinet.
From FIG. l2, l3, and 14 it will be noted that as long as the phase shift 4 =21m where n is an interger, the shape of the frequency and amplitude curves 522 and 524 will remain substantially the same except that the number of nulls at the 90 and 207 positions will be equal to n. Due to the sinusodial slope of the phase shift curve 220, however, these nulls remain closely bunched at a 90 and 270 with relatively broad full amplitude lobes 524A at the and 180 positions. In general, a greater number of nulls around the 90 and 270 positions enhances the effect of creating the sound of a rotating speaker within a square cabinet. Phase shifts of about 1,080 and about 1,440 produce particularly pleasing results.
The effect of the increasing the total phase shift in each cycle is to increase the extent modulation excursions of the frequency. Since the rotational rate, i.e., the scan rate of the light-dependent resistors, remains constant, the overall effect is a simulation of rotatinga speaker around a circle of larger diameter at the same rate of rotation.
As previously mentioned, it is desirable for the resistors 508 and 510 to be selected so as to sum to output from the phase shift network 506 and the music source 502 in approximately a 1:1 ratio. By referring to FIG. 14, it will be noted that such a ratio produces a theoretical null on each side of the 90 and 270 positions of the speaker. As the proportion of the output from the music score 502 is reduced, the amplitude modulation is softened, which is desirable for some types of music. For this reason, it may be desirable to provide an adjustable mixing network to permit the musician to adjust the summation ratio. However, as the output from the music source 202 is reduced below half that from the network 506, the effect of a rotating speaker is substantially lost, with the effect being just straight vibrato.
Quite a different effect occurs as the proportion of signalv from the music source 502 is increased above that portion supplied by the phase shift network 506. By the time the proportions from the music source 502 reaches about 1.5 times that from the phase shift 506, the nonvibrato sound begins to dominate and the effect of a rotating speaker is greatly diminished. By the time the signal from the music source 502 is twice that from the phase shift network 506, the effect of a rotating tremulant effect is produced. As the proportion of nonshifted signal increases, the effect becomes similar to the conventional chorus vibrato presently available on some domestic organs.
FIGS. 15, 16 and 17 illustrate the frequency modulation which are produced when the phase shift network 56 produces a phase shift d =21m+l. In FIG. I4, a phase shift of 540 represented by line 526.'This produces an amplitude modulation represented by the line 528 in FIG. 16 that is of the same phase heretofore described, but has lower excursions of apparent frequency. However, the amplitude as represented by curve 530 is substantially different due to the fact that when the scan cycle is equivalent to the 180' position of the speaker, the signal from the music source 502 is 180 out of phase from the output of the phase shift network 506, thus producing a relatively long null during every other half cycle. This contradicts the condition of substantially maximum amplitude produced by a speaker having equal efficiency toward both the front and rear. It will also be noted that the nulls at the 90 and 270 speaker positions are shifted toward the l position. 1
In the lower degrees of phase shift, for example 540, this condition is not as pleasant to the ear as a rotating speaker. However, as the degree of phase shift is increased, this undesirable effect seems to diminish.
This may be due to the fact that the null at the l80 position is narrowed and thus lasts for a shorter period of time, because the test circuit did not produce the exact degree of phase shift required to produce a noticable null.
An alternative embodiment of the system of the present invention is indicated by the reference numerals 540 in FIG. 20. System 540 includes two of the phase shift networks of 506 of FIG. 10, with the associated summation networks comprised of resistors 508 and 510, connected in series. Corresponding components are therefore designated by the same reference characters in FIG. 20 as in FIG. 10. The effect of the circuit 540 is to provide a large degree of phase shift without producing a large number of nulls at the positions of rotation. Thus, if each of the phase shift networks 506 produces only 360 of the phase shift, for example, only a single null positioned directly at the 90 and 270 positions will be produced.
Similarly if two circuits having 720 of phase shift are connected in series, a total of l,440 of phase shift will be produced with only two nulls at the 90 and 270 positions.
A preferred embodiment of the phase shifted network is indicated at generally by the reference numerals 550 in FIG. 18. The phase shift network 550 is comprised of four identical stages in which corresponding components are designed by the same reference numerals. For example, each stage is comprised of a transistor 552 with the appropriate collector, emitter and base biassing network comprised of resistors 554, 556, 558 and 560. A capacitor 562 couples the collector of the transistor 552 to an output 564. A light-dependent resistor LDR 566 couples the emitter to the output 564. The output 564 is coupled by a capacitor 568 to the input of the next stage. Capacitors 570 and 572 provide DC isolation for the phase shift network.
The resistance of light dependent resistors 566 varies with illumination from a low value to a high value such that each stage produces a phase shift which varies from approximately 0 when the LDR is fully illuminated to approximately l80 when the LDR is dark. The signal from the music source is input at 574 and is summed with the output from the last stage of the phase shift network by a summation resistor 576, as heretofore described. The center tap of the resistor 576 is preferably adjustable so that the musician may vary the ratio in which the two signals are mixed from a ratio of about 2:1 to a ratio of about 1:2. When using four stages as illustrated, the total phase shift is approximately 720, although the phase shift does not reach 0 or does it reach a full 720 due to the inefficiency of the individual stages.
Referring now to FIG. 20, another circuit in accordance with the present invention is indicated generally by the reference numeral 600. The system 600 includes an input 602 for receiving a signal from a music source and an output 604 for connection to an audio generator as heretofore described. The phase shift circuit 600 has a plurality of identical stages S -S which, it will be noted, are identical to the stages of the phase shift network 550. However, the last stage 606 includes a pair of variable resistors 608 and 610. Resistor 608 is connected in series with the light-dependent resistor 612 and resistor 610 is connected in parallel. This resistor circuit provides a means for adjusting the total phase shift of the network to a selected value, within a limited range. For example, the combined resistance of the parallel branches of resistors 608 and 610 determine the minimum phase shift of the stages when the LDR 612 is fully illuminated and therefore at a minimum resistance value. The value of resistor 610 determines the maximum phase shift of the stage when the LDR is dark. It will be noted that the phase shift produced by stage 606 is taken from the linear part of the phase shift curve, and is therefore also useful in shaping the phase shift curve 120 of FIG. 12, for example, as will be presently described.
The signal at the input 602 is also passed through a phase shift stage 614 which has a variable resistor 616 in place of an LDR. The variable resistor 616 may be adjusted so as to synchronize the signal from the output of the phase shift stage 606 with the signal from the music source applied to input 602. Resistors 618 and 620 provide a summation in network as heretofore described for summing the signal from stages 606 and the signal from stage 614. Capacitors 622 and 624 provide DC isolation. Thus the adjustable stages 606 and 614 provide a means for precisely synchronizing the output of stage 614 with the output of stage 614 at the position, i.e., when the LDRs are fully illuminated, by adjusting resistor 616. Then the total phase shift network can be adjusted by resistors 608 and 610, so that the phase relationship between the two signals at the 180 position, i.e., when the LDRs are dark.
It will also be noted that the phase shift stage 606 can be used to shape the phase shift curve 520, for example, of FIG. 12. Since the maximum and minimum resistance of the network between the emitter and the output of the stage are fixed, only the linear portion of the phase shift curve is used. By using the plurality of such phase shifts, the slope of the curve 520 in the vicinity of the 90 and 270 position speaker may be increased substantially without effecting the slope of the curve in the vicinity of the 0 and 180 positions. The shape of the curve 520 may also be adjusted by the manner in which the light dependent resistors are varied.
As previously mentioned, the shape of the amplitude curve 524 is determined by the relative phase relationship of the signals applied to the summation network resistors 618 and 620, for example. Thus by making the phase shift at the l80 position something other than an in-phase condition, the amplitude of the large lobe at the 180 position can be reduced as desired. For example, if a speaker to be simulated is slightly less efficient to the rear of the speaker than in front of the speaker, the total phase shift may be made 780, for example, and thereby produce some attenuation at the 180 position. It will be noted that such an increase in phase shift does not materially change the positions of the nulls at the 90 and 270 positions. Similarly, the total phase shift be made slightly lessv than 720.
Another effect can be produced by substituting a phase shift stage like stage 606 in FIG. for phase shift stage 614, and providing a second light which is cycled at a slower rate for illuminating the LDR. Such a system provides a means for continually shifting the positions of the amplitude lobes on the amplitude curves of FIG. 14 relative to the apparent position of the speaker. This simulates the rotation of a square cabinet about the rotating speaker at a slower speed, and produces an unusual and pleasing musical effect for the more modern types of music.
Whereas the present invention has been described with respect to several specific embodiments thereof, it will be understood that various modifications and changes will be suggested to one skilled in the art, and it is intended to encompass these changes and modifications as fall within the true scope of the appended claims.
What is claimed is:
1. A system for modulating a signal comprising:
a. a plurality of parallel-connected filter means each discriminating against different frequencies,
b. means for applying electrical signals representative of said signal with different phases to said filter means,
c. means for varying the frequency response of at least one of said filter means, and
d. means for recombining the outputs of said filter means with the signal being modulated.
2. The system of claim 1 wherein one of said filter means discriminates against frequencies below a preselected frequency and another of said filter means discriminates against frequencies above said preselected frequency.
3. The system of claim 2 wherein the electrical signal applied to said one filter means is l out of phase with the electrical signal applied to said another filter means.
4. The system of claim 1 wherein the frequency response of a pair of said filter means is varied to modulate both the frequency and the amplitude of said signal.
5. A modulation system for audiofrequency signals comprising:
a. first filter means for discriminating against frequencies below a first rollover audiofrequency,
b. second filter means connected in parallel with said first filter means and discriminating against frequencies above a second rollover audiofrequency such that the cutoff frequency responses of said first and second filter means overlap at a crossover frequency,
c. means for applying audiofrequency signals to said first and second filter means in phase opposition,
d. means for cyclically varying said crossover frequency by increasing the rollover frequency of one of said filter means while concurrently decreasing the rollover frequency of the other of said filter means, and
e. means for combining the outputs of said first and second filter means.
6. The system of claim 5 and further comprising means for selectively varying the frequency of the cyclical variations in the frequency response of said first and second filter means.
7. The system of claim 5 wherein each said filter means comprises:
a series connected capacitor and resistor tuned to discriminate against preselected frequency bands, said resistor being variable to change the preselected frequency bands.
8. A system for modulating the audiofrequency output signal of a musical instrument comprising: 1
a. a first filter network for discriminating against low frequencies connected to receive said output signal,
b. a second filter network for discriminating against high frequencies connected in parallel to said first filter network,
c. first and second modulation networks each connected to one of said filter networks, each modulation network including parallel high and low bandpass filters for operating on signals of opposed phases,
d. means for varying the frequency responses of said modulation networks, and
e. means for combining the outputs of said modulation networks.
9. The system of claim 8 wherein each of said modulation networks includes a tremulant circuit comprising:
a. means for generating two signals in phase opposition each representative of said audiofrequency output signal,
b. first filter means for discriminating against frequencies below a first rollover frequency connected to receive one of said two signals,
c. second filter means connected in parallel with said first filter means for receiving the other of said two signals and for-discriminating against frequencies above a second rollover frequency so that the cutoff frequency responses of said filter means overlap at a crossover frequency, said d. means for varying the magnitude of said crossover frequency by variance of elements in said filter means.
10. The system of claim 9 wherein said first modulation network comprises four of said tremulant circuits connected in series and said second modulation network comprises three of said tremulant circuits connected in series.
11. The method of modulating an electrical signal comprising:
.li a. filtering different frequency bands from a plurality of parallel phase-shifted signals each representative of said electrical signal, b. varying the frequency bands filtered from said phaseshifted signals, I c. combining the filter signals, and
d. varying the rate of variance of the frequency bands filtered from said phase-shifted signals.
12. The method of claim 11 wherein said phase-shifted signals are 180 out of phase.
13. The method of claim 1 l and further comprising:
a. filtering frequencies above a selected first frequency from one of said phase-shifted signals, and
b. filtering frequencies below a selected second frequency from another of said phase-shifted signals.
14. The method of imparting tremulant effects into an audiofrequency output from amusical instrument comprising:
a. generating a pair of parallel phase-shifted electrical signals each representative of said output,
b. filtering frequencies above a preselected reference frequency from one of said electrical signals,
c. filtering frequencies below said reference frequency from the other of said electrical signals,
d. combining the filtered electrical signals, and
e. varying said reference frequency.
15. The method of claim 14 wherein the phases of said electrical signals are 180 out of phase.
16. The method of claim 14 wherein the cutoff frequency responses of said filtered electrical signals overlap at a crossover frequency.
17. The method of claim 16 wherein said crossover frequency occurs at the SO-percent level of each of said cutoff frequency responses.
18. In a tremulant system for musical sound the combination of:
first means for producing a first electrical signal representative of sound produced by a musical instrument,
second means for cyclically shifting the phase of said first electrical signal to produce a second electrical signal in which the phase angle is cyclically varied relative to the first electrical signal by an angle greater than about 540, and
third means for summing said first electrical signal and said second electrical signal to produce a third electrical signal that is both frequency and amplitude modulated.
19. The system of claim 18 wherein the first and second signals are summed in a ratio ranging from about 1:2 to about 2: l.
20. The system of claim 18 wherein the phase of the first electrical signal is shifted through an angle approximately equal to 21m, where n is an integer.
21. The combination defined in claim 18 further characterized by:
fourth means for producing audible sounds in response to the third electrical signal.
22. The combination defined in claim 21 wherein the first means includes a musical instrument,
23. The combination of claim 18 wherein the second means includes means for selectively shifting the phase of the first electrical signal at a first cyclical rate of from about 3 to about l cycles per second, and at a second cyclical rate of from O to about 3 cycles per second.
24. The combination defined in claim 23 wherein the second means further includes means for changing from one cyclic rate to the other cyclic rate in a predetermined manner to stimulate the inertial characteristics of a rotating speaker.
25. The combination defined in claim 18 wherein the second means includes:
means for cyclically shifting the phase of said first electrical signal at a vibrato rate, and
means for iniating the cyclic phase shift by progressively increasing the cyclical rate to stimulate the inertial characteristics of a rotating speaker.
26. The combination defined in claim 18 wherein the second means includes:
means for cyclically shifting the phase of said first electrical signal at a vibrato rate, and
means for terminating the cyclic phase shift by progressively decreasing the cycle rate to stimulate the inertial characteristics of a rotating speaker.
27. The tremulant system comprising means for producing a sound from a stationary speaker that is tremulated at a rate such as to simulate the sound produced by a speaker rotating at said rate, said means including means for progressively initiating and progressively terminating the tremulant rate to simulate the inertial characteristics of the rotating speaker.
28. The method for producing a musical sound from a fixed speaker approximating that produced by a rotating speaker which comprises cyclically shifting the phase of a first signal representative of the musical sound at a rate corresponding to the rate of rotation of the speaker, said phase being shifted through a phase angle in excess of about 540 to produce a second signal, and summing the first signal and the second signal to produce a third signal representative of musical sound produced by a rotating speaker.
29. The method of claim 28 wherein the first and second signals are summed in a ratio between about 2:1 and about 1:2,
30. The method of claim 29 wherein the phase of the first signal is shifted through a phase angle of about =21rn where n is an integer.
31. The tremulant system for producing musical sound comprising:
variable phase shift means for shifting the relative phase of first and second signals each representative of the same musical sound, the variable phase shift means comprising a plurality of variable phase shift stages, each stage comprising a transistor, a biasing network for the transistor, a capacitor connecting the collector of the transistors to the output of the stage, and a variable resistance connecting the emitter of the transistor to the output, the variable resistance of each stage being varied by said means for cyclically varying the degree of phase shift of the phase shift means,
means for cyclically varying the phase shift means so as to vary the degree of relative phase shift from a minimum value to a maximum value at a selected vibrato rate, and
a summation network for summing the first and second signals to produce a tremulant signal.
32. The tremulant system of claim 31 wherein at least one of the stages includes means for selectively limiting the maximum and minimum values of the variable resistance to adjust the total cyclic phase shift produced by the phase shift means.
33. The tremulant system for producing musical sound comprising:
variable phase shift means for shifting the relative phases of first and second signals, each representative of the same musical sound, I means for cyclically varying the phase shift means so as to vary the degree of relative phase shift from a minimum value to a maximum value at a selected vibrato rate, a summation network for summing the first and second signals to produce a tremulant signal, and phase shift means for selectively varying the relative phase of the first and second signals at the extremes of phase shift produced by the phase shift means.
34. The tremulant system for producing musical sound comprising:
variable phase shift means for shifting the relative phase of first and second signals each representative of the same musical sound,
means for cyclically varying the phase shift means so as to vary the degree of relative phase shift from a minimum value to a maximum value at a selected vibrato rate, and
a summation network for summing the first and second signals in an amplitude ratio from about 2:l to about l:2 to produce a tremulant signal.
35. The tremulant system for producing musical sound comprising:
variable phase shift means for shifting the relative phase of first and second signals each representative of the same musical sound, means for cyclically varying the phase shift means so as to vary the degree of relative phase shift from a minimum value to a maximum value greater than the minimum value by about 540 at a selected vibrato rate, and a summation network for summing the first and second signals to produce a tremulant signal. 36. The tremulant system for producing musical sound comprising:
first and second circuits each comprising variable phase shift means for shifting the relative phase of first and second signals each representative of the same musical sound, means for cyclically varying the phase shift means so as to vary the degree of relative phase shift from a minimum value to a maximum value at a selected vibrato rate, a summation network for summing the first and second signals to produce a tremulant signal,
high pass filter means in the first circuit for excluding a low frequency range,
low pass filter means in the second circuit for excluding a high frequency range, and wherein the phase shift means in the first and second circuits are cyclically varied at different vibrato rates.
37. The tremulant system of claim 36 wherein the phase shift means in the first circuit produces a relative phase shift of about 720, and 1 the phase shift means in the second circuit produces a relative phase shift of about 540.
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