|Publication number||US3714655 A|
|Publication date||Jan 30, 1973|
|Filing date||Sep 30, 1970|
|Priority date||Sep 30, 1970|
|Publication number||US 3714655 A, US 3714655A, US-A-3714655, US3714655 A, US3714655A|
|Inventors||Ross G, Susman L|
|Original Assignee||Sperry Rand Corp|
|Export Citation||BiBTeX, EndNote, RefMan|
|Referenced by (5), Classifications (7)|
|External Links: USPTO, USPTO Assignment, Espacenet|
Ross et al.
Jan. 30, 1973 ARRAY ANTENNA SIGNAL PROCESSING SYSTEM  Assignee:
Inventors: Gerald F. Ross, Lexington, Mass; Leon Susman, Sudbury, both of Mass.
Sperry Rand Corporation, New
Sept. 30, 1970 Appl. No. 76,937
|4oo 1. in
Primary Examiner-Benjamin A. Borchelt Assistant ExaminerG. E. Montone Attorney-S. C. Yeaton  ABSTRACT A electromagnetic energy coupling network for reciprocally processing or combining electromagnetic energy flowing in pluralities of transmission lines is disclosed. The coupling element is a multiple port transmission line junction associated with tapered transmission lines and efficiency transferring energy inputs on such lines into a wave flowing only from a single output port. The network may be employed individually or in multiple quantities as an element in complex coupling tree network matrices for the processing of individual impulse signals received by the elements of an antenna array. In application in such signal combining tree matrices, means are provided in the matrix depending upon inherent properties of the coupling element for degrading the adverse effects of internally generated spurious signals.
4030 A A c.
I T T T 50l SWITCHING E 502 503 SYNCHRONIZER 5050 E UTILIZATION APPARATUS PATENIEUmao I973 SHEET 3 OF 5 G b m 4 G M G H II m El B F u E H T T A C m r L M p C s L t A n W u r A l O W n F O. W O W mSo E h 2 62;
(b) RESPONSE OFF ANGLE.
T (C)DEFINITION OF BANDWIDTH FIG? INVENTORS GER/1L0 F. R055 LEO/V SUSMA/V ATTORNEY PATENTEDJAI 30 I973 3,714.6 55 SHEET u 0F 5 a (1) ANOMOLOUS RETURN DUE TO RE-REFLECTION 6OIA ll llllnllnllll IIIIIIIIIII'IVI" OUTPUT WAVEFORM-MODE A-DELAY A SECONDS IN THE OUTPUT LINE 500.
V0 LTS REFLECTION OUTPUT WAVEFORM'MODE B- DELAY ASECONDS IN EACH ANTENNA ELEMENT LINE. VOLTS WVM 602 v 0) v m NO RE-REFLECTION ATTORNEY PATENTEDJAN30 I975 3.714.655 SHEET 5 BF 5 FROM 505 A 706 702 707 7020 D'SPLAY TIME H C BASE I GENERATOR 7o2b L l 704 l 708 701 703 T 705 I I I SAMPLING SAMPLE AND SAMPLE GATE HOLD CIRCUIT i AMPLIFIER l FROM 504 (MODE A) l :GOOA :GOIA T (MODE 6) News 6OIB 7055 725 724 DIGITAL TO I ANALOG MULTIPLIER CONVERTER 750 REGISTER 1 1 ANALOG TO DIGITAL x 7231: J I CONVERTER 4 DISPLAY REGISTER @327 1 r722 3| l PROGRAMMER 1 I MEMoRY INVENTORS GERALD F R055 FROM 505A 72! LEO/V SUSMA/V ATTORNEY ARRAY ANTENNA SIGNAL PROCESSING SYSTEM BACKGROUND OF THE INVENTION 1. Field of the Invention The invention pertains to combining networks or tree matrices for combining or reciprocally processing electromagnetic waves flowing in pluralities of transmission lines and to means for reducing spurious responses therein.
2. Description of the Prior Art Combining or processing networks for efficiently combining coherent electromagnetic energy, especially such energy when of impulse duration, are recently of increasing interest for processing electromagnetic signals collected by multiple-collector array antennas. They are of interest in this and other such applications wherein a plurality of coherently applied signals is processed by combining networks to yield a maximum signal at a single output port. Generally, combining networks of the past have not had capabilities of efficient operation over a multiple-octave instantaneous frequency band when excited by input signal wave forms of sub-nanosecond duration.
The prior art has recognized that, in order to construct a coupling network or matrix for combining or processing a plurality of coherent signals, a coupling element is required having special properties. The element may, for example, be a three-port junction and it must maximize the signal output at one of its ports; i.e., it must with maximum efficiency transfer any energy input at two of its ports to a third or output port. Only this behavior results in the desired minimum over all network distortion or dispersion.
Known junctions have not filled the described need. For example, known types of high frequency or microwave directional couplers introduce time-domain distortion. To use such coupler devices, there must generally be tolerated a compromise trade off of efficiency against distortion. One type of directional coupling network minimizes distortion, but requires an impractical number of coupling elements. Attempts to use other four-port biconjugate network elements in combining or signal processing networks or matrices have generally resulted in poor efficiency and severe dispersion.
SUMMARY OF THE INVENTION The invention pertains to novel high frequency or microwave transmission line coupling networks or matrices and to applications thereof. The invention utilizes a bilateral elemental coupling device which may be used in multiple quantities in tree matrices for forming complex coupling system networks. The elemental coupling device is a three port transmission line junction employing specially tapered input and output transmission lines associated with a tee junction. Signal energy entering the two symmetric ports of the junction is transferred in total to a wave emanating from the third port, and vice versa.
The coupling device has use in coupling matrix or tree network systems for application in novel subnanosecond antenna arrays in which the contributions of a number of effectively discrete but coherent sources are summed by the coupling tree matrix according to a novel technique. An output pulse from a single port of the matrix or network represents a substantially perfect summation of the total energy of the plurality of effectively discrete sources. The inherent properties of such sub-nanosecond matrices are employed with switching, delay, and signal amplitude multiplication means for substantially eliminating the effects of signals spuriously generated within the matrix.
BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is an isometric view, partly in section, of an embodiment of the coupling element employed in the invention.
FIG. 2 is a plan view of the coupler element of FIG. 1 used in explaining its operation.
FIG. 3 is a circuit diagram illustrating a novel application of the coupler element in an antenna array signal processing system.
FIGS. 4 and 5 are graphs used in explaining the operation of the apparatus of FIG. 3.
FIG. 6 is a block diagram showing in detail parts of the processing system of FIG. 3.
FIG. 7 is an explanatory graph.
DESCRIPTION OF THE PREFERRED EMBODIMENT In FIG. 1, there is shown a planar microcircuit useful at microwave or very high frequencies as a bilateral electromagnetic wave transmission line energy coupler circuit. The circuit shown represents a fundamental transmission line coupling network concept which may be used according to the invention as a basic element in complex coupling networks or matrices and elsewhere.
The transmission line device of FIG. 1 comprises at least a dielectric substrate 1 to one surface of which a relatively thin conductive ground sheet 2 may be bonded in any well-known manner. For example, the ground sheet 2 may be formed on one surface of the dielectric substrate 1 by evaporation of a suitable metal in a vacuum chamber from a heated source for distilling the desired conductive metal or by chemical electroplating or by other known metal plating methods.
The transmission line opposite ground plate 2 comprises, for example, planar or microstrip transmission line elements bonded to the second or upper surface of substrate 1. The transmission line system on the upper surface of substrate 1 may be bonded to substrate 1 by well known methods, including methods of the type employed to generate and bond the ground sheet 2.
The planar microstrip circuit of FIG. 1 consists of a three-port transmission line junction having properties which will be further discussed. The three-port junction has symmetric arms 3 and 3a, both conductively joined via a coupling region 4 to a single planar transmission line 5. It is observed that branch lines 3 and 3a stem from the mutual coupling region 4, following oppositely directed doubly-arcuate segments 6 and 6a. Segments 6 and 60 form smoothly curved transmission paths which ultimately become substantially parallel to the single transmission line 5, being conductively continued from the point where parallelism is reached at junctions 7 and 7a by tapered transmission line segments 8 and 8a to further parallel transmission line sections 9 and 9a. It is to be understood that the parallel relation between line segments 5, 9, and 9a is one of convenience for use in many applications, but that other angular relations between these line segments are more convenient for use in other applications.
Sections 9 and 9a have widths transverse to the directions of energy propagation substantially equal to the corresponding width of transmission line 5. The branch transmission line section including doubly arcuate sections 6 and 6a has, however, a lesser width than line segments 5, 9, and 9a. The tapered sections 8 and 8a have non-parallel sides adjusted so as smoothly and continuously to join section 6 to section 9 and section 6a to section 911, respectively.
Referring still to FIG. 1, it should be apparent that it is possible to extend dielectric sheet 1 and ground plate 2 in the direction of transmission line so as to accommodate an extension of the latter or to support additional active or passive high frequency or other circuits in any combination desired to interact with the elements shown in FIG. 1. Likewise, substrate 1 and ground plate 2 may be extended in any desired direction to support additional active or passive circuit elements in any combination desired to interact with transmission lines 9 and 9a.
FIG. 2 is a simple outline only of the microcircuit bilateral coupler circuit shown in FIG. 1, as will readily be seen by comparison of the two figures. Corresponding parts have therefore been labelled with corresponding reference numerals. The purpose of FIG. 2 is to enable explanation of one theory of operation of the coupler network of FIG. 1.
Toward understanding the latter, it should be observed that the character of the tapered wave guide sections 8 and 8a is important to its successful operation. For example, the transient behavior of such tapered transmission lines has been found to include a minimum of distortion, providing that the length L of each tapered section is great compared to the quantity or, which quantity is the product of an input transient or impulse duration 1' and the speed of light c, provided also that the change in impedance level from the high impedance end of the tapered section to the other is less than three to one. The parameter or has been established in the field of microwave transient signal research as a useful parameter for identifying the behavior of transmission line elements exposed to short duration electromagnetic signals. It has been reported in the literature as being of interest in the qualitative evaluation of signal distortion in terms of the length of a tapered transmission line. For example, if or is very small compared to the length of a tapered line, only a negligible distortion will be suffered by impulse signals traversing the tapered line.
The parameter or is also of interest in qualitatively defining the character of the junction region 4 where branch lines 6 and 6a join line 5. Here, for transmission of a relatively undistorted transient or impulse signal,
the general area commonly regarded as the junction region needs to be great compared to 01. The region 4 5 then behaves like a simple resistive discontinuity to transients, rather than having distributed dispersive characteristics that cause signal distortion. With or very small by comparison, the dispersion or smearing of a very short pulse passing through region 4 is small com pared to 1'. While the junction 4 itself is not readily defined in purely geometrical terms, it is readily in terms of the electromagnetic fields propagating across the junction. In essence, it has been described as involving an area centered on junction 4 where there are TEM and other propagation modes present. Departure a short distance from the actual junction region discovers the presence only of the TEM mode fields normally associated with propagation in planar microcircuit lines.
Assume, for instance, that a unit volt signal is incident at port A associated with branch 3 of the coupler of FIG. 2. Such a signal will propagate into planar strip line section 9 in substantially the TEM mode. When the unit signal begins to propagate into the tapered line section 8, its peak voltage value must increase in magnitude as energy must be conserved. For example, it can readily be shown that the voltage amplitude at any distance L from the end 7 of tapered section 8 is:
o/( o) (4) For the indicated Z 0 and Z values, F=, O.5. It follows that the voltage V, transmitted from junction 4 toward the output port C and also toward port B associated with transmission line 3a is given by:
V,== V,j2 (6) In a similar manner, it will be seen that a unit voltage signal incident at port B associated with branch 3a of the bilateral coupler device produces a voltage V, transmitted from junction 4 toward the output port C and also toward port A associated with line 3, this voltage V, also being given by equation (6).
Accordingly, if operating generators of coherent signals are placed at ports A and B equidistant from transmission line junction 4, the total voltage V, contribution from both sources A and B at the output port C is predicted from equations l and (6) to be:
V =V V, 7 Thus, the net voltage at output port C has been increased by a factor of V2.
It also follows that under the circumstances cited above, all of the energy input at ports A and B is delivered to output port C; i.e., the coupler network is 100 percent efficient under such operational conditions. In other terms, the transmitted voltage V, due to a generator at port A which appears from equation (6) to be transmitted toward port B is met at junction 4 by an identical but opposite reflected voltage due to the generator at port A. Energy flow toward ports A and B cannot occur, these being forbidden paths when Z 2 Z The network is 100 percent efficient, providing signals from ports A and B arrive at junction 4 simultaneously.
FIG. 3 represents a novel application of the reciprocal energy combining or dual branch network of FIGS. 1 and 2 in a system employing a signal processing tree matrix of such networks for the purpose of providing a sub-nanosecond antenna array having high directivity, narrow effective beam width, and high efficiency. Such antenna arrays effectively focus incoming radiation by use of an array of collector elements and of combining means for coherently adding the energy received from a specific direction in space by each collector element. Energy coming from other directions adds to produce signals of lower amplitude, as will be seen. The antenna array of FIG. 3 performs the desired summation by use of a plurality of the special energy combining or dual branch networks-of FIGS. 1 and 2, since such networks introduce a minimum of time domain distortion and do not attenuate the antenna response.
In the discussion of the apparatus of FIG. 3, reference will be had to detection of targets relative to one plane only, as it is apparent to those skilled in the antenna art that principles and apparatus useful in scanning in space about one axis may equally well be separately or simultaneously applied about a second scan axis. Further simplification of the drawing has been made by showing in FIG. 3 substantially only one half of the energy collector array and tree network, it being understood that similar halves of the antenna array and tree network are formed on each side of the vertical dash-dot line 18 which extends vertically above the tree network output transmission line 500.
FIG. 3 is intended to illustrate an antenna array of p energy collectors a, 10b, 10p, of which collectors 10a and 1011 are shown in the figure as forming half of the total antenna array. Collectors 10a, 10b, 10p
are at any one instant of time connected through switch arrays 11a, 11b, 11p and 12a, 12b, 12p to the respective input leads of the tree network.
The energy combining bilateral tree network comprises a tree matrix with four successive tiers or stages 100, 200, 300, and 400, which tiers combine transient or impulse signals exciting simultaneously from switch array 12a, 12b, 12p additively, being capable of producing high amplitude picosecond pulses on planar output transmission line 500. Each stage or tier of the combining tree matrix network may comprise a row of dual branch energy coupling networks of the type discussed in connection with FIGS. 1 and 2, and it is to be understood that such devices have been illustrated only schematically in FIG. 3 as a matter of convenience. For example, the dual branch coupling network 103, 103a, 203 of FIG. 3 corresponds to the network of FIG. 1 having arms 3, 3a, and 5, respectively.
Similar dual branch network elements are represented in FIG. 3 in tier or stage as respectively having arms 113, 113a, 203a; 123, 12311, 213, 124, 124a, 213a; and so forth. Tier 300 has two such dual branch networks, one of which networks is illustrated and has arms 303,303a, and 403. The final combining tier 400 has a single dual branch network, of which arms 403 and 500 are shown fully and arm 403a in part.
It is seen with reference to the dual branch network 103, 103a, 203 of tier 100 that arms 103 and 103a have 50 ohm impedance inputs both of which are translated to 100 ohm impedance lines adjacent their common junction with 50 ohm impedance line 203. The remaining dual branch networks of the tree are similarly constructed in accord with the principles set forth in the discussion of FIGS. 1 and 2, including the dual arm output combining element employing arms or branches 403, 403a, 500. While no substrate or ground plane respectively corresponding to the dielectric sheet 1 and ground plane 2 of FIG. 1 are illustrated in FIG. 3 merely as a matter of convenience, it is to be understood that stages 100, 200, 300, and 400 may be bonded to such a dielectric substrate, and may be used with a similar ground plane. Furthermore, additional circuit elements such as energy collectors 10a, 10b, 10p and switch arrays 11a, 11b, 11p and 12a, 12b, 12p may be additionally supported by well known techniques in common upon the same dielectric substrate.
As noted previously, the elemental energy collectors 10a, 10b, ,l0p are individually associated with corresponding elements of switch arrays 11a, 11b, 11p and 12a, 12b, 12p, the conductivity of the elements of these switching arrays being cooperatively controlled through well known mechanical or electrical means indicated by dotted lines 16 and 17 by switching synchronizer 505.
The structure of the switch array systems in relation to collectors 10a, 10b, 10p and to the network input terminals of tier 100 of the tree matrix may be appreciated by considering how switching synchronizer 505 is related to the circuits leading to tier 100 from antenna collectors. 10a and 10b. Collector 10a, for example, is connected to the blade of switch 11a. In the position shown in FIG. 3, the current path from collector 10a is through switch 11a and transmission line 15a. The current path to input line 103 of the dual branch network 103, 103a, 203 is completed through the blade of switch 120.
It is observed, at any one instant of time, that switches 11a and 12a are synchronously positioned by the switching synchronizer 505 through cooperatively operated control means 16 and 17 so as to complete one of two paths between collector 10a'and line 103. The first transmission line path 15a is characterized by anominal standard transmission delay, which may be substantially zero, while a second circuit path of different character is formed when the blades of switches 11a and 12a contact the respective terminals of transmission l-ine 16a. Line 16a has a predetennined delay characteristic greater by a fixed amount A than the delay of transmission line 15a. It will be seen that ele ments 11a, 12a, 15a, and 16a cooperate as a dual state delay means. The fixed delay A is thus seen to be' obtained by introducing lengths of non-dispersive electromagnetic delay lines, such as may be provided by high quality coaxial cable or strip transmission line, of electrical length L such that L/c A, where c is the velocity of propagation in the delay line medium. The time delay A characterizing delay devices 16a, 16b, 16c, 16p is equal to the delay A of the yet-to-be discussed delay device 503, but is otherwise arbitrary in value.
Similarly, switches 11b and 12b operate as a dual state delay means and are synchronously positioned by the switching synchronizer 505 through the agency of the respective control means 16 and 17 to complete one of two current paths between collector 10b and input terminal of line 103a of the same dual branch coupler 103, 1311, 203. A first path via transmission line b, like the path through line 15a, is characterized by a particular nominal delay equal to the nominal delay of line 15a. A second circuit path is found when the blades of switches 11b and 12b contact the respective terminals of transmission line 16b. Line 16b has the same delay characteristic of transmission line 16a; i.e., it appears to be A units longer than line 15b or line 15a.
It is seen that similar alternate current paths are selected by the operation of corresponding first and second switches in an array of dual state delay devices, each under synchronous control of switching synchronizer 505, between collector elements 10a, 10b, 10c, 10p and associated network input terminals of the arms of the respective dual branch couplers throughout the entire tier 100. It is seen that collector elements 10a, 10b, 10p have, in the apparatus illustrated in FIG. 3, equal sensitivities and equal spacing so as to form a lineal antenna array; therefore, lines 15a, 15b, 15p and lines 16a, 16b, 16p all have equal nominal delay characteristics, such as substantially zero. Also, the A value for each line 16a, 16b, 16p has a predetermined constant value. It will be clear to those skilled in the art that other combinations are feasible; for example, antenna arrays are known in the art in which the collector elements are deliberately spaced at different intervals. It will be clear that in the latter kinds of antennas, the delay A may vary according to the varying collector element spacing. It will be understood that switch arrays 11a, 11b, 11p and 12a, 12b, 12p may be mechanical switches operated by mechanical linkages l6 and 17. On the other hand, elements 16 and 17 may be electrical current paths for operation under control of switching synchronizer 505 of known types of latching or other ferrite transmission line or semiconductor diode phase shifters or delay devices.
The respective outputs of switch array 12a, 12b, 12p are connected in pairs to the transmission line input arms of the couplers of tier 100. Stage 100 of the electromagnetic energy propagating tree matrix comprises p/2 elemental couplers, each having two symmetric arms (such as the respective arms 103 and 103a, 113 and 1130, and so forth). The combined outputs of the p/2 elemental couplers of stage 100 respectively appear on p/2 output or third port, 500 ohm arms, such as ports 203 and 203a of the first elemental coupler of stage 200.
Stage 200 consists of p/4 elemental couplers, represented in the drawing by the coupler employing symmetric arms 203 and 203a. These arms are seen to vary in impedance from 50 ohms at their input to 100 ohms adjacent theirjunction with the 50 ohm arm 303.
The third stage 300 of the tree matrix consists of p/8 elemental couplers, represented in the drawing by the elemental coupler employing symmetric arms 303, 303a. These arms 303 and 303a are also seen to vary in impedance from 50 ohms at their input to 100 ohms at their junction with the 50 ohm output port 403.
The final stage 400 of the tree matrix consists of p/ l 6 or a single elemental coupler having input arms 403 and 403a. These arms 403 and 403a also vary in impedance from 50 ohms at their input to 100 ohms at their junction with the 50 ohm output port line 500. It is to be understood that the symmetric arms of the successive stages of the tree matrix are of length always greater than m. It will be understood that the effective junction regions respectively attached to the junctions coupling tiers 100, 200, 300, and 400 are all large compared to or.
In the typical combining tree matrix illustrated in FIG. 3, a quadruple tier matrix has been illustrated. It should be understood that other numbers of tiers, with the possibility of consequent multiplication of the output signal on line 500, may readily be employed. The
voltage amplification for increased numbers of tiers is readily calculated from equation (7). Thus, a unit voltage signal into two associated input ports produces 2 volts across a load at an output port. In a double tier system, a unit voltage input produces 2 volts at the output. The triple tier arrangement produces 2 V2 volts. A quadruple tier system with 16 unit voltage generators at 16 inputs yields 4 volts on the single output line 500. Characteristic of each such matrix and independent of the number of tiers or stages in the matrix is the fact that the distances or propagation times from each first tier coupler input terminal to the output of the last stage or tier is always equal.
Understanding of the invention will be aided by consideration of the operation of the FIG. 3 structure so far discussed, but with reference to FIG. 4. When each of the antenna array collector elements is illuminated by a transient or impulse type of carrier wave of electromagnetic energy, that wave will have a front making an arbitrary angle 0 with respect to an imaginary line drawn through collectors 10a, 10b, 10p of FIG. 3. If each such collector is illuminated equally by such a wave front then, for some predetermined value 6,, of the angle of incidence 0, all of the signals passing, for example, through transmission line paths 15a, 15b, 15p will add coherently at the output transmission line or port 500, as in FIG. 4a. These signals will emerge at port 500 to form a short pulse of duration 7 seconds. Coherent signals simultaneously injected, for example, into arms 103, 103a, at points equidistant from line 203 combine so that the total voltage contribution V from arms 103 and 103a goes into arm 203 and no signals are reflected from the junction into arms 103 or 103a. Every other such coupler junction has the same property. Each such junction is one hundred per cent efficient when signals from adjacent paired energy collectors flow simultaneously into corresponding arms such as arms 103, 103a. It is seen that the total tree matrix has the same property, being made up of like dual branch couplers; thus signals arriving at angle 0,, are added coherently on transmission line 500, as in FIG. 4a.
For other values 0, of the angle of incidence 0, the individual signals received by the respective corresponding collectors 10a, 10b, 10p may arrive at port 500 separated in time, as in FIG. 4b. The larger the value of p (the total number of collector elements 10a, 10b, 10p), the greater the ratio between the maximum signal obtained at port 500 for the peak axis at angle 0,, of an antenna receptivity pattern of finite width and the offaxis contributions to the array of energy collectors. When the angle of incidence 0,, deviates only slightly from the antenna pattern peak sensitivity direction the peak amplitude of the combined signal seen at port 500 drops and the signal energy spreads in the time domain as seen in FIG. 4c. FIGS. 40, 4b, and 40 may be further understood by observing that the antenna pattern width in space may be defined in terms of them. For example, when the amplitude of the signal at port 500 drops to half of its peak value at 0 the antenna receptivity pattern width may be defined or. measured in terms of the time interval at the A voltage points of the curve of FIG. 4c.
It will be understood that the angle 0, of the peak of the antenna pattern may be steered by mechanical steering of the antenna of FIG. 3. Alternatively, electronic steering of the receptivity pattern and consequently of 0, may be achieved in the usual manner by adding electronically variable delay devices or phase shifters for the purpose in each of the lines leading from each of the antenna collector elements a, 10b, 10p.
Signals output at port 500 of the tree matrix final stage 400 are supplied through a switching mechanism including switches 501, 504 to utilization apparatus 510. The blades of switches 501 and 504 are caused to operate synchronously with the respective corresponding blades of switch arrays 11a, 11b, 11p and 12a, 12b, 12p. Synchronous operation is again brought about by mechanical or electrical control means 506, 5061: under operation of control 506a by the switching synchronizer 505. Port 500 is alternately connected through a transmission line 503 having a delay A greater than line 502, and then through line 502. Elements 501, 502, 503, and 504 together form an output dual state delay means. In what will be called the mode A situation, the delay A is removed from the transmission lines leading signals from collectors 10a, 10b,
10p to the respective inputs of tier 100 of the tree matrix. However, in the mode A situation, the delay A is inserted by switches 501 and 504 between port 500 and utilization apparatus 510.
Conversely, in what may be called mode B operation, a delay A is inserted in each of the transmission lines leading from collectors 10a, 10b, 10p to the respective input terminals of tier 100. In mode B operation, the delay A is removed by switches 501 and 504 between port 500 and utilization apparatus 510.
As noted previously, when the angle of incidence 0 is 0 all signals arrive from collectors 10a, 10b, 10p simultaneously and coherently add at port 500. For any other angle of incidence 0 the response at port 500 in the mode A situation is a train of pulses as in FIG. 4b. Such a statement is fully accurate, however, only when the driving point impedances of the antenna collector elements 10a, 10b, 10p are individually substantially equal to the impedances of the respective associated transmission lines 15a, 15b, 15p over a sufficiently extended frequency domain. Such a perfect impedance match is difficult and costly to attain. However, the consequence of relaxing matching tolerances is that a portion of the energy in impulses passing each dual branch coupler junction is reflected back toward the collector elements 10a, 10b, 10p. Similarly, reflected waves entering a given junction region from below continue to see no discontinuity until an antenna collector element is reached. At the antenna collector locations, both types of waves are rereflected, causing the responses to spread and generating a very large number of pulses at various time locations in the output port 500.
To obviate this difficulty, the previously described apparatus is used in a manner making use of the inherent properties of the signals involved so that the effects of undesired signals are substantially eliminated. For this purpose, the apparatus is set to operate in mode B, delay elements 16a, 16b, 16p of A seconds retardation being injected in series in the current paths from each respective antenna collector element 10a, 10b, 10p. The presence of delay elements 16a, 16b, 16p delays each signal flowing from each collector 10a, 10b, 10p toward port 500 by A seconds. However, signals re-reflected by antenna collectors 10a, 10b, 10p after internal reflection in any tier of the matrix network are delayed by 2A, 4A, 6A, or sA sections, where s is an even integer.
Referring to FIG. 5, FIG. 5a illustrates an idealized pattern which might appear on a conventional cathode ray radar or other oscilloscope or which might otherwise represent the time sequence of the input to utilization device 510 with the apparatus receiving an impulse from space while operating in mode A. A signal 600A of K volts amplitude and A time delay corresponding to a signal received by the antenna at time zero is accompanied by a false representation pulse 601A caused by undesired reflections within the system of FIG. 3.
FIG. 5b represents a related idealized time pattern showing the input to utilization apparatus 510 with the apparatus receiving an impulse from space while operating in the mode B condition. A desired signal 6008 again of K volts amplitude and delay A seconds again corresponding to a signal received from space by the antenna at time zero is present. Also present is -a false representation 6013 caused also by undesired reflections within the system of FIG. 3, but shifted in time so as to appear A seconds laterthan the spurious mode A pulse 601A of FIG. 5a.
Multiplication of the signals shown in FIGS. 5a and 5b, using the same zero time reference, yields the enhanced signal 602 of FIG. 50 having an amplitude of K volts and the same A seconds time delay with reference to zero time. Spurious signals 601A and 6018 are greatly reduced in magnitude and may approach zero level in FIG. 5c if the background noise level is sufficiently low.
In essence, the invention provides means for receiving impulse signals from space by an array of cooperating sensors, such as have commonly been exploited in radar or in direction finding, and for processing and combining such signals by means permitting significant enhancement of received impulse signals genuinely representing useful data and also providing significant degradation of spurious responses. A situation met with typically in radar, direction finding, and the like systems involves undesirable responses of an antenna array and signal combining network for a signal arriving off the boresight axis 6 that is, in a direction other than the direction of the axis of symmetry of the antenna receptivity pattern. The response typically consists of a primary response plus various anomalous responses which arise internally of the signal combining system mainly because of mismatches at network junctions in the combining system.
It will be apparent from the foregoing that such spurious responses are substantially, if not entirely, eliminated by the mechanism of the present invention. To accomplish the desired effect, the invention is successively and alternately operated in two particular modes. Switches are operated in a mode A in such a way as to introduce a delay A in the output of the combining network. Switches are again operated, translating the system into mode B operation. Here, the delay A is removed from the output of the combining network and is injected instead in series with each antenna collector element. The time of arrival of the primary response impulse at utilization device 510 is unchanged in shifting cyclically from mode A to mode B operation. However, anomalous returns are delayed by A seconds in the B mode with respect to their timing in mode A. It is seen that simple multiplication of pairs of successive inputs to utilization device 510 yields the desired effect of enhancement of desired signals and degradation of undesired signals.
It is obvious to one skilled in the art that multiplication of successive A and B mode signals is readily achieved by well known means. A variety of known approaches may be exploited and it is therefore considered to be unnecessary to describe here examples of such mechanizations in any detail.
However, it should be clear that the utilization apparatus 510 of FIG. 3 may comprise simply two type A indicator storage tubes. For example, one or more mode A samples looking like FIG. 5a may be stored on a first cathode ray storage tube screen. The apparatus then may be switched to mode B operation and signals like that of FIG. 5b stored on the second storage tube screen. Multiplication may be accomplished mentally, or after suitable measurements are taken, and a graph like FIG. 5c constructed.
Simple analog means are well known for the substantially instant multiplication of pulse signal amplitudes. Such make it readily possible, for example, to operate the invention in the A and B modes, delaying the A mode response by a conventional delay line-amplifier circuit by the fixed frame time of operation on the A mode, and then applying the delayed A mode response and the delayed B mode response in proper synchronism to the inputs of a multiplier whose output is passed for observation on a type A radar screen. Cyclic operation of such a system is evidently feasible.
More sophisticated mechanizations are readily possible using apparatus well known in the prior art. By sequentially operating the antenna array and processing system in modes A and B, and by digitally storing successive respective output wave forms, quantized versions of the received signals for the two modes may readily be stored in individual locations in a general purpose digital processor memory. For example, digital versions of the mode A and mode B wave forms may be cyclically stored in individual memory register circuits. The data is synchronously shifted out of the registers at the appropriate instant of time and an ordinary digital multiplication follows. If the computed result is desired in analog form, it is evident that appropriate conventional digital-to-analog conversion may be employed to construct an image such as that of FIG. 50 for presentation on the screen of a cathode ray tube.
FIG. 6 represents, by way of example, a general method and apparatus for performing the functions of the utilization apparatus 510 of FIG. 3 by employment of digital processing techniques. In FIG. 6, it is shown that the invention may be practiced by the cooperative employment of certain circuits often present in conventional sampling high frequency oscilloscopes and of certain of the circuit elements conventionally found in simple general purpose digital computers.
Referring particularly to the part 700 of the FIG. 6 apparatus for performing the sampling oscilloscope functions, it will be seen that apparatus 700 lends itself to sampling wide band width signals by use of a stroboscopic method by means of which an input wave form may be reconstructed from samples repetitively taken during a number of successive recurrences of an input wave form. An input wave having a given repetitive frequency is continuously sampled at a slightly lesser repetition frequency, and the sampled voltages are stored. In the instance that the sampling system is to provide a display, the sampled wave form voltage is measured and is used to position the electron beam of a cathode ray tube. The sample taken during the next succeeding sampling event is derived at a slightly later point on the input wave form. The cathode ray is moved incrementally in a horizontal sense, but is repositioned vertically according to the magnitude of the newly sampled voyage. Accordingly, there is plotted on the face of the'storage cathode ray display, in point by point fashion, the reconstructed wave form by using as many, for example, as successive sampics to reconstruct the input wave form. In the present invention, the sampled voltages are supplied by the sampling system 700 for processing in a conventional digital computer system 721.
Referring particularly to the sampling system 700 of FIG. 6, the wave form from switch 504 of FIG. 3, which may, in mode A operation, for example, consist of pulses corresponding to the idealized pulses 600A and 601A of FIG. 5a, is applied as an input to the sampling system 700. As is suggested above, no actual display of the signals in sampling apparatus 700 is required for use in the present invention, although such may be convenient.
The sampling system 700 includes a sampling gate 701 and a sample and hold circuit 703 which together comprise what is known in the art as a sampling head 704. Sampling apparatus 700 also includes a time base generator 702 and a sample amplifier 705, and may include an oscilloscope display 706. The input wave form to sampling head 704 comprises, in the system A mode, for example, signal wave 600A and 601A, supplied directly to sampling gate 701 which is enabled by a signal provided by the time base generator 702. The
output from sampling gate 701 is coupled to the input of sample and hold circuit 703, which circuit is also triggered by a signal supplied by time base generator 702. The sample and hold circuit 703 supplies signals corresponding to input signal 600A and 601A, when the apparatus is in the A mode, to the sample amplifier 705. The latter amplifier 705 provides output signals for application after conversion to analog-to-digital converter 720 directly to digital processor 721, as will be explained.
Should it be desired that a presentation of the output of sample amplifier 705 be provided, the latter also provides a signal to the vertical deflection circuits of oscilloscope display 706. The horizontal sweep signal applied to the horizontal deflection circuit of display indicator 708 is supplied via lead 7020. Time base generator 702, which supplies the horizontal sweep signal via lead 702a, also provides unblanking signals to a conventional electron beam intensifier circuit associated with display indicator 708. Erasure signals are supplied in the usual manner.
Time base generator 702 is a conventional circuit for application in sampling oscilloscopes and elsewhere. It generates a sampling wave form illustrated by the timing wave form A of FIG. 7, which is held, for instance, in synchronism with the repetition cycle of the input signal 600A by means of synchronizer signals from terminal 505A being applied to generator 702. The sampling wave form A of FIG. 7 provides a time base synchronized with respect to the input signal comprising elements 600A and 601A, and is used in a conventional manner in time base generator 702 to provide triggering wave forms for sample gate 701 and for the sample and hold circuit 703. I
As previously indicated, time base generator 702 generates the sawtooth sweep wave that is shown at A in FIG. 7. These repetitive signals are synchronized in a conventional manner to the cycles of the repetitive input wave from switch 504 by means of the synchronizer input to generator 702. The wave A therefore provides a time base voltage for the system 700 with respect to input waves such as wave 700A and 601A when the system is operating in the A mode. Time base generator 702 also provides in a conventional manner a slowly increasing ramp voltage B which includes many cycles of wave A. The slowly increasing ramp voltage B is combined in a conventional manner within time base generator 702 with wave A for the purpose of generating for use in sampling head 704 a trigger pulse whenever wave A intersects the slowly increasing ramp B level. The trigger pulses thus generated are used within the sampling head 704 to trigger sampling gate 701 and sample and hold circuit 703. Accordingly, upon each occurrence of the trigger pulse, the sampling gate 701 transmits a voltage sample of the input wave from switch 504 to the sample and hold circuit 703, wherein the sample is stored until the next occurring sample is obtained. Wave form A, being synchronized to the input wave from switch 504, cooperates with the slowly increasing ramp voltage B so that the samples obtained by sample gate 701 slowly pro'gress across the repetitive input wave form. That is, a quantized output wave 705A is generated, which is a quantized version of'the received wave obtained from switch terminal 504, for application to analog-to-digital converter 720. A corresponding wave 7053 may be formed by processing waves 6003 and 601B. If desired, the wave forms 705A or 7058 may be presented on the screen of the cathode ray tube 708 of display 706. For this purpose, the samples held in sample and hold circuit 703 may also be supplied via sample amplifier 705 to the vertical deflection circuit of indicator 706. The slowly increasing ramp B voltages generated within time base generator 702 are used as horizontal deflection signals in display indicator 708. A version of the trigger pulses provided by time base generator 702 to enable sampling head 704 is applied to a control grid of the cathode ray tube 708 for providing unblanking signals for the electron beam. Thus, successively occurring samples obtained across the input wave such as waves 600A, 601A or 6003, 6013 may be displayed on the screen of cathode ray tube 708.
The immediately foregoing paragraphs describe the structure and operation of a conventional system for sampling repetitive wave forms and for reconstructing quantized versions of those signals for digital processing or for other purposes such as display on a cathode ray tube oscilloscope. Such systems are known in the art, though improved sampling systems may be preferred, such as that disclosed by Alexander M. Nicolson in the U.S. Pat. application, Ser. No. 844,021, filed July 23, 1969, entitled: Method and Means for Compensating Amplitude and Time Drifts in Sampled Waveform Systems," issued June 8, 1971 as U.S. Pat. No. 3,584,309 and also assigned to the Sperry Rand Corporation.
For the purpose of processing the successive wave forms 705A and 7058 derived within the sampling apparatus 700, elements of a conventional general purpose digital computer 721 along with digital converters 720 and 725 may be employed. As seen in FIG. 6, computer 721 may contain a programmer memory 722 under control of signals from terminal 505A generated within the switching synchronizer 505 of FIG. 3. It will be apparent to those skilled in the art that computer 721 may be synchronized by signals derived in switch synchronizer 505. Alternatively, it may be preferred that operation of the entire system be synchronized in the conventional manner from the programmer memory 722 located withincornputer72l. In any event, programmer memory 722 is a device having suitable storage elements for storing in a manner well known to those skilled in the art a program for programming further operations upon signals 705A and 7058.
Digital computer 721 additionally contains a second storage means which may take the form of first and second respective registers 723a and 723b, also under the control of timing signals from programmer memory 7 22. Outputs from analog-to-digital converter 720, which receives signals 705A and 7058, are additionally respectively applied to registers 723a and 723b. A further element of the digital computer 721 is the digital multiplier 724. Multiplier 724 accepts the output of registers 723a and 723b when the contents of these registers are shifted upon the occurrence of a command signal from programmer memory 722. The product output of multiplier 724 is then supplied by computer 7 21, still in digital .word form, to the digitalto-analog converter 725. The latter yields, after multiplication within the computer 721 of signals 705A and 7058, a signal devoid of reflections, as ideally illustrated in FIG. 5c. The output signal 750 of converter 725 is supplied, under the control of programmer memory 722, to utilization apparatus 726. Utilization device 726 may comprise, for example, a cathode ray indicator 727.
In operation, it is seen that a series of successive applications of the FIG. 5a signals 600A and 601A to apparatus 700, followed by a substantially equal number of applications of signals 6008 and 6013 of FIG. 5b, yields sets of signals such as signal 705A and 7058 for conversion into digital words by converter 720 and for storage in registers 723a and 723b. When the appropriate number of storage locations in register 723a is filled, programmer memory 722 causes converter 720, by an elementary switching process, to fill register 723b with digital words corresponding to signals 6008 and 6018 of FIG. 5b. Converter 720 and registers 723a and 723b perform their functions in a conventional manner and may be controlled in a conventional manner by programmer memory 722, as may be multiplier 724, converter 725, and display 726.
When register 723a is filled with digital binary words, for example, corresponding to mode A signals 600A and 601A, and when register 723k if filled with data corresponding to the mode B signals 6003 and 6018, the programmer memory 722 then, in a conventional manner, causes the data in registers 723a and 723b to shift into multiplier 724. In the latter arithmetic circuit, the stored digital words are multiplied in the conventional manner to yield binary digital signals representing the product of signals received in the mode A and in the mode B conditions of the system. The output of multiplier 724, being a digital signal, may be supplied to the input of a digital processing system such as a digital servo mechanism or to other apparatus accepting digital input signals. As shown in FIG. 6, the digital product may be converted into an analog signal by converter 725 for display as signal 750 devoid of all undesired reflections or for use in other analog utilization apparatus.
While the invention has been described in its preferred embodiment, it is to be understood that the words that have been used are words of description rather than limitation and that changes within the purview of the appended claims may be made without departing from the true scope and spirit of the invention in its broader aspects.
1. An electromagnetic signal processing system comprising:
electromagnetic energy signal propagating,
branching tree network means,
network input terminal means,
network output terminal means,
first dual state delay means adapted to receive electro-magnetic signals, coupled to said network input terminal means,
second dual state delay means coupled to said network output terminal means,
synchronizer means for switching said first and second delay means between first and second states while maintaining the state of said first delay means opposite to the state of said second delay m n a a utilization means for receiving the output of said second delay means.
2. Apparatus as described in claim 1 wherein said input terminal means comprises plural terminals.
3. Apparatus as described in claim 2 wherein said output terminal means comprises a single terminal.
4. Apparatus as described in claim 3 wherein said network input and output terminal means and said branching tree network means are so constructed and arranged that the energy propagation path from any one of said plural terminals to said single terminal has the same length as the energy propagation path from any other of said plural terminals to said single terminal.
5. Apparatus as described in claim 4 wherein said utilization means comprises cathode ray storage indicator means for individual presentation of signals traversing said signal processing system in said first and second states for visual simultaneous comparison thereof on a time scale basis.
6. Apparatus as described in claim 4 wherein said utilization means comprises:
signal multiplier means having first and second inputs and outputs,
fixed delay means,
means operable in said first state for connecting the output of said signal processing means through said fixed delay means to said first multipIier input, means operable in said second stable state for connecting the output of said signal processing means directly to said second multiplier input, and indicator means for displaying the output of said signal multiplier.
* III R II! t
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|US5084706 *||Dec 15, 1989||Jan 28, 1992||Ross Gerald F||Synchronization of very short pulse microwave signals for array applications|
|US5244869 *||Oct 23, 1990||Sep 14, 1993||Westinghouse Electric Corp.||Superconducting microwave frequency selective filter system|
|US7724189 *||Nov 24, 2004||May 25, 2010||Agilent Technologies, Inc.||Broadband binary phased antenna|
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|US20110113731 *||Dec 9, 2010||May 19, 2011||Graham Packaging Company, L.P.||Repositionable Base Structure for a Container|
|U.S. Classification||342/373, 333/128, 342/374|
|International Classification||H01Q3/30, H01Q3/40|