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Publication numberUS3748605 A
Publication typeGrant
Publication dateJul 24, 1973
Filing dateNov 4, 1971
Priority dateNov 5, 1970
Publication numberUS 3748605 A, US 3748605A, US-A-3748605, US3748605 A, US3748605A
InventorsBaynham A, Dunsmore M
Original AssigneeNat Res Dev
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Tunable microwave filters
US 3748605 A
Abstract
A microwave filter includes one or more bodies of magnetic material placed in a variable magnetic field. Right circularly polarised radiation falls on the body or bodies in a direction parallel to the direction of the magnetic field via a partial-reflection-inducing discontinuity such as a microwave iris. The body or bodies is or are preferably a disc or a number of discs held perpendicular to the direction of the magnetic field. Varying the magnetic field varies the frequency of the filter passband.
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Description  (OCR text may contain errors)

United States Patent 1 1 Baynham et a1.

14 1 July 24,1973

[ TUNABLE MICROWAVE FILTERS [75] Inventors: Alexander Christopher Baynham;

Michael Robert Buchanan Dunsmore,-both of Malvern, England [7 3] Assignee: National Research Development Corporation, London, England 22 Filed: Nov. 4, 1971 21 1 Appl. No.: 195,737

[30] Foreign Application Priority Data Nov. 5, 1970 Great Britain 52,79 6/70 52 us. or. 333/73 x, 333/21 A, 333/73 w, 333/84 R, 333/84 M 51 rm. c1. H0lp 1/20 [58] Field of Search 333/21 A, 24 G, 24.1, 333/73 R, 73 c, 73 s, 73 w [56] References Cited UNITED STATES PATENTS 3,001,154 9/1961 Reggia 333/73 W 7/1953 Luhrs et al 333/24 G UX 11/1966 Weiss 333/24.1

Primary Examiner-Paul L. Gensler Attorney-Cushman, Darby & Cushman [5 7 ABSTRACT 40 Claims, 14 Drawing Figures I 10 j 9 l2 PATENTEDJULZMQB 3.748.605

sum 2 or 6 i PAIENIEDJULMIQH 3, 748,505

saw u or e FIG. 4a

FIG. Sb

. l TUNABLE MICROWAVE FILTERS The present invention relates to tunable microwave filters.

Magnetically tunable, high-Q microwave filters using narrow linewidth ferrimagnetic single crystals (particularly, highly polished spheres of yttrium iron garnet referred to hereinafter as YIG) as resonating elements providing low loss coupling between two microwave circuits with orthogonal RF magnetic fields are by now familiar devices to the microwave engineer. Indeed, they appear to have become virtually indispensable in such applications as panoramic receivers, i.e. tuned radio frequency microwave receivers.

The resonance filter is not, of course, the only tunable filter technique which has been investigated. Previous attempts to solve the problem have included the use of cavities containing ferroelectrics, varactors (variable capacitance reverse biased diodes) and ferrites, as well as ferrite-loaded evanescent waveguides. The device potential of these approaches has been limited since, in general, they can provide tuning ranges of only a few per cent while still maintaining a high unloaded resonator Q factor.

As the resonance filters rely for their operation upon the coupling of microwave energy through a magnetically biassed ferrimagnetic resonator at the frequency corresponding to the execution of its uniform (spin) precession resonance mode, the achievement of low insertion loss and narrow bandwidth requires that the ferrimagnetic resonator itself should have a high unloaded Q factor. However, the absorption losses associated with the ferrimagnet reach their maximum at the uniform precession resonance, so that the use of very narrow linewidth (low loss) materials is dictated, with consequent low power handling capabilities (typically lmW maximum before limiting). Apart from problems of power handling, the resonance filters have two other principal disadvantages. The first principal disadvantage is that the filter develops spurious responses, due in the main to the excitation of the (210) and (540) magnetostatic modes of the ferrimagnetic resonator (mode designation due to L. R. Walker see the article Magnetostatic Modes in Ferromagnetic Resonance, Physical Review, Vol. 105, No. 2, January 1957, pp 390-399). The second principal disadvantage is that the configuration of the microwave circuit in which the resonator is mounted results in the filter refleeting virtually all incident power at frequencies outside the passband.

It is an object of the invention to produce a tunable microwave filter relying on the effects produced by multiple reflections of electromagnetic waves in magnetic layers sandwiched between waveguide irises, or other microwave discontinuities which are capable of producing large reflection coefficients for electromagnetic radiation.

According to the present invention there is provided a microwave filter including at least one body of magnetic material having plane parallel faces with the dimension perpendicular to these faces being small compared with the dimensions parallel to the faces, means for feeding the microwave radiation to be filtered into the body in the form of right circularly polarised electromagnetic waves in a direction having a substantial component perpendicular to the plane parallel sides, a partial-reflection-inducing discontinuity to the transmission of the right circularly polarised electromagnetic waves at the plane parallel surfaces of the body, means for applying a magnetic field in a direction having a substantial component perpendicular to the plane parallel sides and means for varying the magnetic field.

The magnetic material may be ferrimagnetic, ferromagnetic or antiferromagnetic material, or magnetic semiconductor material.

The means for feeding the microwave radiation to be filtered may include microwave transmission media, for example, waveguide or strip transmission line as hereindefined. The term strip transmission line is used to cover all electromagnetic wave transmission systems in which the radiation is bounded or guided by parallel or coplanar metal films or layers deposited or mounted on or between any dielectric or other media. Examples of strip transmission lines are stripline, microstrip, slot line and coplanar waveguide.

Embodiments of the invention are described below with reference to the accompanying drawings, in which:

FIG. 1 is a dispersion diagram for electromagnetic wave propagation in polycrystalline YIG;

FIG. 2 is a simplified illustration of the basic design for a single ferrimagnetic layer interference filter embodying the invention;

FIG. 3 is another dispersion diagram for electromagnetic wave propagation in polycrystalline YIG at various different magnetic field strengths; v

FIG. 4a is a plan view in cross-section of a slotline;

FIG. 4b is a perspective view of the slotline of FIG. 4a;

FIG. 5a is a plan view of the cross-section of a coplanar wave guide;

FIG. 5b is a perspective view of the coplanar waveguide of FIG. 5a;

FIG. 6 and FIG. 7 are respectively longitudinal and transverse cross-sectional diagrams of a tunable microwave filter mounted in a rectangular waveguide;

FIG. 8 is a longitudinal cross-sectional diagram of a tunable microwave filter mounted in a circular waveguide;

FIG. 9a is a plan view of a tunable microwave filter mounted in a slotline;

FIG. 9b is a cross-sectional diagram of the tunable microwave filter of FIG. 9a;

FIG. 10a is a plan view of a tunable microwave filter mounted in a coplanar wave guide;

' FIG. 10b is a longitudinal cross-sectional diagram of the tunable microwave filter of FIG. 10a;

The principle of the invention may be described by reference to FIG. I which is the dispersion diagram for electromagnetic wave propagation in polycrystalline YIG, parallel to an ambient biasing magnetic field of 1.5 X IOA m", and to FIG. 2 which is a simplified illustration of the basic design for a single ferrimagnetic layer interference filter embodying the invention.

The diagram of FIG. 1 applies in the region where the wave numbers are sufficiently low for the effects due to the exchange interaction between neighbouring spins to be neglected, and assumes the ferrimagnetic medium to be unbounded (i.e. of infinite extent). In the design of microwave devices involving the .use of small samples of magnetic materials inserted within or adjacent to microwave transmission media such as waveguide or strip transmission lines (as defined above) it is at present difficult or impossible to solve the propagation characteristics exactly in many cases.

Nevertheless, the fundamental propagation properties can be illustrated by considering the' propagation of plane waves in an infinite medium, since it is a wellknown microwave technique to develop solutions for modes of propagation in bounded transmission media by taking combinations of uniform plane waves which satisfy the boundary conditions. Such boundary conditions typically involve the matching of the electromagnetic field components which can exist on either side of the boundary.

For the purpose of FIG. 1 the wave number is defined as (21rf/c) m where f is the frequency of the wave in the material, c is the velocity of light in vacuo, and e, and L, are respectively the relative permittivity and effective relative permeability of the material, both of which will normally be complex quantities. In a magnetic material, the value of the complex quantity ;1., depends upon the frequency f, the biasing magnetic field intensity H, and the linewidth AH and spontaneous magnetisation M of thev material. In the dispersion diagram the real part k, of the wave number, which determines the propagation characteristics of the wave, is shown with a continuous line and the imaginary part k, of the wave number, which determines the absorption losses in the medium is shown with a broken line. A straight, chain-dotted, line D through the origin of the graph indicates the real part of the wave number for a pure dielectric having the same permittivity as the ferrimagnetic material. (The mode of propagation represented by this line on the dispersion diagram is commonly known as the ordinary wave, whereas the frequency and magnetic field dependent mode is known as the extraordinary wave.)

The real part k, of the wave number increases from zero at zero frequency initially at the same rate as the lineD but increasing in rate to a sharp maximum at a frequency f known as the uniform precession resonance frequency. The imaginary part k of the wave number is also zero at zero frequency and remains small as the-frequency increases, but towards'the uniform precession resonance frequency f. it increases rapidly and also has a sharp maximum at the uniform precession resonance frequency f,,,,,. In other words from zero frequency until the uniform precession resonance frequency f,,,,. is approached the real part k, of the wave number is always greater than the imaginary part In of the wave number.

The uniform precession resonance frequency f,,,, is given by the formula fective permeability in the material is negative. In this region both the real part k, and the imaginary part k, of the wave number decrease rapidly as the frequency is increased from f,,,,, but k, is always greater than k,. The

cut-off frequency f is given by f, 7(Hi M) where M is the spontaneous magnetisation of the ferrimagnetic material, as stated above. On the case of a completely lossless material, between the uniform precession resonance frequency f,,,,, and the cut-off frequency f the real part k, of the wave number would be zero and the electromagnetic wave in the material would become entirely evanescent. As the frequency rises above the cut-off frequency f the real part k of the wave number starts to increase again to become asymptotic to the line D. The imaginary part k, of the wave number continues to fall, becoming asymptotic to the frequency axis.

Prior art resonance filters rely for their operation upon the coupling of microwave energy through a magnetically biased ferrimagnetic resonator at the frequency f, corresponding to the excitationof its electron spin dipoles into their uniform precision resonance mode. This is the point at which the absorption losses in the ferrimagnet attain their maximum value. It follows that for the ferrimagnetic resonator to have the high unloaded Q factor needed for the achievement of a filter with low insertion loss and narrow bandwidth, the use of low loss (narrow resonance linewidth) material is dictated. As a consequence of having a low loss material operating at resonance, only low power levels (typically IOdBm, i.e. IOmW) can be handled without the spin precession angle building up to the point at which it limits due to a catastrophic transfer of energy to degenerate, exchange interaction dependent spin waves. This causes a decline of the effective r.f. permeability in the resonator and the input and output circuits become progressively more decoupled with increasing power level.

In contrast, the types of filter herein described avoid the high resonance absorption losses by operating in a region of the dispersion diagram of the ferrimagnetic material which is well below the uniform precession resonance frequency f,,,,,., and instead rely upon interference effects to define the frequencies of their passbands. FIG. 2 is a simplified illustration of the basic design for such an interference filter, using, in this example, a single layer of ferrimagnetic material. The filter consists of a plane, parallel layer 1 of ferrimagnetic material, with microwave structures 2, 3, on or adjacent to its plane parallel faces. These structures 2, 3 are capable of producing partial reflection of any right circularly polarised electromagnetic waves incident upon them, both when the right circularly polarised electromagnetic wave is propagating from within the ferrimagnetic layer towards the outside and vice versa. The said microwave structures may be waveguide irises (or apertures). Any incident linearly polarised electromagnetic wave 4 propagating towards the filter is converted to a right circularly polarised electromagnetic wave 5 by a suitable transducer 6. Any right circularly polarised electromagnetic wave 7 reflected from the filter is converted by the same transducer 5 back'to a linearly polarised electromagnetic wave 8. Any right circularly polarised electromagnetic wave transmitted through the filter is converted by a further transducer 10 to a linearly polarised electromagnetic wave 11. The microwave structure 2 and the transducer 6 may be the one component e.g. an iris inserted into the appropriate position in a length of waveguide through which electromagnetic energy is propagating. Similarly, the microwave structure 3 and the transducer 10 may be a single component. In the preferred mode of operation of the device, it is mounted in a microwave circuit in such a manner that the incident linearly polarised electromagnetic wave 4 and the reflected linearly polarised electromagnetic wave 8 propagate along separate microwave transmission lines, and any linearly polarised electromagnetic wave 12 incident upon the output side of the filter cannot readily propagate through it. The construction of the device, taking, as it does, the form of a ferrimagnetic layer 1 sandwiched between two microwave partially reflecting structures 2 and 3, is similar to that of the well-known optical Fabry-Perot etalon, and in a similar way there are fixed (real part) wavenumbers at which passbands can occur. The values of these wavenumbers, k,., are defined by k,d z mr where d is the thickness of the layer and n is an integer. (Any small phase changes on reflection of the electromagnetic wave from the faces of the ferrimagnetic layer arising due to the imaginary part of the permeability of the ferrimagnetic layer 1 and the susceptances of the structures 2, 3 have here been neglected, while the question of any spurious effects which might arise due to any form of wave propagation other than that of a plane electromagnetic wave propagating in a direction perpendicular to the plane, parallel faces of the ferrimagnetic layer 1 are considered below). Since the dispersion diagram is magnetic field dependent, through the dependence of the real and imaginary parts of the wave number on the effective relative permeability 1., of the ferrimagnetic material, any variation in the biasing magnetic field must tune the filter. This effect is illustrated in FIG. 3, which is the dispersion diagram for electromagnetic wave propagation in polycrystalline YIG of infinite transverse dimensions, showing the effect of varying magnetic field on the real part k of the wave number, and hence, on the interference filters passband frequency. The positions of four passband frequencies are indicated for a typical wave number defined by the filter dimensions (viz. the thickness of the ferrimagnetic layer). Other passband frequencies defined by small integer multiples of the wave number shown in FIG. 3 are possible in principle, but are unlikely to appear due to their being situated in each case close to the appropriate uniform precession resonance frequency and being therefore, heavily attenuated as a consequence of the large imaginary parts of their wave numbers.

With the low values of the imaginary part of the wave number in the ferrimagnetic material inherent in the below resonance mode of operation, quite high values of unloaded-Q can still be obtained even when using broad linewidth polycrystalline ferrites for very high power handling capabilities and/or economy. Alternatively narrow linewidth single crystal YIG in the form of discs may be used to obtain much higher unloaded- Q's than with the corresponding uniform precision resonance filters, while still maintaining an appreciable power handling capability.

Alternatively the filter may be operated in the region of the dispersion diagram above the cut-off frequency f}, where again the low value of the imaginary parts k, of the wave number in the ferrimagnetic material allows quite high values of unloaded Q to be obtained.

The foregoing description applies basically to filter structures in which the dimensions parallel to the plane, parallel faces of the layer are of infinite extent, or at least very much longer than the wavelength in the in the infinite medium it is necessary, firstly, to ensure uniformity of the wave numbers (and hence uniformity of the magnetic field) within the region of the ferrimag' netic layer in which the electromagnetic radiation exists and, secondly, to prevent the development of spurious responses due to the reflection of electromagnetic radiation from boundaries of the layer transverse to its plane parallel faces.

The presence of magnetic poles at the surface of a magnetised piece of magnetic material, and the resul tant non-zero surface divergence of the magnetisation vector (i.e. Z-M I=0), gives rise to a magnetic field which counteracts the applied field. In uniformly magnetised samples of ellipsoidal shape this so-called demagnetising field is uniform; in other geometrical shapes, such as the present layer, this is not the case. It can readily be seen from symmetry considerations, that at least azimuthal uniformity of the internal magnetic field may be achieved by using a circular cylindrical disc of magnetic material, and this is, therefore, the preferred shape for the magnetic layer. The problem of evaluating the demagnetising fields within such a body has already been considered in the literature (see the article Demagnetising Field in Nonellipsoidal Bodies by K. 1. Joseph and E. Schlomann, published in the Journal of Applied Physics, Volume 36, No. 5, May 1965) and it is thus possible to determine the ratio of disc thickness to disc diameter (commonly referred to as the disc's aspect ratio) required for any desired degree of internal magnetic field uniformity.

As has already been mentioned, ferrimagnetic spheres can exhibit dimensional resonances, normally called magnetostatic modes, when the wavelength in the medium is comparable with the sample dimensions. If the sample size is such that exchange and propagation effects are negligible, then the residual efiects are purely magnetostatic, arising only from the applied magnetic field and the dipolar field of the sample magnetisation. The essential characteristics of magnetostatic modes are that (a) their resonant frequencies depend on the shape, but not on the size, of the resonator (b) the magnetic field satisfies the magnetostatic equations 2 X I! 0 and Z5 0 and (c) the electric field is negligible; these conditions can apply even when the dimensions of the resonator are large compared with a free-space wavelength, provided the mode numbers are sufficiently large. These modes, whose resonance characteristics have been discussed in the literature (e.g. Walker, op. cit.) are responsible for spurious responses in the case of the uniform precession resonance filter. The filter herein described should be at least relatively immune to such spurious responses since, as is shown in the literature (R. W. Damon 8!. H. van de Vaart Propagation of magnetostatic spin waves at microwave frequencies in a normally magnetised disk" in Journal of Applied Physics, Volume 36, No. 11, November 1965) magnetostatic modes only occur in normally magnetised discs for frequencies between f,,,,, and a frequency given by -y V H,(H, M), which is less than f However, in the present filter design, the size of the resonator will be such that for some modes, the magnetostatic approximation is invalid. The modes then become electromagnetic, rather than magnetic, in nature, having electric fields of appreciable amplitude and resonant frequencies which are size dependent. In contrast to the situation of plane electromagnetic wave propagation which exists in a layer with infinite transverse dimensions, the disc may be expected to exhibit dimensional resonances which depend upon its transverse boundaries. Such resonant modes have been discussed in the literature, usually under the title of magnetodynamic modes. Unlike the magnetostatic modes, these dipolar modes are not limited to a frequency band determined directly by the biasing magnetic field.

As with the effect of demagnetising fields on the internal magnetic field uniformity, the theory of magnetodynamic modes available in the literature permits the estimation of the disc radius required to provide a desired degree of suppression of these latter excitations. Due to the need to restrict the disc size on grounds of economy when using single crystal magnetic materials, the preferred approach is to fit a relatively small single crystal disc with a collar of cheaper material having the same spontaneous magnetisation and, ideally, relative permittivity e.g. a single crystal YIG disc with a polycrystalline YIG collar Since the desired resonant modes (of the interference or Fabry-Perot type) tend to have their electromagnetic fields concentrated in the region of the disc adjacent to the irises, while the spurious modes (involving propagation of electromagnetic radiation up to and reflection from the transverse boundaries of the disc) have their electromagnetic fields spread throughout the disc, it is possible to envisage the suppression of the spurious modes by forming the collar from lossy material e.g. very broad linewidth polycrystalline YIG, or, alternatively, a suitably designed uniform suspension of iron filings (or other high spontaneous magnetisation, conducting magnetic material) in an epoxy resin (or other host medium which may be selected on the basis of suitable relative permittivity and ability to be cast).

When seeking to apply the filter principle, as indicated in FIG. 2 and as described above, it is necessary to mount the layer or layers of magnetic material in a microwave circuit or microwave circuits providing the required transducers 6, 10 between linearly polarised electromagnetic waves and right circularly polarised electromagnetic waves together with suitable microwave discontinuities 2, 3 whose precise configurations may depend upon the nature of the transducers 6, 10.

There are three basic methods of exciting circularly polarised electromagnetic waves. The first method concerns electromagnetic energy at microwave frequencies propagating through hollow waveguides (normally of rectangular or circular cross-section). Propagation is in the form of waveguide modes for which the electric and magnetic field patterns are well-defined and can be evaluated theoretically in many cases (e.g. for rectangular and circular cross-sections), provided the frequency of the electromagnetic energy lies within the frequency range for which the waveguide was designed to operate. If the frequency is too low the waveguide is cut-off, while if the frequency is too high more than just the one (fundamental) mode may propagate and the exact electromagnetic field configuration is uncertain. It is a characteristic of such a propagating mode that the lines of magnetic flux occur in the form of closed loops which propagate through the waveguide with a velocity dependent upon the frequency of the electromagnetic radiation and upon the guide dimensions. There are therefore positions on the inside walls of the waveguide at which the amplitude of the radio frequency magnetic field remains constant, but rotates through 211* radians at the samefrequency as the oscillations of the electromagnetic field. It is a well-known technique to couple such a circularly polarised radio frequency magnetic field out of a waveguide by inserting an iris (aperture) into the waveguide wall at an appropriate position.

The second method relates especially to circular waveguide. A circularly polarised radio frequency magnetic field may be envisaged as consisting of two spatially orthogonal linearly polarised fields which are 1r/2 radians out of phase. This indicates the feasibility of obtaining circularly polarised radio frequency magnetic fields by direct synthesis. One well known technique commonly utilised in circular waveguide propagating its fundamental TE, mode, consists of inserting a dielectric vane into the circular guide at an angle of 1r/4 radians to the plane of polarisation of the electric field of the incident electromagnetic field. The component of the incident electromagnetic having its electric field at right angles ot the vane is, to a first approximation unaffected by it, while, for a vane of the appropriate length, the component of the incident electromagnetic wave having its electric field parallel to the vane has its phase delayed by 1r/2 radians. Another well known technique involves the use of a meander line.

The third method concerns strip transmission lines. If an electromagnetic wave propagates through or upon a transmission line in which there is a discontinuity in dielectric constant in a direction transverse to the direction of propagation, then, according to Maxwells equations, there must be a magnetic field component in the direction of propagation which is proportional to the gradient in the electric field created by the dielectric loading. As in the case of propagation in waveguide, the radio frequency magnetic flux exists in the form of closed loops, and there are once more positions in which circularly polarised radio frequency magnetic fields are generated. For example, the transverse electromagnetic mode of propagation in coaxial line is not suitable for the development of non-reciprocalmicrowave devices as it has no longitudinal component of radio frequency magnetic field and so cannot furnish a rotating radio frequency field to drive the spin systems in magnetic media. However, it is a well-known technique to provide this required rotating field by means of assymmetric loading of the'coaxial line with dielectric material. Similarly many of the strip transmission lines known to microwave engineers have inherently asymmetric dielectric loading and hence positions of circularly polarised radio frequency magnetic field. By way of example, the methods of construction and the resultant radio frequency magnetic field configurations of two transmission systems, viz. slotline and coplanar waveguide, are illustrated in FIGS. 4a-b and Sa-b respectively. These particular transmission systems have been chosen since their magnetic field configurations render them particularly attractive for embodiment of the present tunable filter principle, as is described below with reference to FIGS. 9a-b and 10a-b.

Attention is directed to FIGS. 4a-b previously noted, wherein it can beseen that a dielectric substrate 51 has a thin layer 53 of conductor deposited on it. A slot 55 is removed from the layer 53, separating it into two portions 57 and 59. A radio frequency alternating voltage is applied between the portions 57 and 59 and as a result an electromagnetic wave is propagated along the slot 55.

As the electromagnetic wave is propagated an electric field pattern E is set up around the slot in planes at right angles to the direction of propagation. A magentic field pattern M is set up in the vicinity of the slot and concentrated close to a plane containing the direction of propagation and at right angles to the layer 53.

In both FIG. 4b and FIG. 5b the electric and magnetic field patterns have been shown separately for clarity. In fact, they do, in each case, interpenetrate in a manner prescribed by the solution of Maxwells equations for each transmission line geometry.

A coplanar waveguide is described with reference to FIGS. 5a and 5b, wherein a dielectric substrate 61 has a thin layer 63 of conductor deposited on it. Two parallel slots 65 and 67 are removed from the layer 63, separating it into two portions 69 (adjacent to the slot 65) and 71 (adjacent to the slot 67) and a centre strip 73 (bounded by the slots 65 and 67). A radio frequency alternating voltage is applied between the strip 73 (which acts as a live plane) and the portions 69 and 71 (which act as earth planes). As a result an electromagnetic wave is propagated in the direction of the strip 73.

As the electromagnetic wave is propagated an electric field pattern E, is set up which consists of a pattern around each slot 65 and 67 in planes at right angles to the direction of propagation. A saddle-shaped magnetic field pattern M, is set up around the strip 73.

FIG. 6 is a longitudinal cross-sectional diagram and FIG. 7 is a transverse cross-sectional diagram of a tunable microwave filter mounted in a rectangular waveguide. FIG. 6 is a longitudinal cross-sectional diagram on the line VI-VI of FIG. 7 and FIG. 7 is a transverse cross-sectional diagram on the line VII-VII of FIG. 6. A first waveguide 101 has an input port 103 and is terminated by a matching load 105. The waveguide 101 is reduced in height at a region 107, leaving one of the broad walls 109 flat. A microwave iris 111 is cut in the wall 109 between the centre line and the edge of the wall 109 in the region 107 in such a position that a quasi-TEM wave which is approximately right circularly polarised over the operating frequency range will be emitted from the iris 111.

A similar second waveguide 113 has an output port 115 at one end and a matched load 117 at the other. The waveguide 113 is reduced in height at a region 119, leaving one of the broad walls 121 flat. The waveguide 113 is arranged adjacent but not necessarily parallel to the waveguide 101 with their respective broad walls 109 and 121 adjacent and lying in parallel planes. A microwave iris 123 is cut in the wall 121 and positioned exactly opposite the iris 111.

A piece of ferrimagnet 125, having plane parallel sides and a cross-section larger than the irises, is mounted between the irises 111 and 123 with its sides orthogonal to the lines joining the centres of the irises 111 and 123. The ferrimagnetic material is preferably single cystal YIG although narrow linewidth polycrystalline YIG may also be used. Single crystal YIG is needed for broad tuning range devices since operation up to close proximity to uniform precession resonance will be needed. Away from uniform precession resonance, wall current and dielectric losses are the principal determining factor of the unloaded Q of the resonator and so polycrystalline material becomes usable.

As previously described, part of the material of the YIG disc may be comprised of a collar of poor quality polycrystalline YIG and other material having the same spontaneous magentisation as single crystal YIG.

The plane parallel faces of the ferrimagnetic disc are positioned as closely as possible to the inner surfaces of the broad walls 109 and 121 of the waveguide 101 and 113 respectively, and in physical contact with the outer surfaces of the waveguides 101 and 113. Propagation of electromagnetic radiation through a thick iris between the inside of a waveguide and the ferrimagnetic material is analagous to propagation through a length of evanescent waveguide. In order to avoid losses it is therefore desirable to make the thicknesses of the broad walls 109 and 121 as small as possible in the regions of the irises 111 and 123 respectively. One method of achieving this, which is illustrated in FIG. 6 and FIG. 7, is to let the ferrimagnetic disc 25 into the broad walls 109 and 121 as far as possible. Alternatively the plane parallel faces of the ferrimagnetic disc may be silver plated (except for the parts which form the irises 111 and 123) and used to form part of the actual wall of the waveguide. Care must be taken to ensure that a sound electrical contact is made between the silver plating on the plane parallel faces and the conducting material forming the inside surfaces 109 and 121 of the walls of the waveguide, so that leakage of electromagnetic radiation does not occur. Also, the waveguide walls or the silver plating must be many skin depths thick to prevent transmission of electromagnetic radiation between the waveguides and the ferrimagnetic disc at positions other than those defined by the irises 111 and 123.

A variable high magnetic field which has as far as possible a uniform flux density across the irises 11 1 and 123 is induced in a direction along a line joining the centre points of the irises 111 and 123 by means of a magnet 127 constructed as follows. A first pole piece 129 is fitted in a permanent magnet shell 131. The pole piece 129 carries an electrical winding 133 inside the shell 131. The inside end of the pole piece 129 is let into the wall of the waveguide 101 opposite the iris 111 as far as possible. Alternatively the pole pieces 129 may be silver plated and used to form part of the actual wall of the waveguide. A second pole piece 135 is fitted in a second permanent magnet shell 137. The pole piece 135 carries an electrical winding 139 inside the shell 137. The inside end of the pole piece 135 is let into the wall of the waveguide 113 opposite the iris 123 as far as possible. Alternatively the pole piece 135 may be silver plated and used to form part of the actual wall of the waveguide. The permanent magnet shells 131 and 137 are made to fit together in order to surround the waveguides 101 and 113 as far as possible. They may be located together by means of dowels 141.

The action of the arrangement is as follows.

The wave to be filtered is propagated down the waveguide 103 in the direction from the input port 103 towards the iris 111. As stated above, placing the irises 111 and 123 at. the correct distance from the narrow sides of the waveguides 101 and 113 respectively permits the coupling into the ferrimagnetic disc 125 of a quasi-TEM wave which is approximately right circularly polarised over the operating frequency range. Multiple reflections of this wave between the two irises 111 and 123 then give rise to the interference effects needed to develop pass-and stop-bands. Thus the wave emerging from the iris 123 and eventually from the output port 115 will contain only the desired frequency or frequencies.

For a typical X-band (8.0-l2.4GI-Iz) interference filter using R100 waveguide (International Electrotechnical Commission designation), also known as WG16 waveguide (Radio Components Standardisation Committee designation), the irises 111 and 123 are mm diameter circular apertures with their centres located 5.4mm from the sidewall of the-waveguide, while the ferrimagnetic disc 125, is a 15mm diameter by 1.5mm thick disc of single crystal YIG. The biasing magnetic field intensities required to achieve tuning over the band are approximately 4.4 X to 6.0 X 10 A m.

The use of waveguide irises confers two significant advantages. Firstly, the irises present large susceptances to the quasi-TEM wave propagating in the ferrimagnetic material, so providing the large reflection coefficients needed to develop narrow-transmission fringes, and adjustment of the irises dimensions permits any desired compromise between the filter insertion loss and bandwidth. Secondly the irises directional properties can be used to ensure that any energy reflected from the ferrimagnetic resonator tends to be propagated on down the input waveguide 101 in the direction of the matched load 105 rather than being reflected back towards the input port 103, while the transmitted energy tends to propagate only towards the output port 115 of the coupled waveguide 113. This aspect of the operation of the device results in its having a low voltage standing wave ratio at all frequencies, since untransmitted energy is not reflected from the device as in a resonance filter.

For high power applications in which there may be appreciable power dissipation in the ferrimagnetic material irises of moderate thickness may have to be tolerated for their ability to conduct away heat. Where the ferrimagnetic layer is let into the waveguide wall, the circumference of the counter sunk section of waveguide wall must make good thermal contact with the transverse edge of the disc.

In order to ensure that the minimum amount of microwave energy is reflected by the iris 111 in FIGS. 6 and 7 back towards the input port it is preferred to use a non-circular iris. For example, the irises may be elliptical, placed at the correct distance from the axis of both waveguides for the development of a circularly polarised rf magnetic field, and with their major axes oriented with respect to the axis of the waveguide at the angle corresponding to zero reflected signal at the centre of the operating frequency range.

Since the operation of the filter results in untransmitted energy being propagated into the matched load 105, it is possible to devise a filter having two output waveguides, one for the passband via an output port corresponding to the output port 115 and one for the stopband, having an output port where the matched load 105 is in FIG. 6.

In addition, any signal which may be incident upon the output port cannot excite the correct sense of circular polarisation at the iris 123 and so cannot couple into the ferrimagnetic disc 125. The input port is therefor isolated from any signals incident upon the output port.

FIG. 8 is a longitudinal cross-sectional diagram of a tunable microwave filter mounted in a circular waveguide. The filter consists of two microwave irises 2, 3 separated by a disc 1 of ferrimagnetic material. The microwave iris 2 is formed in conducting material 20 which is either deposited on one face of the disc 1 or is an entirely separate piece of material. The material 20 separates the disc 1 from a piece 14 of dielectric material and the microwave iris 3 is formed in conducting material 30 which is either deposited on the other face of the disc 1 or is an entirely separate piece of material. The material 30 separates the disc 1 from a further piece 16 of dielectric material.

The ferrimagnetic material of which the disc 1 is made is preferably single crystal YIG although good quality (narrow linewidth) polycrystalline YIG may also be used. As previously described, part of the material of the disc 1 may consist of a collar of poor quality polycrystalline YIG or other material having the same spontaneous magnetisation as single crystal YIG.

The filter with the piece 14 of dielectric material and the piece 16 of dielectric material is mounted in a circular waveguide 18. However, provided always that the plane, parallel faces of the disc 1 of ferrimagnetic material are entirely covered by highly conducting material 20, 30 (the material may be for example silver) except at the positions of the microwave irises 2, 3, there is no restriction of the diameter of the disc to be the same as the inside diameter of the circular waveguide 18.

The comments made above with reference to FIG. 6 and FIG. 7 in regard to the need to maintain physical contact between the ferrimagnetic disc and the metal layers, and to have irises with minimum thickness also apply in this embodiment.

Two horseshoe permanent magnets, 13, 15 are arranged on opposite sides of the waveguide 18 with corresponding magnetic poles adjacent to each other and common pole pieces 17, surrounding the piece 141 of dielectric material, and 19, surrounding the piece 16 of dielectric material. This arrangement produces a strong magnetic field along the axis of the waveguide 18. A coil 21, wound around the waveguide 18 in the vicinity of the filter, provides means of varying the magnetic field, and hence of tuning the filter.

The dielectric-filled circular waveguide 18 is approached from an air-filled rectangular waveguide (not shown) via a dielectric loaded taper transition 23. Between the transition 23 and the pole piece 17 of the magnets 13 and 15 the waveguide 18 contains a suitably orientated attenuating vane 25 and a suitable quarter wavelength dielectric vane 27. The purpose of the attenuating vane 25 is to remove any waves reflected from the surface of the filter itself having the wrong direction of linear polarisation for propagation back down the rectangular waveguide. The purpose of the quarter wavelength dielectric vane is to convert the incident linearly polarised radiation into right circularly polarised radiation, and to convert any circularly polarised radiation reflected from the filter back to linear polarisation. Beyond the pole piece 19 of the magnets 13 and 15 the waveguide 18 contains a further quarter wavelength dielectric vane 29 and an attenuating vane 31. The purpose of the quarter wavelength dielectric vane 29 is to convert the polarisation form from circular back to linear, and the purpose of the attenuating vane 31 is to remove any residual waves of the wrong direction of linear polarisation. Beyond the attenuating vane 31 the waveguide 18 is attached to a further dielectric loaded transition 33 from dielectricfilled circular waveguide to air-filled rectangular waveguide.

This type of filter reflects all untransmitted energy (and hence has a very high voltage standing wave ratio outside the passband) and does not provide isolation of the input port from signals within the passband incident upon the output port.

As previously noted, FIGS. 9a-b are respectively plan and longitudinal cross-sectional views of a tunable microwave filter mounted in a slotline, the cross-section diagram 9b taken along lXb IXb of FIG. 9a.

A dielectric substrate 201 carries a metal layer 203. A ferrimagnetic disc 205 has two plane parallel faces 207, 209 plated with highly conducting material (preferably silver) except at the position of a central iris 211, 213 respectively. The disc 205 is inserted into the centre of the substrate 201 with its plane parallel faces 207, 209 both perpendicular to the surface of the substrate 201. Two slotlines 215, 217 are formed in the layer 203 and approach very close to the disc 205 in the regions of the irises 211, 213 respectively. In other parts of the layer 203 the slotlines 215, 217 are sufficiently far apart for no appreciable direct coupling of electromagnetic energy to occur between them.

A variable high magnetic field is provided via pole pieces 219, 221 set into the substrate 201 and layer 203.

The depth of insertion of the disc 205 and the design of the slotlines 215, 217 is such that either iris 211 or 213 can be exposed to a right circularly polarised radio frequency magnetic field, depending upon which of the slotlines 215 or 217 is used to carry the input signal. In FIG. 9a the widths of the slotlines 215, 217 and the thickness of the substrate 201 have been exaggerated for clarity. Because of the configuration of the radio frequency magnetic field, as previously described with reference to FIG. 4a-b, there are regions both in the substrate and in the air above it in which the irises can be exposed to right circularly polarised radio frequency magnetic fields.

In action radiation to be filtered enters at an input port 223 at one end of the slotline 215. Radiation at the bandpass frequency is extracted by the action of the disc 205 and launched down the other slotline 217 in the same direction towards a bandpass output port 229. Radiation at other frequencies is directed down the slotline 215 to a bandstop output port 225. The remaining port 227 of the slotline 217 is connected to a matched load (not shown).

Although the iris 211 through which the right circularly polarised radio frequency magnetic fields couple into the ferrimagnetic material of which the disc 205 is made is shown circular, both the irises 211, 213 may be of elongated or elliptical shape, provided always that they have their major axes oriented parallel to the layer 203 and always lie at positions of right circularly polarised radio frequency magnetic field. The lengths of non-circular irises must not be so great as to allow them to extend into regions of non-uniform wave number (or of high loss, when the ferrimagnetic disc is fitted with a collar as described above). The widths of the irises should be such that they do not intercept appreciable non-right circularly polarised radiation.

In the case of slotline the electromagnetic fields are concentrated in the region of the slot so that the pole tips need only be more than about is). away from the slot to have a negligible effect on propagation down it, where A is the free space wavelength of the electromagnetic radiation. Conversely, the width of the metal layers remaining between the slots and the disc should be very much less than 16)..

FIGS. 10a-b show respectively plan and longitudinal cross-sectional diagrams of a tunable microwave filter mounted in a coplanar waveguide, wherein the longitudinal cross-sectional diagram 10b is a cross-section along the line Xb Xb of FIG. 10a.

A dielectric substrate 251 carries a metal layer 253. A ferrimagnetic disc 255 has two plane parallel faces 257, 259 plated with highly conducting material (preferably silver) except at the position of a central iris 261, 263 respectively. The disc 255 is inserted into the centre of the substrate'251 with its plane parallel faces 257, 259 both perpendicular to the surface of the substrate 251. Two coplanar waveguides 265,267 are formed in the layer 253 and approach very close to the disc 205 in the regions of the irises 261, 263 respectively. In other parts of the layer 253 the coplanar waveguides 265, 267 are sufficiently far apart for no appreciable direct coupling of electromangetic energy to occur between them.

A variable high magnetic field is provided via pole pieces 269, 271 set into the substrate 251 and layer 253.

The depth of insertion of the disc 255 and the design of the coplanar waveguides 265, 267 is such that either iris 261 or 263 can be exposed to a right circularly polarised radio frequency magnetic field, depending upon which of the coplanar waveguides 265 or, 267 used to carry the input signal. In FIG. 10, the widths of the coplanar waveguides 265, 267 and the thicknesses of the substrates 251 have been exaggerated for clarity.

In action radiation to be filtered enters at an input port 273 at one end of the coplanar waveguide 265. Radiation at the bandpass frequency is'extracted by the action of the disc 255 and launched down the other coplanar waveguide 267 in the same direction towards a bandpass output port 279. Radiation at other frequencies is directed down the coplanar waveguides 265 to a bandstop output port 275. The remaining port 277 of the coplanar waveguide 267 is connected to a matched load (not shown).

Although the iris 261 through which the right circularly polarised radio frequency magnetic fields couple into the ferrimagnetic material of which the disc 255 is made is shown to be circular, both the irises 261, 263 may be alongated or elliptical shape, provided always that'they have their major axes oriented parallel to the layer 253 and always lying in positions of right circularly polarised radio frequency magnetic field. The lengths of non-circular irises must not be so great as to allow them to extend into regions of non-uniform wave number (or of high loss, when the ferrimagnetic disc is fitted with a collar, as described above). The widths of the irises should be such that they do not intercept appreciable non-right circularly polarised radiation.

In the case of coplanar waveguide, examination of the field configuration shown in FIG. b suggests that, as a first order approximation, the insertion of a highly conducting metal plane approximately midway between the live plane and an earth plane and perpendicular to the substrate should not disturb the field profile in the vicinity of the live plane, i.e. the regions of right circular polarisation will still exist. Therefore, as shown in FIG. l0a-b, the coplanar waveguides 265, 267 are so positioned relative to the disc 255 that in the vicinity of each of the irises 261, 263 respectively one of the earth planes is missing but the metallised faces 257, 259 respectively of the disc 255 lie at distances from the live planes of the coplanar waveguides 265, 267 of approximately half the usual distance between the live and earth planes.

For typical X-band (8.0 to 12.4 GI-Iz) interference filters using slotline and coplanar waveguide the dimensions of the single crystal YIG disc and the biasing magnetic field intensities are the same as for the embodiment in R100 rectangular waveguide, described above with reference to FIG. 6 and FIG. 7. A typical slot width would be 0.5mm, while for the coplanar waveguide a typical live plane width would be 0.75mm with a 0.5mm live to earth plane separation.

The device described above may suffer from a nonlinear relationship between passband frequency and biasing magnetic field intensity. In addition, there will be hysteresis effects in the relationship between the biasing field and the solenoid current. It is therefore desirable to include some magnetic field sensing element (e.g. magnetoresistance or Hall effect probes) in the magnetic circuit of the filter, and to feed the output signal from this probe into a suitably designed electronic control circuit (providing the current for the solenoid) which can ensure a linear, hysteresis-free, relationship between any external programming voltage applied to it and the intensity of the resultant biasing magnetic field. Similarly, the non-linear relationship between frequency and field can be compensated by including a type of circuit known as a diode-resistor function synthesiser within the solenoid current control circuit. Both the control circuit and the diode-resistor function synthesiser involve I techniques well known to those skilled in the design of electronic circuitry.

In order to improve on the insertion loss/bandwidth characteristics attainable with the filter having a single ferrimagnetic disc 25, of ferrimagnetic material, multilayer structures may be used, in which there must be pronounced discontinuities in wave impedance between adjacent layers. The preferred approach is to use several ferrimagnetic discs, comparable with the original single ferrimagnetic disc layer 125, separated by further microwave irises, comparable with the original irises Ill and 123. The use of alternate layers of ferrimagnetic materials having different spontaneous magnetisations is impractical since large differences in spontaneous magnetisation would be required to produce large reflection coefficients for electromagnetic radiation at the interfaces between the alternate layers, and demagnetising field effects would then prevent all the layers from being biased simultaneously into the below uniform precession resonance, region or the above cut-off region. The use of two layers of ferrimagnetic material separated by a single layer of dielectric material is also impractical since the inclusion of a magnetic field dependent layer would result in a filter exhibiting only a small rate of change of passband frequency with magnetic field.

All microwave oscillators and amplifiers generate noise signals to some extent, and the congestion of the available microwave spectrum has made it increasingly necessary to suppress these spurious emissions. Due to the low voltage standing wave ratio obtainable with this filter at all frequencies together with the isolation of the input port from the output, the filter can be used to re duce the tendency to spurious oscillations which may occur with microwave amplifiers, such as travelling wave tubes, when feeding into highly mismatched loads; similarly the isolating action can reduce any tendency towards frequency pulling of a tunable oscillator when feeding into a highly mismatched load. Due to the physical principles on which the filter is based it can be used to reduce spurious emissions at harmonic frequencies. The second harmonic of the passband frequency will normally occur in the frequency range for which the magnetic material exhibits very high absorption, while at this and higher harmonics, not only will efficient excitation of a right circularly polarised wave be unlikely, but also the layer or layers of magnetic material in the filter will not be integral multiples of a half wavelength thick.

The filter may also find use as a frequency dependent feedback element operating in conjunction with a broadband microwave amplifier (for example, a travelling wave tube), thus constituting a high power electronically tunable microwave oscillator. In this application a relatively large insertion loss can be tolerated, so allowing a very narrow bandwidth filter to be designed and used. Alternatively the filter may find use in such devices as panoramic receivers, frequency meters and tunable frequency discriminators.

Since the magnetic material used in the filter must be biased so that the uniform precession resonance is either immediately above or immediately below the frequency range of interest (depending on which region of the dispersion diagram is being used), it follows that very high magnetic fields must be developed in the magnetic material when this frequency range extends up towards hundreds of GI'Iz. This problem may be overcome by utilising magnetic materials which can develop high internal effective magnetic fields. For example, for a uniaxial single crystal antiferromagnet with a biasing field, H, parallel to, and a radio frequency field perpendicular to the axis, there are two resonant frequencies f i given by:

f 7 [(H ZH H HE F H] y [H 1 H] for temperatures well below the Neel temperature. H is the efiective exchange field, which may be of the order of 10 A m, and H is the anisotropy field which normally lies in the range 10 to 10 A m. The antiferromagnet thus behaves as if it were biased with a magnetic field, H These resonance frequencies are respectively right and left circularly polarised and can be varied by the biasing magnetic field. As in the ferrimagnetic case, the material exhibits large effective penneabilities, p. in the vicinity of the resonances, and these permeabilities are dependent on the signal frequency, f, and on the effective biasing magnetic field (H H). Since antiferromagnets tend to have broad linewidths, the relatively low absorption associated with the off-resonance operation of these interference filters is especially important in this case.

We claim:

1. A microwave filter including at least one body of magnetic material capable of propagating spinwaves having plane parallel faces with the dimension perpendicular to said faces being small compared with the dimensions parallel to said faces, means for feeding the microwave radiation to be filtered into said body in the form of right circularly polarised electromagnetic waves in a direction having a substantial component perpendicular to said faces, a partial-reflectioninducing discontinuity to the transmission of said right circularly polarised electromagnetic waves at said faces, means for applying a magnetic field in a direction having a substantial component perpendicular to said faces and means'for varying said magnetic field.

2. A microwave filter as in claim 1 and in which said magnetic material is ferrimagnetic material.

3. A microwave filter as in claim 1 and in which said body of magnetic material is a circular disc.

4. A microwave filter as in claim 2 and in which said body of magnetic material is a circular disc.

5. A microwave filter as in claim 1 and in which said means for feeding microwave radiation to be filtered includes a waveguide.

6. A microwave filter as in claim 5 and in which said waveguide is a rectangular waveguide having an iris so positioned in the waveguide wall as to feed right circularly polarised electromagnetic waves into said body, said iris constituting said partial-reflection-inducing discontinuity.

7. A microwave filter as in claim 5 and in which said waveguide is a circular waveguide having a vane so positioned in said waveguide as to feed right circularly polarised electromagnetic waves along said waveguide.

8. A microwave filter as in claim 7 and in which said partial-reflection-inducing discontinuity is an iris.

9. A microwave filter as in claim 1 and in which said means for feeding microwave radiation to be filtered includes a strip transmission line in which radiation is guided by conducting layers in a dielectric substrate.

10. A microwave filter as in claim 9 and in'which said strip transmission line is a slotline positioned parallel to said faces in a plane perpendicular to said faces and said conducting layers and surface of the substrate are perpendicular to the faces of the filter.

11. A microwave filter as in claim 10 and in which said partial-inducing discontinuity is an iris.

12. A microwave filter as in claim 9 and in which said strip transmission line is a coplanar waveguide positioned parallel to said faces in a plane perpendicular to said faces and said conducting layers and surface of the substrate are perpendicular to the faces of the filter.

13. A microwave filter as in claim 12 and in which said partial-reflection-inducing discontinuity is an iris.

14. A microwave filter as in claim 2 and in which said means for feeding microwave radiation to be filtered includes a waveguide.

15. A microwave filter as in claim 14 and in which said waveguide is a rectangular waveguide having an iris so positioned in the waveguide wall as to feed right circularly polarised electromagnetic waves into said body, said iris constituting said partial-reflectioninducing discontinuity.

16. A microwave filter as in claim 14 and in which said waveguide is a circular waveguide having a vane so positioned in said waveguide as to feed right circularly polarised electromagnetic waves along said waveguide.

17. A microwave filter as in claim 16 and in which said partial-reflection-inducing discontinuity is an iris.

18. A microwave filter as in claim 2 and in which said means for feeding microwave radiation to be filtered includes a strip transmission line in which radiation is guided by conducting layers in a dielectric substrate.

19. A microwave filter as in claim 18 and in which said strip transmission line is a slotline positioned parallel to said faces in a plane perpendicular to said faces and said conducting layers and surface of the substrate are perpendicular to the faces of the filter.

20. A microwave filter as claimed in claim 19 and in which said partial-reflection-inducing discontinuity is an iris.

21. A microwave filter as in claiml8 and in which said strip transmission line is a coplanar waveguide positioned parallel to said faces in a plane perpendicular to said faces and said conducting layers and surface of the substrate are perpendicular to the faces of the filter.

22. A microwave filter as in claim 21 and in which said partial-reflection-inducing discontinuity is an iris.

23. A microwave filter as in claim 3 and in which said means for feeding microwave radiation to be filtered includes a waveguide.

24. A microwave filter in claim 23 and in which said waveguide is a rectangular waveguide having an iris so positioned in the waveguide wall as to feed right circularly polarised electromagnetic waves into said body, said iris constituting said partial-reflection-inducing discontinuity.

25. A microwave filter as in claim 23 and in which said waveguide is a circular waveguide having a vane so positioned in said waveguide as to feed right circularly polarised electromagnetic waves along said waveguide.

26. A microwave filter as in claim 25 and in which said partial-reflection-inducing discontinuity is an iris.

27. A microwave filter as in claim 3 and in which said means for feeding microwave radiation to be filtered includes a strip transmission line in which radiation is guided by conducting layers in a dielectric substrate.

28. A microwave filter as in claim 27 and in which said strip transmission line is a slotline positioned parallel to said faces in a plane perpendicular to said faces and said conducting layers and surface of the substrate are perpendicular to the faces of the filter.

29. A microwave filter as in claim 28 and in which said partial-reflection-inducing discontinuity is an iris.

30. A microwave filter as in claim 27 and in which said strip transmission line is a coplanar waveguide positioned parallel to said faces in a plane perpendicular tosaid faces and said conducting layers and surface of the substrate are perpendicular to the faces of the filter.

31. A microwave filter as in claim 30 and in which said partial-reflection-inducing discontunuity is an iris.

32. A microwave filter as in claim 4 and in which said means for feeding microwave radiation to be filtered includes a waveguide.

33. A microwave filter as in claim 32 and in which said waveguide is a rectangular waveguide having an iris so positioned in the waveguide wall as to feed right circularly polarised electromagnetic waves into said body, said iris constituting said partial-reflectioninducing discontinuity.

34. A microwave filter as in claim 32 and in which said waveguide is a circular waveguide having a vane so positioned in said waveguide as to feed right circularly polarised electromagnetic waves along said waveguide.

35. A microwave filter as in claim and in which said partial-inducing-discontinuity is an iris.

36. A microwave filter as in claim 4 and in which said means for feeding microwave radiation to be filtered includes a strip transmission line in which radiation is guided by conducting layers in a dielectric substrate.

37. A microwave filter as in claim 36 and in which said strip transmission line is a slotline positioned parallel to said faces in a plane perpendicular to said faces and said conducting layersand surface of the substrate are perpendicular to the faces of the filter.

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Classifications
U.S. Classification333/209, 333/236, 333/208, 333/21.00A, 333/204
International ClassificationH01P1/215, H01P1/20
Cooperative ClassificationH01P1/215
European ClassificationH01P1/215