|Publication number||US3750175 A|
|Publication date||Jul 31, 1973|
|Filing date||Nov 25, 1969|
|Priority date||Dec 14, 1967|
|Publication number||US 3750175 A, US 3750175A, US-A-3750175, US3750175 A, US3750175A|
|Inventors||Cooper R, Goode G, Lockerd R, Smith M|
|Original Assignee||Texas Instruments Inc|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (3), Referenced by (62), Classifications (12)|
|External Links: USPTO, USPTO Assignment, Espacenet|
United States Patent 1 Lockerd et a]. 1*July 31, 1973  MODULAR ELECTRONICS 234K711 ii/1958 Ring 343/ 100 SA UX COMMUNICATION EM 3,553,693 1/1971 Lockerd et al. 343/100 SA  Inventors: Robert M. Lockerd; Mark W. Smith, 'p, Examine, Benjamin Borcheh both of Dallas; Ray Cooper; Assistant Examiner-Richard E. Berger F 'S Gwde, both of Attorney-James 0. Dixon, Andrew M. Hassell, Harold Rlchardsonv of Levine, John E. Vandigriff, Ren E. Grossman, Rich-  Assignee: Texas Instruments Incorwnted' ards, Harris and Hubbard, D. Carl Richards, William D. Dallas Harris, Jr., E. Mickey Hubbard, V. Bryan Medlock, Jr., a Harold E. Meier, .Ie W. Mills and Timoth L. Notice: The portion of the term of the patent Bur s y subsequent to Jan. 5, 1988 has ges been disclaimed. 57 B TRACT A  Filed: Nov. 25, 1969 l l s A communication system including a plurality of radi- PP N04 879,775 ating elements formed into an antenna array for trans- Rehted s Application 1 mitting and receiving communication frequency signals  Continuation-impart of Ser. No. 690,638, Dec. 14, "f 3 generate the 1967 Pat 3 553 transmitted signals and process the received frequencies through a manifold arrangement. Each radiating 521 US. Cl 343/100 SA element mnects the fl "wdule 51 Int. Cl. H0lq 3/26 made up of integrated microwave circuitry including a  Field of Search 343/100 SA mixer coupled a and a P Shift" coupled to a beam steering computer. By means of the  Reerences Cited beam steering computer the antenna can be made to UNITED STATES PATENTS scan various preselected areas.
3,386,092 5/1968 Hyltin 343/100 SA X 4 Claims, 11 Drawing Figures 7.725-7.85O GHZ (4mm mod") x3 25OMW POWER l0 MW PHASE ZOMW PRE 2.575-2.6I7GH: MULT. AMP. r SHIFT. AMP 2 MW (-430) (04am 4 an INSTRUCTION LOGIC I62 moo-mas GHz (-50% m u GHz) 020d) some MHz F FILTER MIXER FILTER AMP. use m m I76 Idb 0i 7 I25 (3H: GHZ (46b NF) -40db at 7.725 GHz 3db) lOdb) x3 i0 MW PHASE ZOMW p 2.208 GHZ MULT. SHIFT AMP. 2 MW we I68 172 4 an msmucnou LOGIC PATENHDJUL3HQT5 3.750.175
SHEEI 2 I]? 5 L KZ W L 29 24 2.0 GHz :6 MHZ P A coNTRoL REcEIvE SIGNAL 3 I MA I F GLD ANTENNA I 1 5 'Y (-3db) 28 ELEMENT I F '3 2 PHASE MANIFOLD l EILTER a 7 34 I S I%I I PRE I PFIZ'ESSSNG 30 db) PHASE MANIFOLD SHIFT 36 I THREE I I (-3 db) PHASE I MANIFOLD l SHIFT l FOUR L g u 3 FIG. 3 L-BEAM STEERING TRANSMIT SIGNAL -L0GAL oSGILLAToR 2.0 GHz F ANTENNA 4 BIT CONTROL 2/ ELEMENT coNTRoL 30o mw J MANIFOLD I AMgllFER ossmw mfi g (3cm l-45mw SIGNAL I d I I-2 db) l SIGNAL PRocESSING L 35 l 22 FIG 4 BLOCK n 1 EXTRA PHASE FRONT --I5EAII/I STEERING LoGAL OSCILLATOR -e ScAN DIRECTION QUANTIZED PHASE IFRONT .FIG. 9
PATENIED JUL3 1 ms SHEET '4 0F 5 FIG. 8
PATENTEU 1 3,750,175
SHEET 5 OF 5 FIG. ll
A 7.7257850 GHZ 3db) (HOdb) 3 25OMW POWER IO MW PHASE 20 MW 5 2.575-2.6|7GHz MULT. AMPv SHIFT. AMP. 2 MW I86 I86 I80 I78 (-4db) (+l4db) 4 BIT INSTRUCTION LOGIC I82 V 7000-7125 GHZ (-50db of 1.1 GHZ) (+db) 5001625 MHZ "I FILTER MIXER MP. f FILTER I A 62' /64 6.625 GHZ I74 I76 (4cm NF) -iob of TI I GHz 4 MW -db 0T 7.725GHZ (-3db) (+lOdb) A MULT. SHIFT AMP 2 MW KN I I I I 7 I66 I68 I72 4 B INSTRUCTION LOGIC I70 INVENTOR:
ROBERT M. LOCKERD, ef 0/ ATTORNEY MODULAR ELECTRONICS COMMUNICATION SYSTEM This application is a continuation-in-part application of U.S. Pat. No. 3,553,693, and assigned to the same assignee as this patent application. I
This invention relates to a communication system involving solid state microwave modules, and more particularly to a modular electronic communication system having a multi-element phased array antenna.
While this invention is immediately advantageous in connection with construction and operation of satellite and airborne communication systems, it has application to other communication systems such as those used for mobile ground applications and microwave transmission.
In comparison to conventional communication systerns, a modular electronics system provides the advantages of less weight per watt of effective radiating power, noninertial electronic beam steering (where required), and increased system reliability because of the use of integrated circuits and distributed-function, building block construction. These advantages are of particular importance for satellite communication and airborne communication systems. However, they are also important to ground based systems especially those requiring high reliability.
Satellite and airborne communication systems have been faced with the problems of minimizing weight and increasing reliability while generating high power microwave energy. Other major problems of conventional mobile communication systems have been concerned with auxiliary equipment such as rotary joints, servo motors for the antennas, and the like. Restrictions imposed by such components on reliability exist in the most modern of transistorized systems produced for mobile service. Further, the magnetrons for transmitting, and klystrons for local oscillator service all have been found to restrict the reliability of the system.
In accordance with the present invention, there is provided an improved communications system in which solid state circuitry is so constructed and arranged as to be capable of overcoming the major obstacle heretofore encountered in the development of such a system, namely, the generation of high power microwave energy. This problem is overcome by the use of solid state functional electronic blocks or modules, so constructed as to operate as a modular antenna array which may be responsive to beam steering control and which may be operated at an adequate power level. The electronic blocks may be designed to operate at almost any communications frequencies. At the same time, such construction lends itself to a lightweight multi-element antenna array using electronic beam scanning thereby eliminating wave guides, rotary joints, motors, synchros, gears, and other servo components normally essential to a moving antenna communication system. As a result, a substantial reduction in total volume and weight over known or existing systems is achieved. This is accompanied by a substantial increase in the reliability of the system.
Another advantage of the phase array integrated circuit communication system is the ability to restrict the transmit and receive beam width thereby increasing the antenna efficiency. In conventional satellite or aircraft communication systems, the beam width must be made sufficiently large to completely cover a desired section plus an area outside the desired section necessitated by tolerances in the satellites or aircrafts attitude control system. Because of the much wider beam width in the conventional communication system, a transmitter must have a greater transmitter power to achieve effective communication.
Beam pointing and control in a phased array is accomplished by the setting of phase shifters in series with each radiating element. A beam scanning computer calculates the necessary phase for each element in order to collimate the beam at some position in space and generates an individualized digital code for each element. Typically, the scan computer is a digital device and the variable phase shifter is either a digital device or a digitally controlled analog device. Phased array antennas are formed by radiating elements arranged in any desired pattern, for example, a planar array of rows and columns. In order to control the direction of the antenna beam, an individualized digital code is provided to the phase shifter for each of the radiating elements in the antenna array. In operation, the digital computer generates individual codes for each radiating element depending on its position in the array. This code is applied to the phase shifter to impart a predetermined amount of phase shift to the signal received or transmitted from the element. If each phase shifting code imparts a particular degree of phase shift, the entire beam can be pointed in any desired direction. The phase shifting code may be applied to all the elements simultaneously or on a sequential basis. In the sequential mode, one phase shifter at a time would be updated, while all other shifters would remain unchanged.
Each radiating element in the antenna array is part of an antenna module with individual power generation and phase control circuitry. The use of solid state modules, together with microwave transistors, permits op eration in the X-band and higher. Production of trans mitting and receiving power at the desired frequency involves the use of frequency multipliers and upconverters of integrated circuit construction. Thus, each transmit module includes the necessary circuitry for amplifying relatively low IF energy applied simultaneously to all of the modules, and multiplying the frequency to a higher frequency for transmission from the transmit radiating elements. Each receive module includes circuitry for processing one or more high fre' quency signals received by the receive radiating elements to produce low frequency IlF signals, which are also preamplified. In addition, each module includes phase shifting means for the transmitted or received energy so that the beams to or from the fixed antenna array can be electronically scanned.
The phased'array integrated circuit communication system of this invention as set forth in the appended claims includes a multiple radiating element antenna coupled to a signal processor through amplification and phase shifting devices. For X-band frequency opera tion, a local oscillator output frequency is multiplied and mixed with an IF signal for boththe transmit and receive operation.
A more complete understanding of the invention and its advantages will be apparent from the specification and claims and from the accompanying drawings illustrative of the invention.
Referring to the drawings:
FIG. 1 is a pictorial of a satellite mounted modular electronic communication system employing independent transmit and receive antenna arrays;
FIG. 2 is a basic block diagram of a complete phased array integrated circuit communication system with independent transmit and receive antennas;
FIG. 3 is a block diagram of a multiple signal low frequency modular electronics receiving system;
FIG. 4 is a block diagram of a low frequency modular electronics transmitting system;
FIG. 5 is a block diagram of a X-band modular electronics receive and transmit communication system;
FIG. 6 is an integrated circuit mixer employing a surface-oriented diode wafer and ceramic substrate;
FIG. 7 illustrates a video amplifier construction for the IF preamplifier of FIG. 5;
FIG. 8 is a top view of an intergrated circuit embodiment of a multiplier of FIG. 5;
FIG. 9 is an illustration of beam scanning; and
FIG. 10 is a block diagram of an X-band modular electronic communications system using a single phased array antenna for transmit and receive.
FIG. 11 is another embodiment of the modular electronic communication system according to the present invention.
This invention will be described as it is employed as a satellite communication system contained within a satellite l0, referring to FIG. 1, having a phased array transmitting antenna 11 and a similar phased array receiving antenna 12. A plurality of booms 13-18 provide a means for instructing the satellite 10 to maintain a predetermined position relative to the earth. This orientation is such that the antenna arrays 11 and 12 are always facing toward the earth's surface. Electrical power to operate the communication system of the present invention is provided by means of solar cells 19 covering all outer surfaces of the satellite 10 except for areas covered by the antennas 11 and 12. Solar cell power supplies are now well known in the art and additional description is not deemed necessary.
By using separate transmitting and receive antennas 11 and I2 separated by several feet, greater isolation between transmit and receive modes is possible. Each of the antennas 11 and 12 comprises an array of individual radiating building blocks terminating in a radiating element, such as the crossed dipole 21. Other radiating element configurations that can be used include the slotted dipole configuration, the orthogonal slot configuration, and various open ended wave guides. The antennas 11 and 12 consist of no less than two radiating elements with the maximum number limited only by practical considerations. An antenna array for a 3-degree beam width with 0.75). element-to-element spacing requires approximately 550 radiating elements.
Each radiating element 2la, 21b, 2In of the antenna ll, referring to FIG. 2, is coupled to a manifold 22 through a modular electronic transmitting building block 23a. 23b, .23n. respectively. A central processor 24 composed of a limiter and power amplification section provides a signal to the manifold 22 for distribution to the radiating elements of the antenna 11. For the system shown, the central processor 24 could be either a frequency translationrepeater or demod-remod system, either of which to be a limiting or linear system. The relative amplitude and phase of the signal supplied to each of the radiating elements are controlled to obtain the desired radiation pattern from the combined action of all the elements in the antenna 11. A local os cillator 26 coupled to the manifold 22 provides power to the transmitting building blocks for use in the frequency conversion process where required. A beam steering computer 27, also coupled to the manifold 22, provides the phasing information for beam steering purposes.
A system similar to that described in the preceding paragraph is also provided to process communication signals received by the antenna 12 made up of radiating elements 28a, 28b, 28n. Each signal received by elements of the antenna 12 is transmitted to a manifold 29 through a modular electronic receiving block 31a, 31b, 3ln. The beam steering computer 27 again provides phasing information to the signals received by the antenna 12 prior to processing by the central signal processor 24. Where required, the local oscillator 26, coupled to the manifold 29, converts the transmitted frequency into an IF signal for handling by the central signal processor 24. Thus, the basic modular electronic communication system contains six major elements: the receive antenna array 12, the central processor 24, the transmit antenna array 1 1, the manifolds 22 and 29, the beam steering computer 27, and the local oscillator 26.
Referring to FIG. 3, there is shown a simple low frequency (2.0 Gl-iz) receiving module including a filter and pre-amplifier 32 and four phase shifters 33, 34., 36, and 37, one for each of four possible signals received by a radiating element. The radiating element 28 connects to the input of the pre-amplifier and filter 32. Each of the phase networks 33, 34, 3 6, and 3'7 receives a control signal from and transmits a signal to an associated section of the manifold 29. The various sections of the manifold 29 in turn transmit and receive signals from the signal processor 24.
In an exemplary operation of a multiple phase shifter and manifold receive system, the four receive signals from the element 28 are coupled to each phase shifter 33, 34, 36 and 37. A given phase displacement is ap= plied to each signal in each phase shifter in accordance with the control code connected thereto. The signals from all the phase shifters coupled to one manifold are then summed in such a manner that only one of the received signals will be coherent. The remaining three will be non-coherent resulting in an unintelligent signal (noise), which is rejected by the central processor 24. Each phase shifter and manifold operates in a similar manner to coherently add one received signal and noncoherently add the other three.
A typical Z-GI-lz preamplifier includes thin film resistors and capacitors. and several amplification stages employing chip transistors in a microstrip circuit on a ceramic substrate. Such a device will provide low noise amplification of the signal received at the radiating element 28. Phase shifting with a 4-bit phase shifter. such as phase shifters 33, 34, 36 and 37, is obtained by switching two PIN diodes. one at each end of a quarter wave section of transmission line. Typically, a shifter uses two parallel sets of diodes on each quarter wave section. Thus, two bits of phase shift are obtained by one section of line. With a receiving system as shown in FIG. 3, various amounts of phase shift can be applied to a signal received by the antenna element 28.
For a low frequency communication system. a transmit building block for one radiating element is shown in FIG. 4 and includes an amplifier 38 coupled to a 4-bit phase shifter 39. An antenna element 21 is coupled to the amplifier 38 and a control network 41 connects to the phase shifter 39. A control code from the beam steering computer 27 is coupled to the control network 41 through the manifold 22. The phase shifter 39 receives a signal through the manifold 22 from the signal processor 24. The amplifier 38 and the phase shifter 39 are again microstrip circuits with discrete components on a ceramic substrate as described with respect to the receive building block of FIG. 3.
While the receive and transmit building blocks of a low frequency system are relatively simple, including only an amplifier and a phase shifter, the same building blocks for an X-band communication system requires additional components as shown in FIG. 5. An antenna element 28 receives a circularly polarized wave of about 8.3 GHZ and converts it into the input of an unblanced strip line 42 leading to a mixer 43. Insofar as possible, the antenna element 28 rejects a transmit signal emitting from the antenna element 21 to provide a limited amount of isolation. As discussed previously, a crossed dipole is one of many configurations for the antenna element 28 and provides good isolation between transmit and receive signals. Typically, the element 28 will be mounted on a l X 1 inch block face followed by a tapered coax balun.
Conversion of the RF received signal into an IF processing signal takes place in the mixer 43 by means of a signal from a multiplier 44 connected to the output ofa 2.235 MHz local oscillator through a receive manifold 46. The mixer 43 utilizes a Schottky-barrier, GaAs diode of the type described in the U.S. Pat. No. 3,388,000, issued to Warren P. Waters. Metal semiconductor diodes, which are known in the art as Schottky barriers, are commonly used in high frequency circuits such as, for example, the mixer 43. The Schottky barrier of the above United States Patent is fabricated on a semiconductor substrate to conform to the total integrated circuit technique of a modular electronics communication system. An example of an X-band mixer circuit is shown in FIG. 6; it is a thin film circuit using metallization on a ceramic substrate 50. The Schottky barrier mixer diodes 47 and 48 are mounted as chips on either side of an open one quarter wavelength stub 49 providing a short circuit at the input signal frequency. A complete description of a microwave integrated circuit mixer is given in the U.S. Pat. No. 3,416,042 issued to Philip R. Thomas, et al.
The fundamental operation of a mixer 43 is to convert a microwave frequency of a lower frequency with a minimum of added noise. The conversion for optimum operation should be with minimum loss. Generally speaking, the received microwave signal and a signal from a local oscillator are applied to a semiconductor junction from which the difference in frequency or an IF output is extracted. To optimize the noise level for the receiver system, both the signal to noise ratio of the mixer and the conversion loss in the mixer must be as low as possible.
The frequency multiplier 44 can be of a varactor type with a single idler circuit. In the design of an integrated circuit 3X multiplier, careful attention must be given to harmonic noise problems arising from spurious emission from the multiplier. For example, if the 3X multiplier produces a 4X component, the 4X component would mix with the transmitted signal to produce noise ,at the IF frequency. However, a 3X multiplier circuit using varactor diodes should not directly contain a fourth harmonic if designed with a single idler circuit at twice the input frequency.
Output signals at an IF frequency appear on channel 51 and are applied from the mixer 43 to an IF filter and preamplifier 52. The IF preamplifier is a three or four stage hybrid structure to provide, for example, a 30-dB gain with a 6dB noise figure. Filtering of the IF signal on channel 51 is required at the input of the preamplifier to reduce amplifier loading due to the noise outside the IF band. The principal noise source will be the transmit signal which is converted by the mixer 43. Since this frequency is about one and one-half octaves below the IF signal, a multipole band pass microstrip filter should achieve at least 40-dB of attenuation. A completely monolithic IF preamplifier circuit can be constructed using thin film resistors, capacitors, and inductors and epitaxial transistors on a high resistivity silicon substrate.
FIG. 7 illustrates an integrated circuit 53 of a construction suitable for the IF amplifier 52 of FIG. 5. The circuit 53 is comprised of a substrate 54 of single crystal, high resistivity silicon or other semi-insulating or high resistance semiconductor material having first and second surfaces 56 and 57. The resistance required between the surfaces 56 and 57 will vary with the frequency at which the circuit is operated, the lower the frequency the greater the resistance required. However, for high frequency applications, high resistivity semiconductor material is adequate. The components for the IF amplifier are formed at the surface of the semiconductor substrate 54 using any conventional technique. For example, in addition to epitaxial techniques, a transistor 58 may be formed in the surface by sequentially diffusing N-type, P-type, and N-type regions into the surface 57 of the substrate through openings etched in an oxide film 59. The circuit may also include interconnecting strip conductors such as 61, 62, and 63 which may be placed directly on the high resistivity substrate 54 or on the oxide film 59. The conductors may also form inductors such as indicated by the dotted outline at 64.
An insulating layer 66, such as glass, is deposited over and inherently bonded to the portion of the second surface of the substrate 54 which is exposed, and to the components of the circuit. The insulating layer 66 is therefore integral with the substrate. Metallized films 67 and 68 are inherently bonded to the insulating layer 66 and to the first side 56 of the substrate. When the metallized film 67 and 68 are connected to ground, as represented by the conductors 7.1 and 72, the entire integrated circuit is disposed between two closely spaced ground planes. The ground planes and the circuit are interconnected so as to provide a rugged, sealed package. For high frequency transmission lines, the dielectric properties between the circuit components and each of the ground planes may be made approximately equal for improved performance.
A phase shift network 73 receives the amplified IF signal by way of a channel 74 and delivers output signals of IF frequency by way of channel 76 to the receive manifold 46. A beam steering or phase control voltage is applied to a control network 77 from the manifold 46 by way of channel 78 and produces a 3-bit control signal to the phase shifter 73.
The control network 77 is a register which accepts a one-out-of-eight phase step command through the manifold 46 and stores it until another command is received. The register is composed of three flip-flops and an input logic gate, all in low power integrated circuit form, similar to Texas Instruments Series 54L Logic Modules.
The output signal of the central register of the network 77 drives the phase shifter 73, which may be implemented as 3-series shunt-loaded microstrip transmission line quarter wave sections. Phase shifting is obtained by switching two PIN diodes, one at each end of a quarter wave section transmission line. A PIN diode is primarily capacitive under reverse bias and resistive under forward bias and provides a low power loss phase shifting mechanism. The three phase shift sections provide phase shifts in 45-degree steps from to 360 degrees. That is, a 45 -degree section, a 90-degree section, and a l80-degree section is provided in the phase shift network 73.
Output signals at an IF frequency appear on channel 76 and are applied through the manifold 46 for processing in a central processor 79 by means of a channal 81. A distributive manifold for a modular microwave system made up a large number of building block modules, such as the one shown coupled to the manifold 46 and including the radiating element 28, is described in the US. Pat. No. 3,438,029 issued to Troy D. Fuchser et al. As described in the referenced patent, the manifold 46 is a submanifold of a complete distributive manifold system. In addition to the module associated with the radiating element 28, as shown in FIG. 5, the manifold 46 would be coupled to three other radiating elements of the antenna array 12 of FIG. 1. One quarter of the total number of said submanifolds are coupled to a main manifold which in turn are coupled to a fourway divider. In FIG. 5, the channel 81 is intended to represent the main manifold and four-way divider system coupling the radiating element 28 to the central processor 79.
For the transmit section of a modular electronics communication system, an IF frequency signal is trnasmitted from the central processor 79 through a transmit manifold 82 by means of a channel 83 to a phase shift network 84 by means of a channel 86. Again, the channel 83 is intended to represent a complete manifold system for dividing a transmit signal into a plurality of transmit signals to each radiating element of the antenna array 11. A beam steering or phase control code is applied to a control network 87 from the manifold 82 by way of a channel 88 and supplies a 3-bit control signal to the phase shift network 84. The phase shift network 84 and the control network 87 are similar in construction and operation to the corresponding phase shift network 73 and control network 77 of the received module as described previously.
An IF signal from the phase shift network 84 is coupled to an IF amplifier 89 by means of a channel 91. The IF amplifier 89 is similar to the preamplifier 52 described with reference to FIG. 7, except that it will be designed to operate at higher power levels.
The upward frequency conversion process performed on the output of the IF amplifier 89 is basically the same as the downward shifting done in the mixer 43, except that a lower side band up-converter is employed for the actual translation process. A 4X multiplier 92 supplies the carrier frequency to an upconverter 93. The 4X multiplier 92 is similar in construction to the 3X varactor type multiplier 44 in the receive module.
A 4X multiplier circuit is shown in FIG. 8 and employs a varactor diode 94 operating as a quadrupler with idlers at second and third harmonics. More particularly a tuned circuit 96 and 97 may be considered to be resonant at the second harmonic and the tuned circuit 98 and 94 at the third harmonic. The varactor diode 94 and the strip line transmission circuits forming inductance and capacitance are formed on a semiconductor substrate. The substrate 101 has about one-half of its area covered by a highly conductive surface layer 102. The layer 102 is then covered by a thin dielectric layer 103 so that layer 102 serves as a common plate for all but two condensors in the multiplier.
The input L section is formed by the strip transmission line 104 which extends over the thick dielectric portion of the substrate 101 to the plate 106 of the input capacitor. The capacitor 106 overlays the relatively thin dielectric layer 103 to form a condensor with the common conductive layer 102. A loop 107 forms an inductance over the thick dielectric layer and leads to a capacitor plate 108 over the thin dielectric layer. Similarly, a loop 109 leads to the capacitor plate 111. A transmission line filter system will thus be character ized by long thin transmission lines over a thick dielectric section to provide primarily inductance characteristics. Wide transmission line sections overlaying thin high dielectric layers form zones in the transmission line system primarily capacitive in nature. A loop 112 extends from plate 111 over the thick dielectric to the juncture with a loop 96 which leads to a capacitor plate 97. Loop 112 also leads to one terminal of the varactor diode 94. A strip extending from the juncture 98 and loop 116 then leads to a capacitor plate 117. The capacitor plate 117 is positioned on top of a conductive layer 118 which overlays one-half of a condensor plate 119. Condensor plate 124 similarly overlays the plate 1 19. The transmission line loop 121 then extends to the output capacitance plate 122 with the matching conductance 123 extending from the plate 122. The plate 119 is capacitively coupled to the capacitor plates 117 and 124 and to the high conductive layer 102. The process of forming an integrated circuit multiplier is thoroughly described in the US. Pat. No. 3,386,092 of Tom M. I-Iyltin.
The 4X multiplier 92 quadruples the 2.235 GI-lz output of a local oscillator as amplified by a LO amplifier 126 coupled to the transmit manifold 82 by means of a channel 127. The up-converter 93 is a varactor type similar in some respects to the varactor multiplier circuits 44 and 92. It is of the lower side band type rather than the more conventional upper side band type to reduce harmonic noise production in the receiver IF band pass and reduce the factor by which the local oscillator must be multiplied prior to the conversion process. The converted IF signal from the amplifier 89 is then transmitted as an RF frequency signal from the radiating element 21 coupled to the up-converter 93 by means of an unbalanced strip line 128.
By way of example, the operation of the system of FIG. 5 is such that the 8.325 GH: 1 50 MHz circularly polarized wave applied to the antenna element 28 will be converted to a 1.620 0H: t 50 MHz signal in the mixer 43. The frequency conversion function by the 3X multiplier 44 changes the 2.235 GI-Iz local oscillator signal into a 6.705 GHz signal coupled to the mixer 43.
The input power level to the multiplier 44 is 4 mW, to provide the required 2 mW input to the mixer 43. The IF signal of 1.620 GI-Iz 50 MHz is amplified in the IF filter preamplifier 52 having three or more stages of transistor amplification to generate a gain of 30 dB.
Again, by way of example, the transmit section emits a circularly polarized 7.320 GHz 50 MHz signal from the radiating element 21 from the up-converter 93. The up-converter 93 receives a 1.620 GI-Iz t 50 MHz, mW signal from the IF amplifier 89 and an 8.940 GI-Iz, 100 mW signal from the 4X multiplier 92. The L0 amplifier 126 has a gain of dB at 2.235 GHz, with an output power of 250 mW, while the IF amplifier 89 has a gain of 20 dB. Thus, the power generation chain of the transmit section consists of one stage of amplification at 2.235 GI-Iz followed by a 4X varactor multiplier with an output of 8.940 GI-Iz.
As explained previously, phased array antennas such as l 1 and 12 are formed by radiating elements arranged in any desired geometry, for example, triangular spacing. They may be planar or non-planar (conformal). In order to control the phase of the antenna transmit or receive beam, phase control codes are provided for each of the radiating elements in the antenna array. These codes are generated by the beam steering computer 27 such as shown in FIG. 2. Thus, the beam pointing and control in a phased array is accomplished by the phase setting of a phase shift network in series with each radiating element such as network 73 and 84 of FIG. 5. A beam scanning computer calculates the necessary phase for each element in order to collimate the beam at some position in space. The scan computer can be a digital device and the variable phase shift networks either a digital device or a digitally controlled analog device. Because of their digital nature, phase shift networks may be classified as follows:
Phase Shifter Minimum Discrete Phase Shift l-bit 180 Z-bit 90 3-bit 45 4-bit 22.51 5-bit l 12 Thus, the larger number of bits available to control a phase shifter, the more accurately the array beam may be positioned. It is believed that a 3-bit phase shift network will provide sufficient accuracy for beam steering for the system of this invention.
Referring to FIG. 9, there is illustrated the relationship between beam pointing angle and the required phase shift network setting which is given by:
Modern electronically scanned arrays using digital means for controlling the phase of each antenna element frequently use binary devices, such as diodes, for the actual phase control. The two state characteristic of these devices naturally leads to binary phase quantization. In FIG. 9, the actual quantized phase front for each of the seven elements shown is given by the stairstep curve 131. For a control network such as 77 and 87 of FIG. 5, the minimum step size, Q, for the curve 131 is given by:
Q =21r/2' radians where n the number of control bits. Increasing the number of control bits reduces the quantization error at the expense of increased complexity, insertion loss, driving power, cost, and weight of the overall system. Therefore, there is usually some tradeoff between the accuracy with which the array beam may be positioned and the quantization error.
Radar phased array antenna beam steering, which is similar in many respects to the beam steering required in the system of this invention, is adequately described in the literature. For example, the work of Merrill I. Skolink entitled Introduction to Radar Systems, McGraw-Hill, contains a section on phased array antennas and beam steering. A phased array antenna scan control system is also described in the U.S. Pat. No. 3,345,631 issued to Leo A. Chamberlain, .lr.
Referring to FIG. 10, there is shown a duplexed operation employing a single antenna 132 for the received and transmitted signals which are separated by a circulator 133, sometimes known as a diplexer. The circulator 133 is s three-port device that has the property that a wave incident in port one is coupled into port two only, a wave incident in port two :is coupled into port three only, and so on. Ideal circulators are matched de vices, that is, all ports except one terminate in matched loads, the input impedance of the remaining port is equal to the characteristic impedance of its input line, and hence presents a matched load. 11. J. Carlin in an article entitled Principles of Gyrator Networks, Polytech Institute Brooklyn, Vol. 4, 1955, describes how a lossless, matched, nonreciprocal three-port microwave junction as an ideal three port circulator. A practical realization of a three-port circulator involves a symmetrical Y-junction of three identical strip-line type transmission lines with an axial magnetized ferrite rod or disk at the center of the Y-junctzion. Thus, if an 8.3 GHz signal transmitted to port one of the circulator 133 is incident in one leg of the Y-junction, it is coupled to the antenna element 132 through only the second leg of the junction. The 7.3 GHZ signal received by the antenna element 132 will be incident in the second leg of the Y-junction and coupled to only the third leg. Typical characteristics that can be obtained from a circulator are insertion loss of less than 1 dB, and isolation between the transmit and receive signal of from 30 to 40 dB.
Additional separation between the 8.3 GHz, 500 mW transmit signal output of an up-converter 134 from the received signal is performed by a notch filter 136 coupled to the circulator 133 by means: of a channel 137. The notch filter 136 is of a microstrip configuration providing on the order of to dB isolation between the transmit signal and the 7.3 GHz receive signal. A conversion of the 7.3 GHz signal from the filter 136 to a 1.6 GHZ signal of IF frequency is performed in a mixer 138 coupled to receive a 5.7 GI-Iz, 2 mW sig nal from a 3X multiplier 139. The 3X multiplier 139 receives a 1.9 GHZ local oscillator signal on channel 141 through a manifold 142.
Additional filtering of the IF signal from the mixer 138 is performed in a notch filter 143 coupled to an IF filter and preamplifier 144 by means of a channel 146. A phase shift network 147 receiving control signals over a channel 148 couples the 1.6 GHz signal to a central processor (not shown) through the manifold 142.
The transmit channel of the system of FIG. is similar to the transmit channel of the system of FIG. 5. In addition to the up-converter 134, it includes a local oscillator amplifier 149 for power amplification of a 2.235 GI-Iz local oscillator signal and a 3X multiplier 151 for changing the local oscillator signal into a 6.7 GHZ signal connected to the up-converter 134. A phase shift network 152 applies the appropriate phasing information to a l.6 GI-Iz signal on channel 153 in accordance with control information on channel 154. An IF amplifier 156 couples the transmit signal from the phase shifter network 152 to the up-converter 134 wherein a conversion process takes place to the 8.3 GHZ transmit signal.
Again, a module containing the components shown in FIG. 10 is required for each radiating element of the antenna array 132. The operation and construction of the various components of FIG. 10 have been described previously with respect to FIG. 5, except for the I notch filters 136 and 143, and the diplexer 133.
Referring to FIG. 11, there is shown another embodiment of a duplexed operation employing a single antenna 160 for the received and transmitted signals. To obtain acceptable signal separation, different transmit and receive frequencies are employed as in the previously described systems. In addition, about 10 dB signal separation is obtained by circularly polarizing the transmit received signal and an attenuated transmit signal. This output is one sense and circularly polarizing the receive signal with the opposite sense. For additional signal separation, a filter 162 is designed to pass the frequency of the received signal and attenuate the frequency of a transmittedsignal. For example, the filter 162 may be designed to attenuate a transmitted frequency between 7.725 to 7.850 GI-Iz by approximately 40 dB and pass a 7.125 GHz signal received by the antenna 160 with a loss of about ldB. Thus, the output of the filter 162 will consist of both the output is processed through a receive channel similar to that described with respect to FIG. 10.
The output of the filter 162 is converted into an IF frequency by a mixer 164 which also receives a signal from a multiplier 166 connected to a phase shift network 168. The phase shift network 168 is similar to that described previously. In the embodiment of FIG. 11, the phase of the received signal is controlled substantially simultaneously with the conversion of the RF signal received at the antenna 160 into an IF processing signal by varying the phase of the local oscillator signal multiplication of this signal in the 3X multiplier 166 produces a mixer input signal with the desired phase shift. The input to the preamplifier 172 is the output of a local oscillator (not shown). It should be noted that if preamplifier 172 is capable of providing the desired local oscillator frequency via phase shifter 168 to mixer 164, the 3X multiplier 166 would be unnecessary and the actual desired amount of phase shift for the signal received at the antenna would have to be provided by phase shifter I68.
Typically, the local oscillator may have an output of approximately 2.208 GHz at 2mw. This signal is amplified in the amplifier 172 to a power level of 20 mw. As a result of passing through the phase shift network 168, the power level at the input of the multiplier 166 is 10 mw. In the multiplier 166, the 2.208 GHz oscillator output is increased in frequency to 6.625 GI-lz at a power level of 4mw. Considering a received signal of 7.125 GI-Iz, the output of the mixer 164 will be an IF signal at 500 MHz 2 62.5 Gl-Iz plus some level of unwanted transmitter component at 1 162.5:625 MHz.
Additional filtering of the IF signal from the mixer 164 is carried out in an IF filter 174 coupled to an IF amplifier 176, reducing the unwanted transmitter component another 50 dB. Output signals at an IF frequency from the amplifier 176 are applied through a manifold (not shown) for processing in a central processor of the type described previously.
The transmit channel of the system of FIG. 11 includes a preamplifier 178 coupled to the output of the signal processor. A phase shift network 180 applies the appropriate phasing information to the output of the amplifier 178 in accordance with control information coupled to a 4-bit logic network 182. Instructions to the 4-bit logic network 182 are received from the central processor. The output of the preamplifier 178, with the appropriate phase shift information applied thereto, is connected to the input of a power amplifier 184 which in turn connects to a multiplier 186. The multiplier 186 performs the same function as the upconverter described previously. It converts the low frequency output of the central processor into a high frequency transmit signal.
Considering that the transmitted frequency is in the range of from 7.725 to 7.850 GI-Iz, then with a 3X multiplier, the output of the central processor is a signal at a frequency in the range from 2.575 to 2.617 GI-Iz. Typical power levels for a system of the type illustrated are 2mw to the amplifier 178 and 20 mw to the phase shift network 180. The power level of the signal transmitted from the phase shift network 180 to the power amplifier 184 is on the order of 10 mw with the output of the power amplifier at a 250 mw level. It should be understood, that the values discussed with regard to FIG. 11 are given only by way of example.
In addition to the use of modular electronics technology to satellite communcation systems, there are several other areas for its application; the most appealing of which is the implementation of skin-mounted conformal arrays for aircraft. A modular electronic communication system, capable of high antenna gains and coupled to a beam steering compunter, allows relative small tactical aircraft to communicate freely through a satellite system. Presently, only large, equipment filled aircraft are capable of communicating through such a satellite system. Possibilities exist in this context for the implementation of a combined navigation and communication system employing exclusively satellite terminals. The modular electronics array also is compatible with implementation as an electronically adaptive system or a retrodirective system which permits automatic tracking of satellites from aircraft terminals.
The frequency flexibility of a modular electronics communication system with its high speed, inertialess beam steering also lends itself for adaptation to other applications. For example, spacecraft systems with a high gain, steerable antenna are envisioned as becoming important as data rates and communication distances increase. These same characterisitcs, combined with the reliability and relative ruggedness of an integrated circuit system, make it equally suitable for portable mobile or shipboard applications.
Another area contemplated as within the scope of the present invention includes a system for generating several radiation patterns from one antenna array simultaneously. This, for example, permits the simultaneous, narrow-beam tracking of several satellites by one earth based array. A multibeam function may be achieved in several ways. The simplest of which is to use designated sections of the antenna array for each beam. Another approach would be to use multiple phase shifters and manifolds in the transmit section in a manner similar to the receive section of FIG. 3. With this approach, the entire antenna array is used for each transmit beam. The phase shift networks for each group of radiating elements be independently controlled by the beam steering computer.
While several embodiments of the invention, together with modifications thereof, have been described in detail herein and shown in the accompanying drawings, it will be evident that various further modifications are possible in the arrangement and construction of its components without departing from the scope of the invention.
What is claimed is:
l. A communication system having a central processor for generating a transmitted signal and processing a received signal comprising:
an antenna array of a plurality of transmitting/receiving elements, and transmit/receive module for each element of said antenna array, said module including phase shift means for shifting the phase of a transmitted signal generated by said central processor independent of signals received at the antenna array, power amplification means for increasing the power of said transmitted signal, frequency converting means for converting a low frequency signal to a high frequency transmit signal, a transmit channel coupling said phase shift means and said power amplification means to said frequency converting means to convey phase shifted transmitted signals of increased power to said element, said module also including phase shift means for selecting the phase of the received signal processed by the central processor independent of the transmitted signal, a filter for separating a received signal frequency from other signals, and a receive channel being coupled to said filter and said phase shift means for conveying a received signal to said central processor, said receive channel including a mixer coupled to said phase shift means.
2. A communication system as set forth in claim 1 wherein said transmit/receive module includes in the receive portion a multiplier to change the frequency of the output of the phase shift means prior to coupling to said mixer.
3. A communication system as set forth in claim 2 wherein said transmit/receive module includes in said receive channel an IF filter-amplifier coupled to said mixer.
4. A communication system as set forth in claim 1 wherein said frequency converting means is a multiplier.
* I: i t
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|U.S. Classification||342/371, 342/377|
|International Classification||H01Q3/34, H01Q21/00, H01Q3/38, H01Q3/30|
|Cooperative Classification||H01Q21/0025, H01Q3/34, H01Q3/38|
|European Classification||H01Q3/34, H01Q21/00D3, H01Q3/38|