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Publication numberUS3851244 A
Publication typeGrant
Publication dateNov 26, 1974
Filing dateDec 18, 1973
Priority dateDec 18, 1973
Publication numberUS 3851244 A, US 3851244A, US-A-3851244, US3851244 A, US3851244A
InventorsMounce G
Original AssigneeElectronic Ass Of Canada Ltd
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Microwave moisture measuring apparatus
US 3851244 A
Abstract
Microwave moisture measuring apparatus wherein the output of a variable frequency microwave source is beamed by lens-corrected horns through a moisture containing material and then delayed in time and finally heterodyned with the original output of the microwave source to produce a predetermined beat frequency which can be amplified and then linearly detected.
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Description  (OCR text may contain errors)

MICROWAVE MOISTURE MEASURING APPARATUS George R. Mounce, Willowdale, Ontario, Canada Assignee: Electronic Associates of Canada,

Ltd., Downsview, Ontario, Canada Filed: Dec. 18, 1973 Appl. No.: 425,937

Inventor:

US. Cl 324/585 A int. Cl G0lr 27/04 Field of Search 324/58.5 A, 58 A, 57 H References Cited UNITED STATES PATENTS 8/1966 Heald 324/585 A Nov. 26, 1974 3,50l,692 3/l970 Kluck 324/585 A 3,534,260 10/1970 Walker 3,693,079 9/1972 Walker 324/585 A Primary Examiner-Stanley T. Krawczewicz Attorney, Agent, or Firm-Shenier & OConnor [57] ABSTRACT Microwave moisture measuring apparatus wherein the output of a variable frequency microwave source is beamed by lens-corrected horns through a moisture containing material and then delayed in time and tinally heterodyned with the original output of the microwave source to produce a predetermined beat frequency which can be amplified and then linearly detected.

28 Claims, 9 Drawing Figures 22-Z3GH IOOH BACKGROUND OF THE INVENTION In microwave systems for measuring moisture in sheet materials such as paper and the like, free water absorbs microwave energy due to molecular resonance. The greatest attenuation of microwave energy occurs when the exciting frequency corresponds to the natural resonant frequency of the molecules. One such absorption peak occurs in the band from to GHz. One system of the prior art employs a microwave source operating in this frequency band which is coupled to a first horn antenna on one side of the paper web. A second horn antenna on the other side of the paper web is aligned with the first antenna and receives energy radiated from the first antenna. The output of the receiving antenna is applied to a crystal detector. Since the energy radiated from the first antenna passes through the paper web, it will be attenuated as a function of the moisture content of the web. The transmitting and receiving horns are usually directed at right angles to the web.

Such system suffers three defects. The first defect is that it is impossible to prevent some microwave energy from being reflected back to the transmitting antenna either from the receiving antenna or from the paper web. Such reflections will change the apparent power radiated from the transmitting antenna, depending upon the phase of the reflected signals which will vary with any change in the spacing between the two antennas, with any variations in the position of the paper web relative to the two antennas, and with any change in frequency. Since it is impossible to maintain an absolutely constant spacing between all parts of the system, the apparent power radiated from the transmitting antenna will vary because of the changing phase angle of the reflections. This changes the output from the crystal detector even though the moisture content of the paper web may be constant.

The second defect is that a conventional transmitting horn antenna produces a divergent beam having a convex wavefront, while a conventional receiving horn antenna responds best to a convergent beam having a concave wavefront. This inherent mismatch in the curvature of the two wavefronts greatly increases the amplitude of reflections between the two horns.

The third defect is that crystal detectors used as straight rectifiers have a very limited sensitivity, since they are operated in a square-law region because of the relatively low power output of the microwave source. For the levels of moisture commonly encountered in wet or damp paper webs, the attenuation is so great that the output of the detector is extremely small. Moreover, operation of detectors in the square-law region at low power levels results in a high incremental impedance, producing correspondingly large noise voltages which further limit the minimum detectable signal.

In order to reduce the effect of reflections upon the output of the detector, prior art systems have employed a frequency modulated microwave source. The extend of frequency modulation is such as to change the phase of reflections by 360 or any integral multiple thereof. The energy output from the receiving horn will thus pass through an integral number of cycles of variation;

and if the variation is small, the average output of the detector will accurately represent the moisture content of the damp paper web. However, in most cases the amplitude of reflections is not small compared with the average amplitude. Since the detector is operated in a square-law region, its average output in the presence of strong reflections will always be greater than the average amplitude of signals collected by the receiving horn. Accordingly, the simple frequency modulation systems of the prior art produce excessive outputs from the detector in the presence of strong reflections.

SUMMARY OF THE INVENTION My invention contemplates the provision of a heterodyne or beat freqnency system wherein the detector is subjected both to strong signals from the frequency modulated microwave source and to delayed and attenuated signals from the receivng horn antenna. The strong signal from the microwave source acts as a local oscillator on the detector and forces it out of the highimpedance, high-noise, square-law region into a lowimpedance, low-noise, linear region. This provides high sensitivity irrespective of the extent of attenuation in the damp paper web. Furthermore the adverse effect of strong reflections is substantially eliminated, since linear detection insures that the average output of the detector will accurately represent the average amplitude of signals at the receiving horn over an integral number of cycles. My invention further contemplates the provision of dielectric converging lenses for the transmitting and receiving horns so that the compensated horns will produce and respond best to parallel beams having planar wavefronts. This greatly improves the match between the two homs and correspondingly reduces the amplitude of reflections.

One object of my invention is to provide microwave moisture measuring apparatus employing a heterodyne or beat frequency detector.

Another object of my invention is to provide microwave moisture measuring apparatus employing linear detection.

a further object of my invention is to provide microwave moisture measuring apparatus in which the adverse effects of reflections are substantially eliminated.

A still further object of my invention is to provide microwave moisture measuring apparatus in which mismatch reflections between horns are greatly reduced by dielectric lenses.

Other and further objects of my invention will appear from the following description.

BRIEF DESCRIPTION OF THE DRAWINGS In the accompanying drawings which form part of the instant specification and which are to be read in conjunction therewith and in which like reference numerals are used to indicate like parts in various views:

FIG. 1 is a schematic view showing a first embodiment of my invention employing two pairs of horn antennas. I

FIG. 2 is a graph showing the variation in frequency of the microwave source as a function of time.

FIG. 2a is a graph showing the corresponding variation in the output of the detector as a function of time.

FIG. 3 is a graph on an enlarged scale showing variations in the detector output due to both strong and weak reflections.

FIG. 4 is a schematic view illustrating a second embodiment of my invention employing only a single pair of horn antennas.

FIG. 5 shows the design details of a converging dielectric lens having two refracting surfaces.

FIG. 6 shows the design details of a converging dielectric lens having a single refracting surface.

FIG. 7 is a front view of a stepped approximation to a double refraction lens which gives good results in practice.

FIG. 8 is a side view of the lens of FIG. 7.

DESCRIPTION OF THE PREFERRED EMBODIMENTS Referring now more particularly to FIG. 1 of the drawings, my system may be used for measuring the moisture content of a horizontally disposed damp paper web 8. A 100 Hz oscillator 10 provides an output waveform of trapezoidal shape. The flat top and bottom of the trapezoid may each have a duration of 2.5 milliseconds; and the rising and falling ramps may each have a duration of 2.5 msec. Oscillator 10 controls the frequency of a variable frequency microwave source 12 such as a backward wave oscillator. The frequency of BWO 12 is varied by oscillator 10 between 22 GHz and 23 GHz which lies in the middle of the absorption band of the water molecule. Energy from oscillator 12 is coupled through a microwave isloator 14 to the primary input port of a directional coupler 16. The primary output port of directional coupler 16 is coupled to an upwardly directed transmitting horn antenna 20 which is positioned below web 8. Positioned above web 8 and aligned with the axis of horn 20 is a receiving horn antenna 22 which is coupled to a delay line 24. The output of delay line 24 is coupled to a downwardly directed second transmitting horn antenna 26 which is positioned above web 8. Positioned below web 8 and aligned with horn 26 is a second receiving horn antenna 28. Horn 28 is coupled through a microwave isolator 30 to a secondary input port of directional coupler 16. The secondary output port of directional coupler 16 is coupled to a linear diode detector 32. The output of detector 32 is coupled to an amplifier 34 the output of which is applied to a band-pass filter 36. Filter 36 may have a sharp lower cutoff at 13 KHz and a sharp upper cutoff at 19 KHz. The output of filter 36 is coupled to a linear diode detector 38, the output of which is applied to a low-pass filter 40 which may have a sharp cutoff at 180 Hz. I-Iorns 20, 22, 26, and 28 are mounted on a U-shaped member 9 which may be moved left and right to scan across the width of web 8. At the mouth of horn 20 is moutned a converging dielectric lens 20a. Lens 20a may, for example, be of the single refraction type having a plane outer surface and a curved inner surface. Converging lenses 22a, 26a, and 280 are similarly mounted at the mouths of respective horns 22, 26, and 28. The beams from lens-corrected horns 20a and 26a are thus parallel with planar wavefronts.

In operation of the embodiment of FIG. 1, isolator 14 prevents reflected energy from being applied backwardly to oscillator 12 so that its stability will not be impaired. Most of the energy applied at the primary input port of directional coupler 16 passes to the primary output port thereof; but a certain portion, de-

pending upon the coupling factor, appears at the secondary output port and is applied to heterodyne detector 32. The energy at the primary output port of directional coupler 16 is radiated by horn 20 and passes through web 8 to receiving horn 22. The energy received by horn 22 is less than that transmitted by horn 20 because of attenuation or absorption caused by the water molecules in web 8. The energy from receiving horn 22, after delay in line 24, is radiated from horn 26 and passes downwardly through web 8 to receiving horn 28. The energy received by horn 28 is less than that transmitted by horn 26 because of attenuation or absorption in web 8. The purpose of isolator 30 is to prevent passage of energy backwardly from coupler [6 to horn 28, while permitting energy from horn 28 to pass essentially unattenuated to the secondary input port of directional coupler 16. Most of the energy at the secondary input port of directional coupler 16 passes to the secondary output port thereof and thence to heterodyne detector 32.

Detector 32 thus receives two signals. A first signal is that appearing at the primary input port of directional coupler 16. This first signal has a large amplitude and may be considered the equivalent of a local oscillator signal. The second signal is that appearing at the secondary input port of directional coupler 16. this second signal is greatly attenuated since it has passed through web 8 twice. This second signal is also delayed in that it has passed through coupler 16, horn 20, horn 22, delay line 24, horn 26, horn 28 and isolator 30.

Assume that the overall time delay for the passage of the signal from the primary input port of directional coupler 16 to the secondary input port of directional coupler 16 is 0.04 microsecond. During the flat top and bottom portions of the trapezoidal modulating waveform from oscillator 10, the signals at the primary and secondary input ports of directional coupler 16 will have the same frequency, that is, either 22 GHz or 23 GHz. Hence during these periods the two signals applied to detector 32 will produce a difference or beat frequency of zero. During the period when the output of oscillator 10 is rising and, for example, the frequency of oscillator 12 is increasing from 22 GHz to 23 GHz, the frequency at the secondary input port of directional coupler 16 will be less than that at the primary input port thereof. Correspondingly during the period when the output of oscillator 10 is falling and the frequency of oscillator 12 is decreasing from 23 GI-Iz to 22 GHZ, the frequency at the secondary input port of directional coupler 16 will be greater than that at the primary input port thereof. It may be shown that the difference frequency between the signals at the primary and secondary input ports of directional coupler 16 is equal to the product of the rate of change of frequency of microwave oscillator 12 and the overall time delay of .04 usec) for energy to pass from the primary input port of directional coupler 16 to the secondary input port thereof. For both the positive and negative ramps of the trapezoidal output of oscillator 10, the rate of change of frequency of oscillator 12 is 1 GHz per 2.5 msec or 4( I0) I-Iz/sec. Accordingly the difference in frequencies of signals at the input ports of directional coupler 16 during the rising and falling ramps of the trapezoidal output of oscillator 10 is 4(l0)(0.04)(l0) 16 KHz. This is the center frequency of band-pass filter 36.

Referring now to FIG. 2, there is shown the 100 Hz waveform of oscillator which controls the frequency modulation of oscillator 12. FIG. 2a shows the corresponding output from detector 38. It will be noted that the output from detector 38 comprises a 200 Hz square wave. During those periods when the output from oscillator 110 is constant, the difference frequency output from detector 32 is zero; and the output from detector 38 is also zero. During those periods when the output of oscillator 18 is changing, detector 32 provides a difference frequency output of 16 KHz which lies in the pass band of filter 36; and detector 38 may provide, for example, a positive output. It will be noted that the output from detector 38 is not a perfect square wave sincesults in short duration rising and decaying transients in the output of detector 38.

While oscillator 10 may provide a pure triangular waveform, a clipped triangular or trapezoidal waveform with flat top and bottom portions is preferred, since this produces extended periods of time for which the output of detector 32 is zero. This permits energy stored in the reactive elements of filter 36 to completely dissipate so that each burst of 16 KHz energy from heterodyne detector 32 can be treated as a separate entity by band-pass filter 36. If the output of oscillator 10 were a pure and unclipped triangular waveform with no flat dwell portions during which the energy stored in filter 36 could decay, then the output of detector 38 would be dependent upon the phase shift between successive 16 KHz bursts from detector 32. If there were no phase shift between successive bursts from detector 32, then the output from detector 38 would be a constant direct-current. If, however, successive 16 KHz bursts from detector 32 had any relative phase shift other than 0, then the output of detector 38 would momentarily decrease and then rise again to its original value. The worst case would occur when the phase shift between successive KHz bursts from detector 32 is 180. In this case the output of detector 38 would momentarily drop to zero and then rise again to its original value.

It is desired that low-pass filter 40 integrate over one full cycle of the 200 Hz square wave output from detector 38. Accordingly filter 40 should have a cutoff frequency less than 200 Hz, as for example 180 Hz, in order to eliminate 200 Hz and higher harmonic ripple in the direct-current output.

Assume that the time delay for passage of signals from the throat of horn to the throat of horn 22 is 0.002 usec. For any mismatch between horns 20 and 22, this same amount of time will be required for reflections from the throat of horn 22 to be returned to the throat of horn 20. The total time delay for reflections back and forth between the throats of horns 28 and 22 will be 0.004 usec. Similarly the total time delay for reflections back and forth between the throats of horns 26 and 28 will also be 0.004 usec. During those periods when the frequency of oscillator 12 is changing, horns 22 and 28 receive two signals of different frequencies, due to horn mismatch reflections, which are coupled to the secondary input port of directional coupler 16. The frequency difference between these two signals is equal to the product of the rate of change of frequency of microwave oscillator 12 and the total time delay for reflected signals to pass back and forth between the throats of a pair of opposed horn antennas. Thus the frequency difference between the two signals applied to the secondary input port of directional coupler 16 is 4(10)(0.004) (l0) 1,600Hz.

As may be seen by reference to F IG. 3, during the period when the frequency of oscillator 12 is changing, the output of detector 38 will exhibit a 1,600 l-Iz variation about its average pulse amplitude. During each 2.5 msec period when the frequency of oscillator 12 is changing, there will occur 2.5( 10)'( 1,600) 4 cycles of 1,600 Hz variation due to reflections. It is desired that there be an integral number of cycles of variation so that the total area under each pulse will be constant irrespective of the magnitude of reflections and hence of the amplitude of the 1,600 Hz variation. When dielectric lenses are used, the 1,600 Hz variation is substantially sinusoidal as shown greatly exaggerated in scale by curve 38 of FIG. 3. However, when the dielectric lenses are removed, there results a 1,600 Hz variation of substantially sawtooth waveform as shown approximately to scale by curve 38a of FIG. 3. The variation in waveform 38a may be expressed approximately sin 6+%sin20+/asin 30,

where 0 27r( 1,600) r radians. Waveform 38a includes not only a 1,600 Hz fundamental component but also appreciable 3,200 Hz and 4,800 I-Iz second and third harmonic components. These harmonic components result because the horn mismatch is so severe that two and three round-trip reflections between horns 20 and 22 still result in an appreciable amplitude of the reflected signal. The use of lens-corrected horns greatly reduces the amplitude of all mismatch reflections. The attenuation of the second and third harmonic components is so great that they are substantially suppressed; and only the fundamental component is discernible.

The system of FIG. 1 requires appreciable space because of the use of four horn antennas. Furthermore it does not measure the moisture in a single area but instead measures the average moisture in two distinct areas, which somewhat limits the discriminiation of the measurement. For these reasons it would be advantageous if only a single pair of horn antennas were required.

Referring now to FIG. 4 of the drawings, the primary output port of the directional coupler 16 is connected to the 0 port of a polarization separator 17, having a port and a common port. The common port of polarization separator 17 is coupled to a horn antenna 19 which is positioned below web 8. Horn 19 is directed upwardly, but at an appreciable angle from the vertical. Positioned above web 8 and aligned with the axis of horn I9 is a horn antenna 21 which is coupled to the common port of a polarization separator 23, having a 0 port and a 90 port. The output from the 0 port of polarization separator 23 is coupled to delay line 24. The output of delay line 24 is applied to a section of wave guide 25 which is twisted through 90 thereby to rotate the plane of polarization through 90. The output from the 90 twist section'25 is applied as an input to the 90 port of polarization separator 23. The output from the 90 port of polarization separator 17 is coupled to a section of wave guide 27 which is twisted through 90 thereby to rotate the plane of polarization through 90. The output from 90 twist section 27 is coupled through microwave isolator 30 to the secondary input port of directional coupler 16. I-Iorns 19 and 21 are mounted on a U-shaped member 9 which may be moved left and right to scan across the width of web 8. At the mouth of horn 19 is mounted a dielectric lens 19a; and at the mouth of horn 27 is mounted a lens 210. Lenses 19a and 210 are of the double-refraction type having a curved outer surface and a planar inner surface.

In operation of the embodiment of FIG. 4, microwave enegy from source 12 passes through isolator 14, to the primary input port of directional coupler 16. Most of this energy passes to the primary output port of coupler l6 and thence through polarization separator 17 to horn 19. This energy is radiated from horn l9 and passes through web 8 at an appreciable angle from the normal. For this forward transmission path, horn 19 transmits; horn 21 receives; and the plane of polarization is The energy received by horn 21 passes through polarization separator 23 and delay line 24 to 90 twist section 25. The energy from twist section 25 passes through polarization separator 23 to horn 21. The energy is radiated from horn 21 and passes through web 8 to horn 19. For this return transmission path, horn 21 transmits; horn 19 receives; and the plane of polarization is 90. The energy received by horn 19 passes through polarization separator 17 to 90 twist section 27. Section 27 rotates the plane of polarization back to 0 and the energy passes through isolator 30 to the secondary input port of directional coupler 16. The 90 twist section 27 is provided so that both the primary and secondary input ports of coupler 16 will receive energy with the same plane of polariazation. This insures that the summation of signals at the secondary output port of coupler 16 is governed solely by its coupling factor and is independent of the orientation of detector 32 within the secondary output port of coupler 16.

Horns 19 and 21 as well as the short sections of wave guide joining horn 19 to the common port of separator 17 and joining horn 21 to the common port of separator 23 should have symmetry about the X and Y axes in order to pass with equal facility energy having a polarization of either 0 or 90. Accordingly, horns l9 and 21 may have, for example, either a square or a circular cross section.

In FIG. 4, the 0 polarized radiation from horn l9 passes upwardly through web 8 at an angle of approximately 45 from the normal; and 90 polarized radiation from horn 21 passes downwardly through web 8 at the same angle from the normal. This effectively increases the apparent thickness of web 8 by approximately 41 percent and results in a greater attenuation of microwave energy appearing at the secondary input port of coupler 16. Furthermore any 0 polarized energy from horn 19 which is reflected from web 8 will be directed downwardly and to the left and will not be returned directly to horn 19. Similarly any 90 polarized energy from horn 21 which is reflected from web 8 will be directed upwardly and to the right and will not be returned directly to horn 21. This greatly reduces the coupling of web reflections back to either of horns 19 and 21 and consequently reduces the adverse residual effect of variations in the position of the web relative to the two horns. Because of the greatly increased power gain of lens-corrected horns, the size of the horns may be reduced by 40 percent while at the same time achieving a 4 db increase in overall gain. The reduction in horn size permits a corresponding reduction in horn separation where, as in FIG. 4, the beams pass through the web at an appreciable angle from the normal.

- The 0 polarized energy from horn l9 and the polarized energy from horn 21 both pass through web 8 in the same area. This increases the discrimination of the microwave moisture measurement.

Referring now to FIG. 5, there is shown the design details for a double-refraction lens. In the equations, f is the focal length of the lens which is substantially equal to the length of the horn, d the maximum thickness of the lens which should be an integral number of half wavelengths, and n is the refractive index of the lens material which is equal to the square root of its relative dielectric contant.

FIG. 6 shows the design details for a single-refraction lens. The lenses of FIGS. 5 and 6 are of the planoconvex type. The lens of FIG. 5 has a shape factor of +1, while the lens of FIG. 6 has a shape-factor of -1. If the convex surfaces are limited simply to being spherical, then the lens of FIG. 5 is preferable since it is closest to the shape factor of +0.714 for minimum spherical aberation. However, optical precision is neither needed nor required. As a practical matter, either of the lenses of FIGS. 5 and 6 is more than adequate; and in fact even relatively crude approximations to these lenses yield excellent results in practice.

FIGS. 7 and 8 show a step-wise approximation to a piano-convex lens which gives satisfactory results in practice. Lens 41 comprises a 4.000 inch square base section 42, a 3,455 inch diameter circular intermediate section 43, and a 2.300 inch diameter circular top section 44. Each of sections 42, 43, and 44 may have a thickness of 0.186 inch, so that the maximum lens thickness is 0.558 inch. The lens may be integrally formed of Teflon, which is a brand of tetrafluoroethylene polymer made by E I. du Pont de Nemours and Co. Lens 41 is designed for use with a horn having a square cross section with a four inch mouth and a length of 11 inches.

It will be seen that I have accomplished the objects of my invention. My microwave moisture measuring apparatus employs a heterodyne or beat frequency detector which is subjected both to a strong signal from the frequency modulated source and to a delayed signal which has been attenuated by virtue of its having passed twice through the moisture containing web. The strong signal from the microwave source causes the beat frequency detector to operate in a substantially linear manner. This increases the sensitivity of the detection of the attenuated and delayed signals. The use of lens-corrected horns greatly reduces mismatch reflections and increases the power gain. The adverse effect of residual reflections is substantially eliminated, since linear detection insures that the average output of the detector represents the average amplitude of signals over an integral number of cycles. The effect of residual reflections may be rendered negligible by narrowing the pass band of filter 36 from 6 KI-Iz to ZKHz. Filter 36 should have a sharp lower cutoff at 15 K112 and a sharp upper cutoff at 17 KHz. Thus filter 36 can respond only to modulating frequencies up to 1 KHz or 1,000 I-Iz which is less than the 1,6000 Hz fundamental frequency component of mismatch reflections. The output of detector 38 is now as shown in FIG. 2a with negligible ripple discernible during each positive pulse.

It will be understood that certain features and subcombinations are of utility and may be employed without reference to other features and subcombinations. This is contemplated by and is within the scope of my claims. It will be further understood that various changes may be made in details without departing from the spirit of my invention. It is therefore to be understood that my invention is not to be limited to the specific details shown and described.

Having thus described my invention, what I claim is:

l. Microwave moisture measuring apparatus including in combination a variable frequency source of microwave energy, control means for cyclically varying the frequency of the source, means including a lenscorrected horn for radiating energy from the source through a moisture containing material, said material attenuating the energy passing therethrough as a function of its moisture content, a first detector, and means coupling both the source and the attenuated energy to the first detector.

2. Apparatus as in claim 1 further including a bandpass filter, a second detector, and means including the filter for coupling the first detector to the second detector.

3. Apparatus as in claim 1 wherein the control means cyclically varies the source frequency at a certain repetition frequency, the apparatus further including an amplifier, a band-pass filter, a second detector, means including the amplifier and the band-pass filter for coupling the first detector to the second detector, a lowpass filter, and means coupling the second detector to the low-pass filter.

41. Apparatus as in claim 3 wherein the low-pass filter has a cut-off frequency less than twice said repetition frequency.

5. Apparatus as in claim 1 wherein the construction of the control means is such that the source frequency cyclically varies in accordance with a generally triangular waveform having equal positive and negative slopes.

6. Apparatus as in claim 1 wherein the construction of the control means is such that the source frequency cyclically varies in accordance with a trapezoidal waveform having flat top and bottom portions of appreciable extent. I

7. Apparatus as in claim ll wherein the means coupling the attenuated energy to the first detector includes a delay line.

8. Apparatus as in claim 1 wherein the means coupling the source and the attenuated energy to the first detector includes a directional coupler.

9. Apparatus as in claim 1 wherein the radiating means includes a microwave isolator.

10. Apparatus as in claim 11 wherein the means coupling the attenuated energy to the first detector includes a microwave isolator.

ll. Apparatus as in claim ll wherein the radiating means includes means for passing source energy twice through the material.

12. Apparatus as in claim ll wherein the radiating means includes a first transmitting horn antenna disposed on one side of the material, a first receiving horn antenna disposed on the other side of the material and oriented to receive energy from the first transmitting horn, a second transmitting horn antenna disposed on said other side of the material, means coupling the first receiving horn to the second transmitting horn, and a second receiving horn antenna disposed on said one side of the material and oriented to receive energy from the second transmitting horn.

13. Apparatus as in claim 12 wherein the means coupling the first receiving horn to the second transmitting horn includes a delay line.

14. Apparatus as in claim 1 wherein the radiatingmeans includes a first and a second horn antenna disposed on opposite sides of the material and oriented to transmit and receive energy from one another, a first and a second polarization separator each having a common port and a pair of polarization ports, means coupling the source to one polarization port of the first separator, means coupling the common port of the first separator and the first horn, means coupling the common port of the second separator and the second horn, and means including a first polarization rotator for coupling one polarization port of the second separator and its other polarization port.

15. Apparatus as in claim 14 wherein the means coupling the two polarization ports of the second separator includes a delay line.

16. Apparatus as in claim 14 wherein the means coupling the attenuated energy to the first detector includes the other polarization port of the first separator.

17. Apparatus as in claim 14 wherein the means coupling the attenuated energy to the first detector includes a second 90 polarization rotator and means coupling the other polarization port of the first separator to the second rotator.

18. Apparatus as in claim 14 wherein said material comprises a generally planar web and wherein the horns are so oriented that radiation therebetween passes through the web at an appreciable angle from a normal to its plane.

19. Apparatus as in claim 1 wherein the source operates in the frequency band from 20 GHZ to 25 GHz.

20. Apparatus as in claim 1 wherein the variable frequency source comprises a backward wave oscillator.

21. Apparatus as in claim 1 wherein said material comprises a web and wherein the radiating means comprises a pair of horn antennas disposed on opposite sides of the web and mounted on a U-shaped member adapted to be moved transversely across the width of the web.

22. Apparatus as in claim 1 wherein the construction of the lens-corrected horn is such as to provide a parallel beam having a planar wavefront 23. Apparatus as in claim 1 wherein the means coupling the attenuated energy to the first detector includes a second lens-corrected horn.

24. Apparatus as in claim ll wherein the corrected horn includes a plano-convex converging lens.

25. Apparatus as in claim 1 wherein the corrected horn includes a double refraction converging lens.

26. Apparatus as in claim 1 wherein the corrected horn includes a single refraction converging lens.

27. Apparatus as in claim 1 wherein the corrected horn includes a stepped approximation of a converging lens.

28. Apparatus as in claim 1 wherein the means coupling the attenuated energy to the first detector includes a second horn, the apparatus further including a band-pass filter and means coupling the detector to the filter, the filter having a pass band sufficiently narrow to reject the fundamental component of modulation caused by mismatch reflections between the two horns.

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Referenced by
Citing PatentFiling datePublication dateApplicantTitle
US4220915 *Jun 28, 1978Sep 2, 1980Rca CorporationResistivity measurement system
US4319185 *Dec 5, 1979Mar 9, 1982Sentrol Systems Ltd.Delay line microwave moisture measuring apparatus
US4716360 *Aug 16, 1985Dec 29, 1987Advanced Moisture Technology, Inc.Moisture detector apparatus and method
US4767982 *Jun 1, 1987Aug 30, 1988Master Chemical CorporationConcentration detection system
US4789820 *Jul 11, 1986Dec 6, 1988Hercules IncorporatedApparatus and method for sensing multiple parameters of sheet material
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US4820970 *Jun 3, 1987Apr 11, 1989Swanson Claude VApparatus and method for using microwave radiation to measure water content of a fluid
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US5059914 *Jul 11, 1990Oct 22, 1991Aerospatiale Societe Nationale IndustrielleMicrowave device for the continuous measurement of the viscosity of a viscous medium
US5315258 *Mar 30, 1992May 24, 1994Kajaani Elektroniikka OyMethod and apparatus for determining the moisture content of a material
US5333493 *Aug 15, 1990Aug 2, 1994Commonwealth Scientific And Industrial Research OrganisationMoisture content by microwave phase shift and mass/area
US7078913 *Dec 31, 2003Jul 18, 2006The United States Of America As Represented By The Secretary Of AgricultureMultipath resistant microwave moisture sensor
US8104498Oct 22, 2010Jan 31, 2012Rain Bird CorporationSoil moisture sensor and controller
US8671969Dec 23, 2011Mar 18, 2014Rain Bird CorporationSoil moisture sensor and controller
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Classifications
U.S. Classification324/640, 324/605
International ClassificationG01N22/04, G01N22/00
Cooperative ClassificationG01N22/04
European ClassificationG01N22/04