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Publication numberUS3904997 A
Publication typeGrant
Publication dateSep 9, 1975
Filing dateSep 13, 1973
Priority dateSep 13, 1973
Publication numberUS 3904997 A, US 3904997A, US-A-3904997, US3904997 A, US3904997A
InventorsStinehelfer Sr Harold Eugene
Original AssigneeMicrowave Ass
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Trapped-radiation microwave transmission line
US 3904997 A
Abstract
A novel microwave transmission line is described, in which a conductor strip is supported on one side of a high-dielectric substrate body, the other side of which is not backed by a ground-plane conductor. The side of the dielectric body with the conductor strip is faced toward and spaced a distance from a ground plane conductor. A channel is provided in the ground plane conductor, and the dielectric body is in contact with the conductive material of the ground plane conductor along two paths at the sides of the channel. The conductor strip is enclosed in the channel, between the side paths, "suspended" or spaced from the ground plane body. Radiation from the conductor strip is minimized by trapping in the channel and in the dielectric body.
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Description  (OCR text may contain errors)

United States Patent 1191 Stinehelfer, Sr.

[451 Sept. 9, 1975 Harold Eugene Stinehelfer, Sr., Burlington, Mass.

[75] Inventor:

[73] Assignee: Microwave Associates, Inc.,

Burlington, Mass.

[22] Filed: Sept. 13, 1973 [21] Appl. No.: 397,156

OTHER PUBLICATIONS Brenner, H. E., Numerical Solution of Tem-Line Problems Involving Inhomogeneous Media," MTT-IS, 1967, pp. 485-487.

n tl\\\ Glance et al., A Waveguide to Suspended Stripline Transition, MTT-Zl, 2-1973, pp. 117-118. Schneider et al., Microwave & Millimeter Wave Hybrid Integrated Circuits for Radio Systems, B.S.T.J., Vol.48, 1969, pp. 1703-1713.

Primary Examiner-Alfred E. Smith Assistant Examiner--Wm. H. Punter Attorney, Agent, or FirmAlfred H. Rosen; Frank A. Steinhilper 5 7 ABSTRACT A novel microwave transmission line is described, in which a conductor strip is supported on one side of a high-dielectric substrate body, the other side of which is not backed by a ground-plane conductor. The side of the dielectric body with the conductor strip is faced toward and spaced a distance from a ground plane conductor. A channel is provided in the ground plane conductor, and the dielectric body is in contact with the conductive material of the ground plane conductor along two paths at the sides of the channel. The conductor strip is enclosed in the channel, between the side paths, suspended or spaced from the ground plane body. Radiation from the conductor strip is minimized by trapping in the channel and in the dielectric body.

15 Claims, 21 Drawing Figures PATENTEU SEP 9 I975 sum 1 OF 5 PAIENTEUSEP 9191s snme UF 6 TRAPPED-RADIATION MICROWAVE TRANSMISSION LINE BACKGROUND OF THE INVENTION The use of microstrip transmission line, generally in the form of an electrical conductor formed. as by etching on high dielectric permittivity substrate material, has become widely recognized in the designof microwave transmission circuits and integrated circuits, subassemblies and components. A basic or common microstrip structure consists of a conductor strip or other circuit pattern supported on one side of a dielectric substrate backed on the other side by an electricallyeonductive ground plane. The dielectric substrate is usually alumina ceramic because of its high dielectric constant and low loss tangent although other dielectric materials are also used. Often the microstrip circuits are completely enclosed and sealed in an electricallyconductive enclosure. Various forms of microstrip, strip line, and other transmission lines are illustrated in FIG. I of the article of E. G. Cristal et al., MICRO- GUIDE A NEW INTEGRATED CIRCUIT TRANS- MISSION LINE GMTT May l972 pages 2l2-2l4. The features of ceramic-based microstrip which led to its popularity in microwave integrated circuit (MIC) applications are: r

1. its fairly high effective dielectric constant and the consequent miniaturization of microwave circuits; 2. the ease, reproducibility and economy of producing circuits by photo etching methods; 3. the ease of mounting unpackaged semiconductor devices directly on the deposited conductors; and 4. its *open" construction which permits probing and adjustment of the circuit while it is operating. Microstrip does have a couple of drawbacks, however, which limit its applicability. First, its attenuation is relatively high, compared to coaxialline or stripline. Second, microstrip has a tendency to radiate RF energy at discontinuities.

The high loss is a consequence of the high effective dielectric constant. (The effective dielectric constant is a weighted average of the air and ceramic dielectric constants, which properly accounts for the decrease in wave phase velocity and line impedance from the values in air). The high dielectric constant concentrates the electric energy within the dielectric, enhancing the dielectric loss, while the small size of the conductor strip raises the current density, enhancing the copper losses. i

Because of its open, unbalanced configuration, microstrip tends to radiate RF power from discontinuities where higher order modes are excited. Open circuits and high-Q resonators are particularly troublesome. The radiation can be reduced by increasing the dielectric constant, which better confines the fields to the dielectric, but this is done at the cost of increased attenuation. Radiation into free space can, of course, be eliminated by enclosing the microstrip in a shielded box, as is common practice. Nevertheless, the excitation of higher order modes by discontinuities is not changed by the shielding, and the tendency toward radiation becomes manifested as increased coupling or crosstalk among the various circuit elements in.the enclosure.

functions require circuits somewhat greater than a quarter wavelength on a side. it is often difficult to satisfy this criterion without resorting to elaborately shaped enclosures and circuit substrates.

Suspended substrate microstrip line was introduced as aa way of decreasing the effective dielectric constant and. thereby, the loss. Suspended substrate (SS) has a thin strip of conductor deposited on a dielectric substrate, as in microstrip, but the substrate is placed or suspended nearly equidistant between two ground planes. The strip may be on either side of the dielectric. SS has relatively little electric energy stored in the dielectric compared to microstrip; it thus has a lower dielectric loss and wider, lower loss strips for a given im pedance level. Also, because of the reduced effect of thedielcctric on the wave propagation in SS lines; the tolerance on the dielectric material properties its thickness, dielectric constant, uniformity and surface finish which are rather critical in microstrip, are

considerably relaxed.

Of course, SS line does not alleviate the radiation and box resonance problems of microstrip. In fact, because of the lower effective dielectric constant,-SS line has an even greater tendency to radiate and a shielded, narrow enclosure is essentially mandatory.

GENERAL NATURE OF THE INVENTION The present invention provides a'novel transmission line having a channel between a dielectric body and a ground plane conductor body that are spaced a distance apart, there being a conductor strip supported on that surface of the dielectric body which faces the ground plane conductor body, the conductor strip being thereby suspended a distance from the ground plane conductor. The remaining surface of the dielectric body, outside the channel, is not backed by a ground plane conductor. The channel is provided in the ground plane conductor, and the dielectric body is in contact with the conductive material of the ground plane conductor along two paths at the sides of the channel which paths are generally parallel to and spaced from the conductor strip. The width of each path ofcontact is related to the width of the conductor strip such that radiation from the transmission line, e.g.: from discontinuities, is minimized by trapping the electric fields in the channel and in the dielectric material. For convenience, this novel transmission line may be referred to as trapped-radiation microstrip transmission line. i i

The invention improves upon the radiative properties of ordinary microstrip without paying the loss penalty associated with a higher effective dielectric constant. A line according to the invention will have a dielectric constant between those characterizing the equivalent microstrip and suspended substrate lines. Like a suspended substrate line, it will have low loss and relaxed tolerances to dielectric material properties. On the other hand, the fields will be largely confined to the channel region adjacent the conductor strip by virtue of the high dielectric constant in the fringing field zone. This .trapping of the fields in the channel reduces the coupling to the free space outside the channel or above the substrate, significantly reducing the excitation of radiation or box resonances by circuit discontinuities. The fact that a wave propagating in this line gets trapped in the channel causes the wave to propagate in a uniform manner over longer distances than in the case of microstrip or suspended substrate lines.

In addition to maintaining simultaneously low radiation and low loss, the transmission line according to the invention possesses several other advantages and useful features, some of which will be described in detail. It is an attractive transmission line medium for use in microwave integrated circuit technology, for example.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is an isometric sketch of a preferred embodiment of the invention;

FIG. 2 is a section line 22 of FIG. 1;

FIG. 3 is a section of another embodiment of the invention;

FIG. 4a is a representation of a typical electric field pattern in the embodiment illustrated in FIG. 2;

FIG. 4b is a representation of a typical magnetic field pattern in the embodiment illustrated in FIG. 2;

FIG. 5 is a set of graphs showing typical characteristic impedance of microstrip and suspended-substrate transmission lines, and a trapped-radiation transmission line according to the invention.

FIG. 6 is a graphic illustration of the impedance of a transmission line according to FIGS. 1 and 2 as a function of eccentricity of the strip conductor therein;

FIG. 7 illustrates a shorted stub in a transmission line according to FIGS. 1 and 2;

FIG. 8 illustrates structure for incorporating shuntmounted devices and bias feed-through conductors in a transmission line according to FIGS. 1 and 2;

FIG. 8a is an enlarged partial cross-section taken along line 8A8A in FIG. 8 with the parts 14 and 11 closed on each other;

FIG. 9a illustrates the incorporation of a capacitive tuning screw in a transmission line according to FIGS. 1 and 2;

FIG. 9/; illustrates a series capacitive gap in the strip conductor of a transmission line of the invention;

FIG. 94' is the equivalent circuit of FIG. 9b;

FIG. 10 illustrates a structure for launching wave energy from a coaxial line into a trapped-radiation transmission line of the invention;

FIG. 11 is a longitudinal top-section through an embodiment of the invention which incorporates in one structure a branch-line hybrid with off-center conductors;

FIG. 12a is a longitudinal top-section through a coupled-strip coupler employing trapped-radiation transmission line according to the invention;

FIG. 12b is a section on line l2bl2b FIG. 12a;

FIG. 13:: is a longitudinal side-section through an open stub end of a transmission line according to FIGS. 1 and 2;

FIG. 13!) is a top view on line l3b-l3b of FIG. 13a;

FIG. 14a is a longitudinal side-section through a shorted stub end of a transmission line according to FIGS. 1 and 2; and

FIG. 14b is a top view on line l4b14b of FIG. 14a.

DETAILED DESCRIPTION OF THE DRAWINGS In FIGS. 1 and 2, a strip 10 of electrical conductor is supported on a surface 12 of a dielectric substrate body 11, in the same manner as in a microstrip transmission line. The dielectric body 11 is affixed in paths 11a and 11b along the edges of the surface 12 to confronting surfaces 14a and 14b respectively of an electrical ground-plane conductor 14. The ground plane conductor 14 has a channel 13 in it, bounded at its bottom by a bottom wall 15 and at its sides by walls 15a, 15b extending to the surfaces 14a and 14b, respectively. The channel 13 is an elongated passage of rectangular cross-section, bounded on the bottom and two sides by electrical conductor walls 15, 15a and 15b, and on the top by the surface 12 of the dielectric body 11. The strip conductor 10 is suspended from the dielectric surface 12 spaced from the bottom wall 15 of the channel 13.

The relative dimensions of the components in FIGS. 1 and 2 are as follows:

width of conductor I0 W width of aths 1 la-l4a and l lb-l4b W/Z (approx.)

width of space between each side-wall (inner) of sides 15a. 15b and the nearer longitudinal edge of conductor 10 X width of channel l3 (W 2X) S thickness of dielectric body I l (1 thickness of channel 13 h In a practical transmission line having Z 50 ohms d W/3; W/h 3 (approx.); S M2 at fmax; and X 2.2h (approx.)

FIG. 3 is a modification of FIG. 2 in which the dielectric body 11 and strip conductor 10 are the same, but the ground plane member 18 is structurally different, being made of a thin metal channel having a bottom wall 19 and up-standing thin sidewalls 19a and 19b. At the top edge of each sidewall 19a, 19b is a flat electrically conductive plate 20, 21, respectively, of width Y on each of which the meeting path surface 14a, 1411, respectively, for the dielectric member 11 is formed. The portion 20a, 21a of each plate 20, 21, respectively, that extends inward from the sidewalls 19a, 19b, respectively, should preferably be less than W/4 in width. The channel 13 is bounded by the electrically conductive walls 19, 19a and 19b, and the surface 12 of the dielectric body 11.

The dielectric body and the ground plane body may be joined together in any fashion known to the art, now or hereafter. For example, metalized strips (16a, 16b) like the conductor 10 can be laid down on the dielectric member in the paths 11a, 1112.

FIGS. 4a and 4b illustrate typical field patterns of trapped-radiation transmission lines according to the invention. These are, respectively, computer-generator plots of the electric and magnetic fields in a line according to FIGS. 1 and 2 but, the lines being symmetrical about a longitudinal centrally-located plane between the side walls of the channel, these figures show only one half of the transmission line, each being bisected along the axis of symmetry. The magnetic field pattern (FIG. 4b) is equivalent to the electric equipotential pattern, and similarly the electric field pattern (FIG. 4a) is equivalent to the magnetic equipotential pattern. The patterns were computed by the finite element program for solving the two-dimensional La- Place s equation developed by Sylvester and coworkers at McGill University, (Z. Csendes and P. Sylvester, Dielectric Loaded Waveguide Analysis Program, IEEE Trans. Microwave Theory and Techniques, Volume MTT-l9, p. 789, Sept. I971; and Z. Csendes and P. Sylvester, A Finite-Element Field-Plotting Program, IEEE Trans. MTT, Vol. MTT-ZO, p. 294, April 1972). It will be seen that the electric field lines in FIG.

4a and the magnetic field lines in FIG. 41; exhibit a high degree of trapping in the channel 13 and in the dielectric 11, particularly in the portion adjacent the coupling path 11b- 141) at the side wall 1512.

Comparison with similarly-derived field patterns for micro-strip line and suspended substrate line, all three lines being in the same-size enclosure. and employing dielectrics having the same dielectric constant and physical thickness, and strip conductor widths chosen to correspond to substantially the same line impedance, demonstrates that field trapping in lines according to the invention is greater, and therefore radiation is more greatly restricted than in the prior lines mentioned. Thus, it is found that in all three kinds of transmission line most of the electric field is concentrated in the dielectric. This proportion is greaten in micro-strip, less in suspended substrate, and intermediate in the trapped-radiation line of FIGS. 1 and 2. However, when one considers the electric field above the dielectric substrate (i.e.: at the side opposite the side bearing the strip conductor the trapped-radiation line of the present invention has the weakest fields. The differe'nce is dramatic relative to micro-strip, and less marked but still significant relative to suspended substrate. The suspended substrate field in this region is only slightly greater than in the line according to FIGS. 1 and 2, but it falls off less rapidly as one moves off the center line. Thus can be seen, at least qualitatively, the

reduced penetration of electric fields into the air above The invention attains drastic lateral confinement in the channel 13 even though the effective dielectric constant may be only about half that of microstrip line.

In addition to reducing the lateral extension of fields, the trapped-radiation line channel (l3, 13), together with the dielectric interface, i.e., outer surface of the body 11, above the line deflect the fields back to the channel edges, effectively eliminating or greatly reducing the radiative coupling to the region above, or outside, the dielectric substrate. It is possible to increase the effective dielectric constant of the trappedradiation lines of the invention, eg; to 4 or 5, by increasing the substrate thickness or its dielectric constant, and thereby further reduce the radiation while keeping the loss below that of microstrip.

In FIG. 5, curve 31 shows the measured characteristic impedance Z,, of a set of trapped-radiation transmission lines according to FIGS. 1 and 2, in which the strip conductor 10 and widths W as follows:

0.020 inch 0.040 inch 0.060 inch 0.080 inch, and

0. 100 inch In each case the conductor was on a dielectric substrate ll of alumina having thickness d 0.020 inch and dielectric constant e 9.0; and the depth 11 of the channel 13 was 0.020 inch. Each channel had width S that was 0.090 greater than W. Except for channel width S, the geometry corresponds to that of FIGS. and 4b, where W was 0.050, and S was 0. I50 inch. In FIG. 5, Z,, is plotted as a function of the ratio W/lz, and the characteristic impedance is seen to fall between 80 ohms and 40 ohms, being 50 ohms when W/ll (approx.).

Curve 32 (in dashedline) shows for comparison the impedance of suspended substrate line on the same dielectric, as estimated from the curves of H. E. Brenner Use a Computer to Design Suspended Substrate ICs", Microwaves, Vol. 7, No. 9, p. 38, Sept. 1968. FIG. 5 illustrates that the characteristic impedance for these two lines is closely similar.

Curve 33 shows for comparison the characteristic impedance of microstrip line, as calculated from Wheelers formulas, (See M. V. Schneider, Microstrip Lines for Microwave Integrated Circuits", Bell Sys. Tech. Journal, Vol. 48, p. l42l, MayJune 1969).

Trapped-radiation lines according to the invention have four impedance-determining parameters, as is apparent from FIG. 5. These are:

By contrast, suspended substrate has only three such parameters, and microstrip has only two. These additional variable parameters make it possible to develop transmission lines having a wide variety of specifications. The channel width S can be increased until the trapping effect is lost, or decreased until the width W of the strip conductor 10 must be so narrow that losses are unacceptably high and mechanical tolerances become unacceptably close. The dielectric can be made thinner to the point where leakage flux and radiation become unacceptably high, or thicker to the point where effective dielectric constant and consequent loss become unacceptably high and the possibility of box resonance entirely within the dielectric is introduced or becomes unacceptable. In general, acceptable ranges of dimcnsisons have been indicated above, in connection with FIGS. 1 and 2, and FIG. 5, for frequencies up to 18 GHz.

Since there is no inherent reason for the strip conductor 10 to run along the center line of the channel 13, the offset, or eccentricity. A of this conductor is a variable parameter which offers one more degree of freedom in designing transmission lines according to the invention. FIG. 6 shows the measured effect of offsetting a 50 ohm strip conductor up to half the channel width. The characteristic impedance is plotted as a function of the offset A, and curve 35 shows that 2,, can be varied, essentially linearly, from 50 ohms to less than 47 ohms. FIG. 6 illustrates the comparatively relaxed tolerances that are possible in trapped-radiation lines of the invention. For typical channels, that is, those which have width S about 2 to 3 times the width W of the strip conductor 10, the lateral positioning tolerance of the conductor is not critical; a 15% offset causes only a one-ohm error in a 50 ohm line. Similarly, the channel width itself is not critical. The strip conductor having width W= 0.060 .inch was 50 ohms in a channel having width S 0.150 inch, and approximately 49 ohms in a channel having width S 0.130 inch.

Trapped-radiation lines according to the invention have many advantages over microstrip and suspended substrate lines. For example, short-circuited stubs are rarely used in microstrip because they are inconvenient to make. In order to return the open end of a stub to ground, either it must be positioned at the edge of the dielectric board (or substrate), or a hole must be provided through the board and a separate grounding connection must be made. On suspended substrate, one does not have even the option to drill a hole through the board to the ground plane. By contrast, a short circuit in lines according to the invention can be simply printed (e.g.: etched out) with the rest of the circuit, as is illustrated in FIG. 7, where parts correspond to parts in FIG. 2 have the same reference characters. The stub conductor 41 runs from the main line conductor to the edge of the channel 13, which is widened in the region 13a in the vicinity of the stub to accommodate its length. The contact paths 11a and 1 1b of the dielectric body 11 are metallized in the same manner as the conductor 10, in a common printing or etching operation, for ease in making fixed electrical contact to the meeting path surfaces 14a, 14b of the ground plane member 14. Since shorted stubs in microstrip are usually simulated by an extra quarter wavelength of open stub, they do not have the same frequency response of true shorted stubs, and they radiate. By contrast, a shorted stub as shown in FIG. 7 has none of these deficiencies, giving it a clear advantage in such circuit applications as intcrdigital and comb-line filters. This shorted stub also offers the possibility of a sliding short by filling the widened channel region 13a with a movable metal block (not shown).

The advantage of the invention that makes it an easy matter to fabricate shorted stubs also makes it easy to shunt-mount discrete circuit elements, such as semiconductors, resistors, capacitors, directly on the dielectric substrate 11. FIG. 8 (and FIG. 8a) illustrates several possible arrangements, which can be used together or in vairous combinations. The main line conductor 10 and one of the metallized meeting paths 11a can be fitted with confronting short contact tabs 43, 44, and a diode 45 can be mounted on one of these tabs 44. A conductor 46 can then be connected from the diode to the other confronting tab 43, thereby connecting the diode in shunt from the main line conductor 10 to ground. A slot 48 may be provided through the upper surface 141) of the wall b of the ground plane conductor 14, to accommodate a bias feed-through conductor 49 across the path 111;, the metallizing of which is interrupted for this purpose. The feed-through conductor connects to a long thin conductor 51 which connects at its remote end to the main line conductor 10, and has a length parallel to that conductor sufficient to provide a choke at the operating microwave frequency. This is a bias choke, and a by-pass capacitor 52, which may for example be an RF bypass semiconductor chip capaci tor, is connected between the feed-through conductor 49 and ground via the ground-contact path 11b metallizing, to complete a bias network.

FIG. 9 illustrates an arrangement, which is easily possible in the present invention, that provides the ability to do final tuning of a microwave circuit with a screwdriver, using a non-radiating screw 61 accessible from outside the finished transmission line structure. Again, parts in common with FIG. 2 bear the same reference characters, F IG 9a being a longitudinal section through an embodiment like FIG. 2. A gap 62 is provided in the main conductor 10, and the screw 61 is threaded through the bottom 15 of the ground plane member 14, where it is located precisely with reference to the gap. This is a capacitive screw, which is grounded, and with it the coupling of a series gap 62 can be varied, as shown. Similarly the effective length of an open stub (see FIG. 13) can be varied. In either case, the radiation trapping properties described above will be found to be effective.

The series gap 62 and its equivalent circuit are illustrated in FIGS. 9b and 9c, respectively, The equivalent circuit consists of a series capacitive susceptance B and two shunt susceptances B in a 71' network. Two sets of reference planes P P and T are shown for this circuit. For narrow gaps it is convenient to use a single reference plane through the center line T of the gap, and for this reference the shunt capacitors are negative to account for missing line capacitance. For wide gaps 62 it is more accurate to consider a pair of reference planes P and P at the ends of the gap. Here, while the same equivalent 11 network applies, the shunt capacitors are positive to account for fringing at the open ends. In the limit, of course, the gap may be treated as infinitely wide, and then it is conventianal to treat the open circuit capacitance as an equivalent length of line and to'define a corresponding line length correction.

A series of gaps 62, each with its tuning screw 61, appropriately spaced along the main line conductor 10, will provide a gap filter.

Transition from a coaxial line to a trapped-radiation line of the invention may follow the technique used with microstrip, and is illustrated in FIG. 10. A coaxial line 61 is represented by its inner and outer conductors 62, 63, respectively, the inner conductor being connected to the main line conductor 10 of the trappedradiation line, and the outer conductor being connected to the ground plane member 14. Test models made according to FIG. 11, in which the trapped radiation line of the invention was designed to have Z, 50 ohms, and in which the coaxial line was used to launch microwave energy into the new line, indicated VSWR per transition to be about 1.2 at X-Band, and 1.4 at 18.0 GHz, prior to any development work aimed at reducing the VSWR.

The branch line hybrid coupler is often used for 3dB couplers since it affords the desired tight coupling. FIG. 11 illustrates a circuit configuration that uses a pair of off-center main line conductors 70, 71 mounted on a dielectric substrate 74 and suspended in a common channel 72, the channel being formed in a groundplane member (not shown) which corresponds to the ground-plane member 14 in FIGS. 1 and 2, and has side walls 73, 73. In FIG. 11, the main line conductors and 71 are connected by branch line conductors 75, 76, which are formed with the main line conductors on the substrate 74. At higher frequencies, when the diameter of the hybrid ring can become so small that the coupled circuits may be effectively merge together, the coupler is reduced to a suspended substrate circuit and undesired coupling across the ring will degrade performance. The provision of an island-like conductive element 78, projecting from the bottom of the groundplane member (not shown) into the center of the hybrid ring will reduce such undesired coupling at all higher frequencies. Additional islands 77 and 79 may be provided between the main lines 70, 71, as shown, to further enhance the reduction of undesired coupling.

A simplearrangement of a pair of coupled strips (or main line conductors) 80, 81 is illustrated in FIGS. 12a and 12/). This arrangement is useful for directional couplers, as illustrated, and for half wave resonator filter circuits, to cite a few examples. The coupled strips are located closely. spaced in the same channel 13 of FIGS. 1 and 2, for example. Because the lateral confinement of the electric and magnetic fields by the channel .provides excellent isolation among various circuits'that may be formed on the same substrate 11', the feed lines 82, 83, 84 and 85 to the pair of coupled strips 80, 81 can be easily, simply and cleanly separated from each other. Each feed line is located in its own channel 82, 83, 84 or 85, respectively, thereby helping to reduce parasitic reactances and undesired coupling between the two lines. By properly proportioning the thickness of the dielectric 11 and the depth of the channel 13 at the coupled-line pair 80, 81, it is also possible to equalize the phase velocities of the even-and odd-mode waves. This has the advantage of making broader-band higher directivity couplers easier to achieve in lines according to the invention than in microstrip.

FIGS. 13:: and 1312 show how to make an open stub in a transmission line according to the invention. A section of FIG. 13a taken along line 22 will look identical to FIG. 2. The channel 13 is terminated or blocked at one end 13.5 and the main line conductor terminatcs short of that end. A non-radiating turning screw 61, like the same component in FIG. 10a, serves to capacitively terminate the stub. Trapping lines 101, 102 show how the dielectric member 11 and the screw 61 (which is grounded) reduce radiation from the free end of the stub.

FIGS. 14a and 1412 show how to make a closed stub. The technique is similar to that in FIGS. 13a and 1312. except that the top surface 91 of the end wall 13b obstruction is fitted with a groove 90, and the main line conductor extends into that groove to make electrical contact with the ground-plane member 14.

The embodiments of the invention which have been illustrated and described herein are but a few illustrations of the invention. Other alternative circuit arrangements may be made within the scope of this invention by those skilled in the art. No attempt has been made to illustrate all possible embodiments of the invention, but rather only to illustrate its principles and the best manner presently known to practice it. Therefore, while certain specific embodiments have been described as illustrative of the invention, such other forms as would occur to one skilled in this art on a reading of the foregoing specification are also within the spirit and scope of the invention.

I claim:

1. An electric wave transmission line comprising a dielectric body and an electricallyconductive body spaced apart a distance (/1) in fixed relation with respective first surfaces of each confronting each other and forming in part the enclosing walls of an enclosed elongated channel of width (S), said conductive body having two side walls bounding said channel extending toward said dielectric body into contact therewith along two paths each having a width (y), an electrical conductor strip of width (W) supported on said first surface of the dielectric body within and extending along the channel, the dielectric body having a second surface outside the channel and opposite to said first surface, which second surface directly confronts the surrounding region free of any intervening electric conductor, said width (y) being approximately W/Z, said width (S) being less than halfa wavelength at the operating frequency, and being substantially two to three times greater than said width (W), said distance (/2) being less than the quantity (SW)/2.

2. A transmission line according to claim 1. including electrically conductive closure means across said channel in electrical connection with said conductive body, said conductor strip terminating short of said closure means and thereby having an open end confronting said closure means, an electrically conductive'adjustable means supported on said conductive body movably relative to said open end for confining the microwave radiation from said open end substantially to said channel and said dielectric body.

3. A transmission line according to claim 1 composed ofa trough-shaped conductive body having up-standing side parts and a substantially planar dielectric body, said dielectric body being in contact along two spacedapart paths on said first surface thereof with the ends of said side parts, said conductor strip being between said paths.

4. In a transmission line according to claim 3, a gap in one of said paths, an electrical conductor passing through said gap out of electrical contact with said conductive body and extending to a point on said conductor strip, for bringing a bias voltage to said condutor strip from outside said transmission line.

5. In a transmission line according to claiam l, at least one gap in said conductor strip, and an electrically-conductivc adjustable means supported on said conductive body movable adjacent to and movable relative to each such gap for tuning the gap.

6. In a transmission line according to claim 3, a second conductor strip on said first surface of said diclec tric body in one of said paths. for making electrical contact with said conductive body, and supported on at least one of said conductor strips an electric wave modifying member.

7. A transmission line according to claim 1 including electrically-conductive closure means across said channel in electrical connection with said conductive body, said conductor strip extending into electrical contact with said closure means.

8. A transmission line according to claim 1 having two electrical conductor strips supported on said first surface of said dielectric body within and extending in spaced-apart relation along said channel, in a wavecoupling to each other.

9. A transmission line according to claim 8 in which at least two bridging conductors connecting said conductor strips are also supported on said first surface, said bridging conductors being spaced apart along said channel.

10. A transmission line according to claim 1 in which said conductive body is made of thin metal, and elongated flat sheets of metal each having a width Y at least approximately W/Z are fitted one to each of said ends of said side walls.

11. A transmission line according to claim 1 in which said conductive body is a rigid block of metal containing said channel, said side walls having said width Y that is at least approximately W/2.

12. A transmission line according to claim 1 in which the thickness d of said dielectric body is at least W/3.

13. In a transmission line according to claim 1, diode means in the space between said dielectric body and said conductive body, and means connecting said diode means between said conductor strip and said conductive body in the vicinity of one of said paths.

14-. In a transmission line according to claim 1, a stub transmission line section connected between said conmeans in said passage means, a reactive connection from said bias conductor means to said conductor strip, and a capacitive connection from said bias conductor means to said conductive body in the vicinity of said 15. In a transmission line according to claim 1, pasone path.

sage means through one of said paths, bias conductor UNXTED STATES PATENT OFFICE CETEMCAT CF CECTION Patent No. 3,904,997 Dated September 9 1975 Q Inventor(s) Harold Eugene Stinehelfer, Sr.

It is certified that error appears in the above-identified patent and that said Letters Patent are hereby corrected as shown below:

- Column 1, line 65, change "enclou" to enclo- Column 2, line 6, change "aa" to -a Column 4, line 17, of "width of paths lla-l4a and Q llb-l4b after "width" insert yline 48, change "computer-generator" to -computergenerated- .1 Column 5, line 53, change "and" to had- Column 8, line 51, delete "be" line 55, after "bottom" insert wall- Q Column 9, line 21, change "turning" to --tuning an talc this first ay 9? June1976 {MAM Arrest:

Q RUTH (I. MASON C. MARSHALL DAMN Anesring Officer Commissioner vj'iarems and Trademarks

Patent Citations
Cited PatentFiling datePublication dateApplicantTitle
US2721312 *Jun 30, 1951Oct 18, 1955IttMicrowave cable
US2937347 *Jan 2, 1958May 17, 1960Thompson Ramo Wooldridge IncFilter
US3400405 *Jun 1, 1964Sep 3, 1968Sylvania Electric ProdPhased array system
US3530411 *Feb 10, 1969Sep 22, 1970Bunker RamoHigh frequency electronic circuit structure employing planar transmission lines
US3617955 *Apr 8, 1969Nov 2, 1971Bell Telephone Labor IncTemperature compensated stripline filter
US3768050 *May 19, 1971Oct 23, 1973Motorola IncMicrowave integrated circuit
Referenced by
Citing PatentFiling datePublication dateApplicantTitle
US4131894 *Apr 15, 1977Dec 26, 1978Ball CorporationHigh efficiency microstrip antenna structure
US4211986 *Jul 17, 1978Jul 8, 1980Tokyo Shibaura Denki Kabushiki KaishaStrip line coupler having spaced ground plate for increased coupling characteristic
US4254383 *Oct 22, 1979Mar 3, 1981General Electric CompanyInverted microstrip phase shifter
US4776087 *Apr 27, 1987Oct 11, 1988International Business Machines CorporationVLSI coaxial wiring structure
US5105055 *Oct 17, 1990Apr 14, 1992Digital Equipment CorporationTunnelled multiconductor system and method
US5173666 *Mar 27, 1992Dec 22, 1992The United States Of America As Represented By The Secretary Of The ArmyMicrostrip-to-inverted-microstrip transition
US5319329 *Aug 21, 1992Jun 7, 1994Trw Inc.Miniature, high performance MMIC compatible filter
US5977915 *Jun 25, 1998Nov 2, 1999Telefonaktiebolaget Lm EricssonMicrostrip structure
US6370030Jul 10, 1998Apr 9, 2002Telefonaktiebolaget Lm Ericsson (Publ)Device and method in electronics systems
US6624722 *Sep 12, 2001Sep 23, 2003Radio Frequency Systems, Inc.Coplanar directional coupler for hybrid geometry
US6714104Mar 30, 2000Mar 30, 2004Nokia Networks OyInverted microtrip transmission line integrated in a multilayer structure
US6812411Mar 12, 2001Nov 2, 2004Siemens AktiengesellschaftPrinted circuit board configuration with a multipole plug-in connector
US6822532 *Jul 29, 2002Nov 23, 2004Sage Laboratories, Inc.Suspended-stripline hybrid coupler
US7106151 *Jul 24, 1998Sep 12, 2006Lucent Technologies Inc.RF/microwave stripline structures and method for fabricating same
US8228139Mar 18, 2009Jul 24, 2012Powerwave Technologies Sweden AbTransmission line comprised of a center conductor on a printed circuit board disposed within a groove
CN1293667C *Dec 19, 2003Jan 3, 2007上海交通大学Inverted microstrip transmission line based on micro electromechanical system and its producing method
EP0288767A2 *Mar 29, 1988Nov 2, 1988International Business Machines CorporationMethod for forming a shielded transmission line
EP2105988A1 *Mar 18, 2009Sep 30, 2009Powerwave Technologies Sweden ABTransmission line and a method for production of a transmission line
WO1999003315A1 *Jun 26, 1998Jan 21, 1999Ericsson Telefon Ab L MA device and method in electronics systems
WO2000062368A1 *Mar 30, 2000Oct 19, 2000Nokia Networks OyInverted microstrip transmisson line integrated in a multilayer structure
Classifications
U.S. Classification333/116, 333/238, 333/263, 333/33, 333/235, 333/247
International ClassificationH05K1/02, H01P3/08
Cooperative ClassificationH05K1/0237, H01P3/084
European ClassificationH01P3/08B2