|Publication number||US4115740 A|
|Application number||US 05/850,830|
|Publication date||Sep 19, 1978|
|Filing date||Nov 11, 1977|
|Priority date||Nov 18, 1976|
|Also published as||CA1089543A, CA1089543A1, DE2751669A1, DE2751669C2|
|Publication number||05850830, 850830, US 4115740 A, US 4115740A, US-A-4115740, US4115740 A, US4115740A|
|Inventors||Tadao Yoshida, Tadao Suzuki|
|Original Assignee||Sony Corporation|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (1), Referenced by (29), Classifications (21)|
|External Links: USPTO, USPTO Assignment, Espacenet|
This invention relates to a pulse amplifier circuit and, more particularly, to a push-pull pulse amplifier using field effect transistors and wherein a power loss to the load caused by simultaneous conduction of the field effect transistors is substantially eliminated.
Push-pull amplifier configurations are advantageous in that higher power levels can be obtained therefrom with minimal distortion. Hence, push-pull power amplifiers find ready application in audio equipment. In one type of push-pull amplifier, field effect transistors (hereinafter FET' s) are used as the amplifying elements because FET's generally exhibit switching characteristics which are improved over bipolar transistors. However, inter-electrode capacitance, particularly gate-source capacitance and gate-drain capacitance, of an FET may, in many instances, be detrimental to the overall operation of the FET push-pull amplifier.
The aforementioned inter-electrode capacitance is particularly noticeable when pulse signals are supplied to the gate electrode of the FET, and such capacitance may be considered to be the inherent input capacitance of the FET which is present at its gate electrode. This input gate capacitance cooperates with the resistance of the drive circuit which is coupled to the gate electrode of the FET so as to form an RC delay circuit. This means that the amplified pulse signal produced by the FET exhibits sloping or rounded flanks. That is, the time constant of the RC delay circuit imparts a significant delay to the pulse signal which is supplied to the FET gate electrode, thereby delaying the positive and negative transitions (i.e. the rise time and fall time) of the amplified pulse. This distortion is undesirable and can result in a deteriorated or degraded output.
When a depletion-type FET, such as a junction FET, and particularly a vertical channel junction FET, is used in a switching mode in the aforementioned push-pull pulse amplifier, this FET is more highly conductive when its gate-source voltage VGS is zero. With this zero gate-source voltage, the FET may be considered to be ON, and its drain-source voltage VDS is substantially zero. The FET is rendered non-conductive, that is, it is turned OFF, when its gate-source voltage increases to the FET pinch-off voltage. When the FET is OFF, its drain-source voltage is a maximum value, and if the FET is connected such that its drain-source circuit is connected in series with a DC supply voltage, this drain-source voltage will be substantially equal to that DC supply voltage. Because of the aforementioned RC time constant attributed to the input capacitance of the FET and the resistance of the drive circuits connected thereto, if the voltage which is applied to the gate electrode of the FET changes abruptly from a zero level to the pinch-off level, the FET will not be turned OFF immediately. Rather, a time delay is encountered until the FET is OFF. Similarly, if the voltage which is applied to the gate electrode changes abruptly from the pinch-off level to the zero level, the FET will not be turned ON until after a time delay. Hence, the ideal linear relation between drain-source voltage VDS and gate-source voltage VGS will not be attained. Rather, a change in VGS is delayed with respect to a change in VDS, thereby appearing as a non-linear relationship. Consequently, when these FET's are used in a push-pull pulse amplifier, the delay encountered in turning OFF one FET while turning ON the other means that there will be some overlap in time during which both FET's are ON. Hence, a portion of the current which otherwise would flow to the load which is driven by the FET's will be diverted so as to flow through the FET which is in the process of turning OFF. This means that a portion of the input power for driving the load is lost to the FET which should be OFF but, because of its turn-off time delay, is still ON.
The problem of turn-on and turn-off delay also is encountered in bipolar transistors. When a bipolar transistor is in saturation, the minority-carrier density stored in the base region is relatively high. In order for the transistor to be operated in its switching mode, this abnormal carrier density first must be removed before the transistor can be turned OFF. Hence, a relatively long delay may elapse before the bipolar transistor responds to a turn-off signal applied thereto. It has been proposed (Japanese Patent Publication No. 5113/64, published Apr. 21, 1964) to provide a separate minority carrier discharge circuit which is actuated when the bipolar transistor is to be turned OFF. This circuit consists of a diode which is connected to the transistor base electrode and which is poled in a direction such that when the diode is forward biased, the minority carriers may be discharged therethrough. However, this known prior art is not concerned with the problem of overlap in the ON states of transistors which are connected in push-pull relation. Hence, this prior art does not recognize the defect of power loss that may be caused by such ON-state overlap in push-pull transistors. Furthermore, although there is a turn-off delay in bipolar transistors caused by abnormally high minority current densities in the base region when the transistor is in saturation, there is no comparable turn-on delay. This differs from an FET wherein the gate input capacitance thereof causes both a turn-on and a turn-off time delay.
Although the existence of input gate capacitance of an FET has been known, there appears to have been no proposal to utilize this capacitance so as to avoid the problem of power loss mentioned above. Generally, the input gate capacitance of an FET is substantially independent of bias current and, therefore, cannot be easily controlled (i.e., minimized) by judicious selection of the bias current. Nevertheless, it has been known that the time constant of the input gate capacitance and the drive circuit resistance affects the turn-off time of the FET. In the text "FET Applications Handbook" by Eimbinder, Tab Books (1970), it is noted that when a turn-off signal is applied to an FET, an excess charge is provided on the stray capacitance and this charge must be eliminated and the capacitance recharged to supply potential before the FET can be considered OFF. The time required to eliminate the excess charge is the turn-off delay. This text also recognizes that a turn-on delay is caused by the requirement to discharge the gate-source capacitance before the FET can turn ON. However, this text notes that the rise time of the circuit, that is, the turn-on time, is much faster than the fall time, that is, the turn-off time. To account for this turn-off delay, a diode is connected to the gate electrode of the FET and is reverse biased when the FET is to be turned ON, while being forward biased to turn the FET OFF. A capacitor is in parallel with the diode to provide a discharge current path for the input gate capacitance of the FET. Even though these turn-on and turn-off delays are recognized, the problem of power loss is not. In fact, this text states that when two FET's are used to drive a single load in a multiplex type of application, it is advantageous to provide some overlap during which both FET's are ON. Accordingly, the turn-off time should be at least as long as the turn-on delay in order to assure this overlap. See Chapter 17 and particularly Sections 17-3 and 17-9 of this text.
In view of the aforementioned deficiencies of the prior art, it is an object of the present invention to provide an improved pulse amplifier using FET's which solves the problem of turn-on and turn-off delays attributed to the input gate capacitance of the FET's.
Another object of this invention is to provide a push-pull pulse amplifier using FET's wherein power loss caused by an overlap in the ON times of the FET's is substantially avoided.
A further object of this invention is to provide a pulse amplifier using FET's wherein the drive circuits coupled to the gate electrodes of the FET's, when combined with the input gate capacitance, exhibit different charge and discharge time constants.
An additional object of this invention is to provide a push-pull pulse amplifier using FET's wherein drive circuits are connected to the respective FET's, these drive circuits exhibiting a higher impedance when the FET is turned ON than when the FET is turned OFF, these impedances cooperating with the inherent input gate capacitance of the FET's to provide a higher discharge time constant for turning ON the FET and to provide a lower charge time constant for turning OFF the FET's.
Various other objects, advantages and features of the present invention will become readily apparent from the ensuing detailed description, and the novel features will be particularly pointed out in the appended claims.
In accordance with this invention, a pulse amplifier is provided with two FET's, each having a gate electrode and source and drain electrodes, and each FET exhibiting an inherent input gate capacitance. The FET's are connected in push-pull relation such that either their drain or source electrodes are connected in common to an output terminal and the other of the drain and source electrodes is adapted to receive a DC energizing voltage. First and second drive circuits are provided to supply pulse signals to the respective gate electrodes of the FET's, each of the drive circuits exhibiting a higher impedance when a pulse is supplied to turn the respective FET ON and a lower impedance when a pulse is terminated to turn the respective FET OFF. The higher impedance cooperates with the input gate capacitance of the respective FET to provide a higher discharge time constant to turn the FET ON and the lower impedance cooperates with the input gate capacitance of the FET to provide a lower charge time constant to turn the FET OFF, whereby the FET's are not ON concurrently.
The following detailed description, given by way of example, will best be understood in conjunction with the accompanying drawings in which:
FIG. 1 is a schematic diagram of one embodiment of the present invention;
FIG. 2 is a schematic diagram of an equivalent circuit which is useful in understanding the operation of the present invention;
FIGS. 3A-3C are waveform diagrams representing signals which are produced at various locations of the circuit shown in FIG. 1; and
FIG. 4 is a schematic diagram of another embodiment of the present invention.
Referring now to the drawings, and in particular to FIG. 1, there is illustrated one embodiment of a pulse amplifier in accordance with the present invention. The pulse amplifier is formed of FET's Q1 and Q2 which, in the illustrated embodiment, are complementary and are connected in push-pull configuration. Thus, FET Q1 is a p-channel device and FET Q2 is a n-channel device. The drain electrodes of FET's Q1 and Q2 are connected to a common output terminal, and their respective source electrodes are adapted to receive a DC energizing voltage. The gate electrode of each FET is connected to receive a pulse signal, such as a pulse width modulated (PWM) signal, or the like. In general, the pulse signals applied to the gate electrodes of FET's Q1 and Q2 may be characterized as rectangular wave signals, as will be greater described below.
Although a particular push-pull configuration formed of complementary FET's connected as common-source devices is shown, other push-pull circuits may be used, such as a push-pull circuit formed of two n-channel FET's or two p-channel FET's, or a push-pull circuit formed of complementary FET's connected as common-source and common-drain devices, respectively.
The output terminal connected in common to the drain electrodes of FET's Q1 and Q2 in FIG. 1 is adapted to be connected to a load 3. As an example, if the illustrated amplifier circuit is used as an audio power amplifier, the load may be formed of a low-pass filter and a loudspeaker system.
A source of DC energizing voltage B1 is connected to the source electrode of FET Q1 so as to apply a positive DC energizing voltage thereto. Another source of DC energizing voltage B2 is connected to the source electrode of FET Q2 so as to supply a negative DC energizing voltage thereto. Sources B1 and B2 are connected in series, and the junction defined thereby is coupled to load 3, as shown. For example, this junction may provide load 3 with a reference voltage, such as ground. As will be appreciated, depending upon which of FET's Q1 and Q2 is ON, current will flow from source B1, through the source-drain circuit of FET Q1 to load 3; or current will flow from load 3 through the drain-source circuit of FET Q2 to voltage source B2.
Each of FET's Q1 and Q2 is a depletion-type FET, such as a junction FET, and preferably a vertical-channel FET. One example of a vertical channel FET which can be used is described in our U.S. Pat. No. 4,021,748 and exhibits triode-type dynamic characteristics.
Suitable circuitry is provided to supply pulse signals to FET's Q1 and Q2. As an example, this circuitry may include an audio signal preamplifier. In order to simplify the explanation of the illustrated embodiment, the pulse signal supply circuitry is depicted as signal sources S1 and S2 for supplying the positive portion of a pulse signal to FET Q1 and the negative portion of the pulse signal to FET Q2, respectively. Of course, it should be appreciated that conventional pulse signal supply elements, such as an input transformer, a phase splitter, or the like, may be used to supply the pulse signals to these FET's. Hence, if the amplitude of the pulse signal, such as a PWM signal, varies from a positive level to a negative level, pulse source S1 is the equivalent circuit which supplies only that portion of the pulse signal between zero and the positive level, and pulse source S2 is the equivalent circuit which supplies only that portion of the pulse signal from zero to the negative level. Pulse source S1 is connected to the gate-source circuit of FET Q1 by a drive circuit 4, and pulse source S2 is connected to the gate-source circuit of FET Q2 by a drive circuit 5.
Each of drive circuits 4 and 5 is adapted to exhibit a relatively high impedance when a pulse is supplied therethrough to turn the respective FET ON, and to exhibit a relatively low impedance when the supplied pulse terminates to turn the respective FET OFF. In the illustrated embodiment, these impedances are resistive impedances formed of resistors 6a and 8a connected in series in drive circuit 4, and formed of resistors 6b and 8b connected in series in drive circuit 5. One of these resistors in each of the drive circuits is connected in parallel with a diode. Accordingly, in drive circuit 4, a diode 7a is connected in parallel with resistor 8a and is poled in a direction so as to be forward biased when FET Q1 is turned OFF. In drive circuit 5, a diode 7b is connected in parallel with resistor 8b and is poled so as to be forward biased when FET Q2 is turned OFF. The resistance of each of resistors 6a and 6b is represented as Rs. This resistance may be equal to the source impedance of pulse sources S1 and S2, respectively; or may be equal to a discrete resistance plus the source impedance. Resistors 8a and 8b have equal resistance represented as Rg. In the illustrated configuration wherein diodes 7a and 7b are connected in parallel with resistors 8a and 8b, resistance Rg is much greater than resistance Rs (Rg>> Rs).
The effective input gate capacitance of FET's Q1 and Q2 is dependent, to a large extent, upon the gate-source capacitance. If it is assumed that the input gate capacitance is represented as Ci, then during the FET turn-on and turn-off operations, pulse source S1, drive circuit 4 and FET Q1 may be represented as the equivalent circuit of FIG. 2. In this equivalent circuit, capacitor 9a corresponds to the input gate capacitance of FET Q1 and has a capacitance Ci. A similar equivalent circuit, not shown, may be used to represent the turn-on and turn-off modes for FET Q2. Since the operation of FET's Q1 and Q2 is substantially the same, only one need be described.
In operation, let it be assumed that the pulse signal which is supplied to the circuit shown in FIG. 1 is rectangular having a 50% duty cycle as shown by the waveform in FIG. 3A. When this pulse is positive, pulse source S1 supplies a positive-going pulse to FET Q1 whose level increases from zero to the pinch-off voltage Vp. At the same time, pulse source S2 supplies a positive-going pulse to FET Q2 whose level decreases from -Vp to zero. Conversely, when the supplied pulse is negative, pulse source S1 supplies a negative-going pulse from level Vp to zero and pulse pulse source S2 supplies a negative-going pulse of increasing magnitude from zero to -Vp. Each of the FET's is turned ON when its gate voltage is equal to zero and is turned OFF when its gate voltage is equal to the FET pinch-off voltage. That is, FET Q1 is ON when its gate voltage is zero and is OFF when its gate voltage is +Vp. FET Q2 is ON when its gate voltage is zero and is OFF when its gate voltage is -Vp.
Initially, let it be assumed that FET Q1 is ON and FET Q2 is OFF. Thus, the gate voltage of FET Q1 is zero and the gate voltage of FET Q2 is -Vp. Now, as the pulse produced by pulse source S1 increases from zero to Vp, diode 7a is forward biased. This essentially short-circuits resistor 8a, resulting in a relatively low resistive impedance in drive circuit 4. This low resistive impedance, which is substantially equal to resistance Rs of resistor 6a cooperates with capacitance Ci of capacitor 9a (FIG. 2), that is, with the input gate capacitance of FET Q1, to establish a relatively low RC time constant T1. Thus, the input gate capacitance of FET Q1 is rapidly charged because of this low time constant, and FET Q1 is turned OFF rapidly. This is represented by the vertical leading edge of the pulses shown in FIG. 3B. The input gate capacitance of FET Q1 may be thought of as being charged to the pinch-off voltage Vp.
When the pulse supplied by pulse source S1 terminates, FET Q1 is driven to be turned ON. As may be appreciated, diode 7a is reverse biased when the pulse supplied by pulse source S1 terminates because of the pinch-off voltage Vp which is stored on the input gate capacitance of FET Q1 and which is applied to the cathode of this diode. Hence, diode 7a is non-conductive, and a discharge path for the input gate capacitance, that is for capacitor 9a in FIG. 2, is through both resistors 8a and 6a. That is, when the pulse supplied by pulse source S1 terminates, resistor 8a is switched back into drive circuit 4. The resistive impedance of this drive circuit now is much higher than during the FET turn-off mode. Consequently, a relatively high time constant T2 equal to (Rs+Rg)Ci is provided in drive circuit 4. This means that a significant delay must elapse until the input gate capacitance, that is, capacitor 9a, is sufficiently discharged so that the gate voltage of FET Q.sub. 1 is low enough to turn this transistor ON. The delay in turning ON FET Q1 is represented by the negative slope of the trailing edges of the pulses shown in FIG. 3B.
While FET Q1 turns OFF and ON in the manner shown in FIG. 3B when pulses are supplied as shown in FIG. 3A, FET Q2 operates in a complementary manner. That is, when FET Q1 is turned OFF, FET Q2 is turned ON and, conversely, when FET Q1 is turned ON, FET Q2 is turned OFF. This is shown in FIG. 3C. That is, it is recalled that, initially, it had been assumed that FED Q2 is OFF. This means that the input gate capacitance of FET Q2 is charged to the pinch-off voltage -Vp. When pulse source S2 supplies a positive-going pulse to FET Q2, the voltage at the anode of diode 7b is more negative than the voltage which is supplied by pulse source S2. Hence, diode 7b is reverse-biased so as to be non-conductive. Accordingly, drive circuit 5 is formed of resistors 6b and 8b, in series, and the discharge time constant for the input gate capacitance of FET Q2 is equal to (Rs+Rg)Ci. Therefore, a significant discharge delay must elapse until the input gate capacitance of FET Q2 is sufficiently discharged to turn this FET ON. When FET Q2 is to be turned OFF, the voltage applied to the cathode of diode 7b by pulse source S2 is more negative than the discharged voltage across the input gate capacitance of FET Q2. Hence, diode 7b is forward biased so as to be conductive, thereby short-circuiting resistor 8b. This means that the charge time constant for the input gate capacitance of FET Q2 is equal to RsCi, and this charge time constant is substantially less than the aforementioned discharge time constant. Consequently, FET Q2 is turned OFF more rapidly than it is turned ON.
When the waveforms of FIGS. 3B and 3C are compared, it is appreciated that, because of the longer turn-on time constant for each of the FET's, FET Q1 will be turned OFF well before FET Q2 is turned ON, and FET Q2 is turned OFF well before FET Q1 is turned ON. Therefore, there is no overlap in the times during which both of these FET's are ON. Since FET's Q1 and Q2 are not ON concurrently, current is not diverted from load 3; and there is no power loss.
An alternative embodiment of drive circuits 4 and 5 is illustrated in FIG. 4. This alternative embodiment can be used to replace drive circuit 4, for example, and another similar circuit would be used to replace drive circuit 5. Since drive circuits 4 and 5 are similar in construction and operation, the alternative embodiment shown in FIG. 1 will be assumed to be the replacement for drive circuit 4. In the embodiment of FIG. 1, diode 7a is used as a switch to selectively connect resistor 8a to or from the circuit. In the embodiment of FIG. 4, transistors 10 and 11 are used as switching devices to selectively connect reisitors 12 and 13 to or from the circuit. In particular, transistor 10, which may be a bipolar transistor, has its base electrode connected to pulse source S1, its collector electrode supplied with a DC energizing voltage B and its emitter electrode connected through resistor 12 to a terminal 14. Terminal 14 is to be connected to the gate electrode of FET Q1. Transistor 10, which is shown as an NPN transistor, may be considered to be an emitter-follower, and is energized by a relatively positive DC energizing voltage. Transistor 11, which is shown as a PNP transistor, also may be considered to be an emitter-follower having its emitter electrode connected through resistor 13 to terminal 14 and having its collector electrode connected to the negative terminal of voltage supply source B so as to be supplied with a relatively negative DC energizing voltage. The resistance of resistor 12 is equal to Rs and the resistance of resistor 13 is equal to Rs+Rg.
In operation, when pulse source S1 supplies a positive-going pulse to turn FET Q1 OFF, this positive pulse turns ON transistor 10 so as to establish a charge path for the input gate capacitance of FET Q1 through resistor 12. The charge time constant T1 for the input gate capacitance of FET Q1 is equal to RsCi. Hence, the input gate capacitance is charged rapidly to the pinch-off voltage Vp, thereby rapidly turning OFF FET Q1.
When the pulse supplied by pulse source S1 terminates, transistor 10 is turned OFF so as to remove resistor 12 from the charge path to the input gate capacitance of FET Q1. However, when this pulse terminates, transistor 11 is turned ON so as to connect resistor 13 into the discharge path for the input gate capacitance of FET Q1. Accordingly, the input gate capacitance discharged through resistor 13, and the time constant for this discharge circuit is equal to (Rs+Rg)Ci. It is appreciated that the charge time constant T1 and the discharge time constant T2 for the input gate capacitance of FET Q1 when the drive circuit shown in FIG. 4 is used are substantially equal to time constants T1 and T2 when the drive circuit shown in FIG. 1 is used. Thus, the delay in turning ON FET Q1 is much greater than the delay in turning this FET OFF. A similar result is achieved if the embodiment shown in FIG. 4 is used as drive circuit 5. Therefore, since FET Q1 will be turned OFF more rapidly than the turning ON of FET Q2, and since FET Q2 will be turned OFF more rapidly than the turning ON of FET Q1, the problem of power loss is avoided because both transistors will not be ON concurrently.
It should be noted that when the embodiment shown in FIG. 4 is used as drive circuit 5, transistor 10 is turned ON when FET Q2 is to be turned ON, and transistor 11 is turned ON when FET Q2 is to be turned OFF. Hence, when the embodiment of FIG. 4 is used as drive circuit 5, the resistance of resistor 12 is equal to Rs+Rg, and the resistance of resistor 13 is equal to Rs so that the larger time constant will be established when FET Q2 is to be turned ON.
While the present invention has been particularly shown and described with reference to preferred embodiments, it will be readily apparent that various changes and modifications in form and details may be made by one of ordinary skill in the art without departing from the spirit and scope of the invention. For example, resistors 6a and 6b in FIG. 1 may be omitted so as to provide a very short charge time constant for the input gate capacitances of FET's Q1 and Q2. As another alternative, other suitable switching devices can be used in place of diodes 7a 7b to selectively switch resistors 8a and 8b into and out of the drive circuits. Similarly, transistors 10 and 11 can be replaced by other equivalent switching elements; and these transistors may be connected in other configurations to couple one or the other of resistors 12 and 13 between pulse source S1 (or S2) and the gate electrode of FET Q1 (or Q2). It is, therefore, intended that the appended claims be interpreted as including the foregoing as well as various other such changes and modifications.
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|U.S. Classification||330/264, 327/170, 330/251, 330/300, 327/179, 327/427, 330/269|
|International Classification||H03K5/02, H03F3/20, H03F3/217, H03F3/34, H03F3/30, H03F3/345|
|Cooperative Classification||H03F3/217, H03F3/30, H03K5/023, H03F3/2171|
|European Classification||H03F3/217, H03K5/02B, H03F3/217B, H03F3/30|