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Publication numberUS4122399 A
Publication typeGrant
Application numberUS 05/858,118
Publication dateOct 24, 1978
Filing dateDec 7, 1977
Priority dateDec 7, 1977
Publication number05858118, 858118, US 4122399 A, US 4122399A, US-A-4122399, US4122399 A, US4122399A
InventorsGeorge Ludwig Heiter, Hotze Miedema, Edwin Charles Moore
Original AssigneeBell Telephone Laboratories, Incorporated
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Distortion generator
US 4122399 A
Abstract
The problems of high cost, high loss and incomplete distortion compensation are resolved in a distortion generating circuit which employs a nonlinear phase modulator (70, FIG. 7) and a linear phase shifter (71) as a means of generating selected distortion signal components. The circuit includes an output coupler (30) and an output coupler (31) interconnected by means of a pair of wavepaths (32, 33). The nonlinear phase modulator (70), which includes a nonlinear reactive element (37), is disposed in one of the wavepaths (32). The linear phase shifter (71) is included in the other wavepath (33). By the appropriate adjustment of the linear phase shift in wavepath (33) the entire gamut of distortion characteristics can be compensated. Distortion generating circuits of the type disclosed can be employed as either predistorters or as postdistorters to compensate for the nonlinearities in electromagnetic signal devices, such as amplifiers.
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Claims(5)
What is claimed is:
1. A distortion compensating circuit for generating selected nonlinear amplitude and nonlinear phase distortion comprising:
an input coupler (30) having an input port (1) and two output ports (2, 3);
an output coupler (31) having two input ports (2', 3') and an output port (1');
a first wavepath (32), coupled to a nonlinear circuit (70), connecting an output port (2) of said input coupler (20) and an input port (2') of said output coupler (31);
and a second wavepath (33), coupled to a linear phase shifter (71), connecting the other output port (3) of said input coupler (30) and the other input port (3') of said output coupler (32);
Characterized in that:
said nonlinear circuit (70) is solely a phase modulator;
and in that the phase shift produced by said phase modulator is a function of the power of the signal in said first wavepath.
2. The circuit according to claim 1 wherein an input signal applied to port 1 of the input coupler (30) produces two signal components at port (4') of the output coupler (31);
the one component that traverses said second wavepath (33) is a replica of the input signal; whereas the other component that traverses said first wavepath (21) is
Characterized in that:
said nonlinear circuit (70) serves solely to phase modulate said other signal component.
3. The circuit according to claim 1
Characterized in that:
said nonlinear circuit (70) includes at least one voltage dependent reactance (37) embedded in a resonant circuit;
and in that said nonlinear circuit (70) is coupled to said first wavepath (32) by means of a circulator (34).
4. The circuit according to claim 1
Characterized in that:
said nonlinear circuit (70) includes a pair of voltage dependent reactances, each of which is embedded in a resonant phase modulating circuit (80, 81);
and in that said resonant circuits (80, 81) are coupled to said wavepath (32) by means of a 3 dB, quadrature coupler (50).
5. The circuit according to claim 1
Characterized in that:
the phase shift introduced by said phase shifter (71) is selected as a function of the combined nonlinear amplitude and nonlinear phase characteristic to be compensated.
Description
TECHNICAL FIELD

This invention relates to distortion compensating generators and, in particular, to such generators that are capable of generating selected distortion components for compensating a broad range of nonlinear transfer characteristics.

BACKGROUND ART

Predistortion and postdistortion techniques, for cancelling the distortion introduced by the nonlinear transfer characteristc of electromagnetic signal devices, are well known in the art. Typically, in such circuits a distortion generator is included in either the input or the output circuit of the device for the purpose of introducing compensating distortion components which serve to cancel the distortion components produced by the signal device. Obviously, the distortion compensating generator must be capable of operating over the same dynamic range and the same frequency band as the device to be compensated. In addition, the relative magnitudes and phases of the compensating components generated should be such as to cancel all the significant distortion components produced by the signal device without itself introducing other spurious distortion components. Finally, all of this must be done at a reasonable cost if it is to have any commercial value. For example, U.S. Pat. No. 3,755,754 dicloses an arrangement for compensating an amplifier employing a velocity modulation tube by means of a second velocity modulation tube or other device that has a distortion characteristic that is substantially similar to that of the amplifier to be corrected. Such an arrangement, however, can be prohibitively expensive if the only way of matching the distortion characteristic of the amplifier is to add a second, like amplifier to the circuit.

U.S. Pat. No. 3,952,260 provides an alternative distortion correcting circuit employing a less expensive diode as the nonlinear element. In this circuit, however, the diode is forward-biased and serves as a nonlinear resistance. Aside from the added loss introduced by the forward-biased diode, there appears to be no means for compensating nonlinear phase distortion. Thus, only partial compensation is possible by this arrangement.

U.S. Pat. No. 3,383,618 similarly employs a diode as a distortion compensating generator. Used in this manner, a relatively large component of the useful signal is required to drive the diode in order to produce the necessary distortion components. A correspondingly large signal component may then interact destructively with the useful signal at the output of the distortion generator thereby reducing the total output signal. In addition, the diode itself, in order to produce amplitude distortion compensating components, must serve as a nonlinear resistor and, thereby, introduces loss to the system.

Thus, the prior art distortion generators tend to be relatively costly and inefficient.

SUMMARY OF THE INVENTION

The problems of high cost, high loss and incomplete distortion compensation are resolved in a distortion compensating generator comprising an input coupler and an output coupler interconnected by means of a pair of wavepaths. One wavepath includes a linear phase shifter. The other includes a nonlinear distorter. The distorter is characterized in that the nonlinear circuit portion is a nonlinear phase modulator, and in that the phase shift introduced by the linear phase shifter is selected as a function of the nonlinear characteristic of the signal device to be compensated. In accordance with the present invention, the entire gamut of nonlinear characteristics can be accommodated by the simple expedient of changing the phase shift introduced by the linear phase shifter.

IN THE DRAWINGS

FIG. 1 shows, in block diagram, a distortion generator used as a predistorter to generate distortion cancelling signal components;

FIG. 2 shows, in block diagram, a distortion generating circuit in accordance with the present invention;

FIGS. 3, 4, 5 and 6 are vector diagrams characterizing the operation of the distortion circuit of FIG. 2 for different values of θo ; and

FIGS. 7 and 8 show two specific embodiments of the invention.

DETAILED DESCRIPTION

Referring to the drawing, FIG. 1 shows the use of the distortion generator as a predistorter 11 for cancelling intermodulation distortion components produced by the nonlinearities in the input-output characteristic of an amplifier 10. Ideally, the output signal of amplifier 10 would be an amplified replica of its input signal. However, because of slight nonlinearities in the amplifier's transfer characteristic, spurious signal components are produced which must be removed. The function of the predistorter is to generate compensating higher order distortion components which are then introduced into the input circuit of amplifier 10 in a manner to combine destructively in the amplifier output circuit with the distortion components produced by the amplifier. The present discussion of the invention relates particular to the manner in which these distortion components are generated by the distortion generator when used as a predistorter.

FIG. 2 now to be considered, shows, in block diagram, a distortion generator in accordance with the present invention comprising: an input coupler 20; an output coupler 26; and two interconnecting wavepaths 21 and 22. The former wavepath 21 includes a nonlinear phase modulator 23 comprising at least one voltage dependent reactive element 27 such as, for example, a varactor diode. The latter wavepath 22 includes a linear phase shifter 24. In operation, an input signal applied to port 1 of coupler 20 is divided into two components. The first, smaller of the two signal components, experiences a signal amplitude-dependent phase modulation as it propagates through wavepath 21. The other of the signal components experiences a signal amplitude-independent phase shift as it propagates through wavepath 22. The two components are then recombined in output port 1' of output coupler 26. The resulting signal, including distortion components, is in turn coupled to the device to be compensated, i.e., amplifier 10.

The circuit is most conveniently analyzed in terms of a single frequency input signal. While the analysis is not rigorous, it does provide a qualitative picture of the operation of the predistorter. Standard formulae can then be employed to relate the single frequency nonlinearities to the intermodulation coefficients. (See, for example, G. L. Heiter "Characterization of Nonlinearities in Microwave Devices and Systems," IEEE Trans. MTT, 21, 12, December 1973.)

Proceeding in this manner, the output signal v0, at the predistorter output terminal 1', is the vector sum of the two signal components v1 and v2 derived from wavepaths 22 and 21, respectively. At the operating frequency, ω0, phase shifter 24 produces a power - independent phase retardation of θ0 in signal v1. Since only relative phase shifts are of interest, v1 can be used as the reference signal in which case one can consider that the phase shifter 24 produces a relative phase advance of θ0 in signal v2. In addition, signal v2 is phase modulated by modulator 23 which produces an rms phase deviation θ2 that is a function of the instantaneous voltage of the signal applied thereto. Accordingly, v0, v1, and v2 can be expressed as

v0 = v1 + v2 ;                              (1)

v1 = v1 cos ω0 t;                     (2)

and

v2 = V2 cos (ω0 t + θ0 + θ2). (3)

θ2, the rms phase modulation, is given by

(d2 d1 k0)(V1)2 /2R0 ;       (4)

where

d1 and d2 are the coupling ratios of couplers 20 and 26, respectively, and have values that are much less than one;

k0 is the phase conversion coefficient of the diode in degrees/watt; and

R0 is the transmission line impedance of the diode circuit.

The specific relation for θ2 given by equation (4) is for a particular phase modulator, as will be explained in greater detail hereinbelow. If other modulator circuits are employed, similar relationships can be derived.

The magnitude V0 and the phase, θ, of the predistorter output voltage are given by

V0 ={[V1 +V2 cos (θ02)]2 + [V2 sin (θ02)]2 }1/2(5)

and

tan θ=[V2 sin (θ0 + θ2)]/[V1 + V2 cos (θ0 + θ2)]              (6)

For the practical case where V2 << V1, equations (5) and (6) simplifify to

V0 ≈ V1 [1 + (V2 /V1) (cosθ0 cosθ2 - sinθ0 sinθ2)]    (7)

and

tan θ ≈ (V2 /V1) (sinθ0 cosθ2 + cosθ0 sinθ2)                      (8)

From equations (7) and (8), the effect of different phase shifts can be noted. For example, for θ0 = 0 or 180 (and noting that θ2 and V2 /V1 are both very small quantities) V0 is essentially equal to V1 and θ ≈ θ2. Thus, for these cases the predistorter produces essentially only a signal amplitude-dependent phase shift of equal or opposite polarity, respectively. For θ0 = 90 or 270, the output voltage V0 is approximately equal to √V1 2 + V2 2, and tan θ is constant. Thus, in these cases only nonlinear amplitude variations of the compressive or expansive type, respectively, are produced. It should be noted at this point that these amplitude variations are produced by purely reactive means. For other values of θ0, the entire range of nonlinearities can be simulated, as illustrated in FIGS. 3, 4, 5 and 6.

FIG. 3 shows the case where the fixed phase shift θ0 is greater than zero but less than 90. Assume, for the purpose of discussion, that it is a property of the particular nonlinear phase modulator that as the input signal level increases, the rms phase shift θ2 increases. This causes θ to increase and V0 to decrease. The resulting nonlinear amplitude characteristic is compressive (i.e., v0 decreases with increasing signal), and phase characteristic is inductive (AM-PM conversion such that θ increases with increasing signal).

With θ0 between 90 and 180, as in FIG. 4, v0 decreases and θ decreases with increasing signal level, producing a compressive, capacitive AM-PM nonlinear characteristic.

FIG. 5 illustrates the expansive, capacitive AM-PM characteristic obtained for 180 < θ0 < 270, and FIG. 6 illustrates the expansive, inductive AM-PM characteristic obtained when 270 < θ0 < 360. Thus, knowing the nature of the nonlinearity of the signal device to be compensated, the phase shifter 24 is designed to obtain the appropriate offset θ0 for producing the appropriate type of compensating nonlinear characteristic in the distortion generator. Once the proper offset angle is established, the ratio of linear-to-distortion power output from the predistorter (V2 /V1)2 is matched to that of the device by either changing the bias voltage on the nonlinear diode, or by adjusting the coupling ratio d1 of the input coupler.

FIG. 7 shows in greater detail a predistorter in accordance with the present invention. In this embodiment, the input and output couplers 30 and 31 are four-port hybrid couplers of which one pair of conjugate ports 2-3 of coupler 30 is connected to a pair of conjugate ports 2'-3' of coupler 31 by means of wavepaths 32 and 33. The nonlinear phase modulator 70 is coupled to wavepath 32 by means of circulator 34 whose input port a is connected to port 2 of coupler 30 and whose output port c is connected to port 2' of coupler 31. The intermediate port b of circulator 34 is connected to the phase modulator which comprises a varactor diode 37 and a series resonating inductor 36. Optionally, a filter 35 can be included for suppressing any undesired harmonics of the input signals produced by diode 37. A d.c. return path 42 is provided for the diode.

The linear phase shifter 71 is coupled to wavepath 33 by means of a second circulator 39 whose input port a' is connected to port 3 of coupler 30 and whose output port c' is connected to port 3' of coupler 31. Intermediate port b' of circulator 39 is connected to the linear phase shifter 71 which comprises a length of transmission line 40 that is terminated by means of a movable short circuit 41.

In multifrequency operation, the diode circuit is tuned to resonance at the center of the frequency band of interest, and the short circuit 41 adjusted to provide the desired offset θ0. In those cases where 90 < θ0 < 270, illustrated in FIGS. 4 and 5, there is a component of v2 that combines destructively with v1, reducing the magnitude of the input signal applied to amplifier 10. To avoid this loss, the coupling between the resonant diode circuit and wavepath 38 is advantageously made critical. When this is done, the fundamental component of the incident signal is totally absorbed by the resonant circuit and only the intermodulation components produced by the nonlinear phase modulation process are reflected back to circulator 34 and onto the output port of the predistorter. With no fundamental component reflected from the modulator circuit, there is no component available to interact destructively with signal v1.

Where more distortion power is required, two diodes 82 and 83 can be employed as shown in FIG. 8. In this embodiment, circulator 34 is replaced by a 3 dB, quadrature coupler 50, and the single phase modulator 70 is replaced by a pair of phase modulators 80 and 81, where each includes one of the diodes 82 and 83 embedded in an appropriately tuned resonant circuit. More specifically, port 2 of input coupler 30 is connected to one of the ports 5 of a pair of conjugate ports 5, 6 of coupler 50. Port 2' of output coupler 31 is connected to port 6 of coupler 50. Resonant circuits 80 and 81 are connected to the second pair of conjugate ports 7 and 8 of coupler 50.

In operation, components of the input signal simultaneously energize both phase modulators 80 and 81 whose outputs combine constructively in coupler port 6. By using two diodes in the manner described, twice the distortion voltage is produced.

While the circuit has been characterized as a predistorter, it will be recognized that it can just as readily be used as a postdistorter and be placed after the device to be compensated. Similarly, the particular linear phase shifter described is only illustrative. Other types, such as that described on page 331 of the book entitled Microwave Semiconductor Devices and Their Circuit Applications, edited by H. A. Watson and published by McGraw-Hill Book Company, can also be used.

Patent Citations
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US4068186 *Jun 17, 1976Jan 10, 1978Kokusai Denshin Denwa Kabushiki KaishaCircuit for compensating for nonlinear characteristics in high-frequency amplifiers
Referenced by
Citing PatentFiling datePublication dateApplicantTitle
US4283684 *Apr 16, 1979Aug 11, 1981Kokusai Denshin Denwa Co., Ltd.Non-linearity compensating circuit for high-frequency amplifiers
US4465980 *Sep 23, 1982Aug 14, 1984Rca CorporationPredistortion circuit for a power amplifier
US4481645 *Dec 29, 1982Nov 6, 1984At&T Bell LaboratoriesLinear distortion canceller circuit
US4513250 *May 31, 1983Apr 23, 1985Northern Telecom LimitedSignal cuber
US4878030 *Oct 23, 1987Oct 31, 1989Ford Aerospace & Communications CorporationLinearizer for microwave amplifier
US4902983 *Feb 2, 1988Feb 20, 1990Nec CorporationNonlinear signal generating circuit and nonlinear compensating device using the same
US5075635 *Dec 17, 1989Dec 24, 1991Thomson-CsfDevice for correcting the phase induced by the class c operation of the solid state amplifier and radar system using such a device
US5142240 *Dec 21, 1990Aug 25, 1992Mitsubishi Denki Kabushiki KaishaAmplifier circuit with correction of amplitude and phase distortions
US5793253 *Apr 28, 1995Aug 11, 1998Unisys CorporationHigh power solid state microwave transmitter
US6255908Sep 3, 1999Jul 3, 2001AmplixTemperature compensated and digitally controlled amplitude and phase channel amplifier linearizer for multi-carrier amplification systems
US6353360 *Feb 8, 2001Mar 5, 2002Nec CorporationLinearized power amplifier based on active feedforward-type predistortion
US7478296Jan 29, 2003Jan 13, 2009Janusz RajskiContinuous application and decompression of test patterns to a circuit-under-test
US7493540Jul 20, 2000Feb 17, 2009Jansuz RajskiContinuous application and decompression of test patterns to a circuit-under-test
US7500163Oct 25, 2004Mar 3, 2009Janusz RajskiMethod and apparatus for selectively compacting test responses
US7506232Aug 11, 2006Mar 17, 2009Janusz RajskiDecompressor/PRPG for applying pseudo-random and deterministic test patterns
US7509546Sep 18, 2006Mar 24, 2009Janusz RajskiTest pattern compression for an integrated circuit test environment
US7523372 *Aug 27, 2007Apr 21, 2009Janusz RajskiPhase shifter with reduced linear dependency
US7653851Mar 26, 2009Jan 26, 2010Janusz RajskiPhase shifter with reduced linear dependency
US7805649Mar 2, 2009Sep 28, 2010Mentor Graphics CorporationMethod and apparatus for selectively compacting test responses
US7805651Dec 8, 2009Sep 28, 2010Mentor Graphics CorporationPhase shifter with reduced linear dependency
US7865794Mar 12, 2009Jan 4, 2011Mentor Graphics CorporationDecompressor/PRPG for applying pseudo-random and deterministic test patterns
US7877656Jan 13, 2009Jan 25, 2011Mentor Graphics CorporationContinuous application and decompression of test patterns to a circuit-under-test
US7900104Mar 17, 2009Mar 1, 2011Mentor Graphics CorporationTest pattern compression for an integrated circuit test environment
US8024387Aug 20, 2007Sep 20, 2011Mentor Graphics CorporationMethod for synthesizing linear finite state machines
US8108743Sep 27, 2010Jan 31, 2012Mentor Graphics CorporationMethod and apparatus for selectively compacting test responses
US8533547Jan 25, 2011Sep 10, 2013Mentor Graphics CorporationContinuous application and decompression of test patterns and selective compaction of test responses
EP0047825A1 *Jul 8, 1981Mar 24, 1982ANT Nachrichtentechnik GmbHMethod of linearising micro-wave amplifiers over a large bandwith
EP0277636A2 *Feb 2, 1988Aug 10, 1988Nec CorporationNonlinear signal generating circuit and nonlinear compensating device using the same
EP0370877A1 *Nov 17, 1989May 30, 1990Thomson-CsfCorrection device for phase displacement induced by the functioning of a "solid state" amplifier in class C, and radar chain using such a device
WO1990006625A1 *Nov 17, 1989Jun 14, 1990Thomson CsfService for phase correction induced by the class c operation of a 'solid state' amplifier and chain radar system using such device
Classifications
U.S. Classification330/149, 327/100
International ClassificationH04B1/62, H03F1/32
Cooperative ClassificationH03F1/3252, H03F1/3276, H04B1/62
European ClassificationH04B1/62, H03F1/32P4, H03F1/32P10