|Publication number||US4361797 A|
|Application number||US 06/231,799|
|Publication date||Nov 30, 1982|
|Filing date||Feb 5, 1981|
|Priority date||Feb 28, 1980|
|Publication number||06231799, 231799, US 4361797 A, US 4361797A, US-A-4361797, US4361797 A, US4361797A|
|Inventors||Yoshikazu Kojima, Masaaki Kamiya|
|Original Assignee||Kabushiki Kaisha Daini Seikosha|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (5), Referenced by (18), Classifications (7), Legal Events (4)|
|External Links: USPTO, USPTO Assignment, Espacenet|
The present invention relates to a constant current circuit for supplying a constant current to a power source.
A conventional constant current circuit is illustrated in FIG. 1. The operation principle will be hereinafter described in conjunction with FIG. 1. When the current flowing through a load 1 is represented by Iref, a voltage Vr produced across a resistor 3 having a resistance value of R is expressed by the following equation.
VR =R.Iref1 (1)
The voltage VR produced across the resistor 3 represented by the equation (1) is compared with the output voltage Vref of a constant voltage circuit 5 in a comparator 4 as illustrated in FIG. 1, and a constant current insulated gate field effect transistor 2 is driven by the output therefrom. In this case, the transistor 2 is turned on when Vref is larger than VR and is turned off when Vref is smaller than VR. That is, the constant current insultated gate field effect transistor is operated in such a way that VR is equal to Vref. Therefore, the constant current Iref1 produced by the constant current circuit as illustrated in FIG. 1 will be expressed as follows.
Iref1 =Vref /R (2)
As clearly understood from the equation (2), the conventional constant current circuit has the following disadvantages.
1. Since the comparator and the constant voltage circuit 5 are required, the circuit is complex and cannot be operated with less power.
2. The resistor 3 cannot be fabricated with less dispersion by the present integrated circuit technique so that the circuit is not suitable for integrated circuit.
An object of the present invention is to provide a constant current circuit, in which the defects in the prior art are removed, the dispersion of the constant current Iref1 is smaller, and the number of the elements is reduced so as to easily fabricate as an integrated circuit.
FIG. 1 is a circuit diagram of a conventional constant current circuit;
FIG. 2, FIG. 4, FIG. 5 and FIG. 6 are circuit diagrams of constant current circuits of the present invention; and
FIG. 3 is a graph for illustrating that a constant current value depends upon the power source voltage of the constant current circuit of the present invention.
The present invention will be described in more detail in conjunction with FIG. 2.
When conductance constants of insulated gate field effect transistors(IGFET) 8 and 10 are KP1 and KN1, respectively, and threshold voltages thereof are VTP1 and VTN, the current I1 flowing through the IGFET 8 and the IGFET 10 is expressed by the equation (3). However, the assumption is made that all of the IGFETs are operated in the saturation region and are not operated in the sub-threshold region, in order to simplify the explanation. ##EQU1##
In the equation (3), VGP represents each gate voltage of the IGFETs 8 and 9, respectively, and VGN represents each gate voltage of the IGFETs 10 and 11.
By the similar way, the current I2 flowing through the IGFETs 9 and 11 will be expressed as the following equation (4) when the conductance constants of the IGFETs 9 and 11 are KP2 and KN2, respectively and the threshold voltages thereof are VTP2 and VTN, respectively. ##EQU2##
VGN and VGP will be obtained from the equations (3) and (4) as follows.
VGN =VTN +C1 (VTP1 -VTP2) (5)
VGP =VTP2 +C2 (VTP1 -VTP2) (6)
Wherein, the C1 and C2 are constants which can be obtained from the calculation based on each conductance constant of the IGFETs and they are expressed by the following equations, respectively. ##EQU3##
Therefore, when the voltage expressed by the equation (5) is applied to the gate electrode of the IGFET 12 connected in series to the load 7 through which the constant current should flow, as illustrated in FIG. 2, the current Iref7 flowing through the IGFET 12 and the load 7 will be expressed by the following equation. ##EQU4##
In which, the assumption is made that the conductance constant of the IGFET 12 is KNO and the threshold voltage thereof is equal to VTN which is the same as that of the IGFETs 10 and 11.
As understood from the equation (9), the current Iref7 flowing through the load 7 does not depend upon the voltage VDD of the power source 6 and the resistance value of the load 7. FIG. 3 illustrates a curve showing a dependency of Iref7 for power source voltage VDD. Since all of the IGFET are operated in a saturation region when the power source voltage VDD is more than 1.2 volts, the current Iref shown in FIG. 3 does not depend upon the power source voltage VDD.
In order that each IGFET in the constant current circuit of FIG. 2 is operated in saturation region, the following condition should be satisfied. ##EQU5## wherein, KN =KN1 =KN2
In equation (11), Vl represents a voltage across the load 7. As understood from the equations (10) and (11), the power source voltage VDD =a shown in FIG. 3 is a saturation point of the constant current and the point can be determined in accordance with the selection of the conductance constant of each IGFET. When each conductance constant is determined in such a way that the value of each second term of the right side of equations (10) and (11), a constant current can be obtained by a low power source voltage. The saturating point of the constant current, that is, the power source voltage VDD =a is also adjustable by changing each conductance constant even when KN1 is not equal to KN2. As understood from the equation (9), the current value of Iref7 can be also controlled by the difference between the threshold levels of two IGFETs and the conductance constant of each IGFET. Especially, when it is controlled by the conductance constant KNO of the IGFET 12 connected in series to the load 7, Iref7 can be controlled without the dependency of the saturation operation condition.
As described above, the usable power source voltage range and the constant current value of the constant current circuit of the present invention can be controlled by the conductance constant and the threshold voltage of each IGFET.
When the circuit shown in FIG. 2 is fabricated as an integrated circuit, the range of dispersion of the constant current Iref can be reduced to about 10(%). As clearly understood from the equation (9), the constant current value Iref is the multiplication of the constant determined by the conductance constant of each IGFET and the square of the difference between the threshold levels of two IGFETs. The less than 5% dispersion of the conductance constant can be obtained by the use of present integrated circuit fabfrication techniques. Also, the difference in threshold voltage between two IGFETs can be reduced to less than 5% by the use of the ion implantation technique. Although the consumption current of the constant current circuit is the sum of I1 and I2 represented by the equations (3) and (4), as understood from the equations (3) and (4), I1 and I2 can be reduced by adjusting the value of conductance constant. Therefore, the constant current circuit of the present invention is suitable for a circuit intended to reduce current consumption.
Although a N type IGFET is used as the constant current IGFET in the circuit of FIG. 2, a P type IGFET can be also used. FIG. 4 illustrates an example wherein a P type IGFET is used. This circuit is different from the circuit of FIG. 2 in that a series circuit of an IGFET 14 and a load 13 is connected in parallel to the power source 7 and the voltage VGP is applied to the gate electrode of the IGFET 14. When the conductance constant of the IGFET is represented by KPO and the threshold voltage is represented by VTP2, the current Iref13 flowing through the IGFET 14 and the load 13 is expressed as follows by the use of the equation (6). ##EQU6##
As understood from the equation (12), the circuit of FIG. 4, as well as the circuit of FIG. 2, can produce the constant current Iref13 with less current dispersion when it is fabricated as an integrated circuit.
Although the circuits illustrated in FIG. 2 and FIG. 4 can produce a constant current with less current dispersion by utilizing the difference between two threshold voltages of P type IGFETs, the circuit of FIG. 5 is an example utilizing the difference in threshold voltage between two N type IGFETs. FIG. 6 illustrates another embodiment of the present invention wherein a pair of currents through respective load circuits 7 and 14 are maintained constant. In either case, as understood from the equations (9) and (12), the constant current Iref determined by the following equation can be obtained by the constant current circuit of the present invention.
Iref =A(K)ΔV2 (13)
A(K) is a constant determined by only the conductance constants of IGFETs included in the circuit, and ΔV is a difference in threshold voltage between the different two IGFETs. Although, to simplify the explanation, the discussion has been made for the case that all of the IGFETs are not operated in the sub-threshold region, the constant current value Iref produced by the constant current circuit of the present invention will also be determined by the equation (13), as long as they are operated in saturation region.
The effects of the present invention are as follows:
1. Since the circuit has no comparator and no constant voltage circuit, the circuit has simple structure and is suitable for reduction of the power consumption.
2. Since the constant current value is determined by the product of a constant determined by the conductance constant of the IGFET in the circuit and the difference in threshold level between two different IGFETs, the constant current can be obtained with less current dispersion by the use of the integrated circuit technique.
3. It is possible to reduce the circuit size since it is suitable for fabricating as an integrated circuit.
|Cited Patent||Filing date||Publication date||Applicant||Title|
|US3832644 *||Nov 30, 1971||Aug 27, 1974||Hitachi Ltd||Semiconductor electronic circuit with semiconductor bias circuit|
|US3875430 *||Jul 16, 1973||Apr 1, 1975||Intersil Inc||Current source biasing circuit|
|US3887881 *||Jan 24, 1974||Jun 3, 1975||American Micro Syst||Low voltage CMOS amplifier|
|US4011471 *||Nov 18, 1975||Mar 8, 1977||The United States Of America As Represented By The Secretary Of The Air Force||Surface potential stabilizing circuit for charge-coupled devices radiation hardening|
|US4264874 *||Jan 25, 1978||Apr 28, 1981||Harris Corporation||Low voltage CMOS amplifier|
|Citing Patent||Filing date||Publication date||Applicant||Title|
|US4414503 *||Dec 7, 1981||Nov 8, 1983||Kabushiki Kaisha Suwa Seikosha||Low voltage regulation circuit|
|US4442398 *||Nov 9, 1981||Apr 10, 1984||Societe Pour L'etude Et La Fabrication De Circuits Integres Speciaux-E.F.C.I.S.||Integrated circuit generator in CMOS technology|
|US4450367 *||Dec 14, 1981||May 22, 1984||Motorola, Inc.||Delta VBE bias current reference circuit|
|US4477737 *||Jul 14, 1982||Oct 16, 1984||Motorola, Inc.||Voltage generator circuit having compensation for process and temperature variation|
|US4498041 *||Sep 1, 1983||Feb 5, 1985||Tokyo Shibaura Denki Kabushiki Kaisha||Constant current source circuit|
|US4622480 *||Apr 22, 1983||Nov 11, 1986||Nippon Telegraph & Telephone Public Corporation||Switched capacitor circuit with high power supply projection ratio|
|US4808907 *||May 17, 1988||Feb 28, 1989||Motorola, Inc.||Current regulator and method|
|US4839577 *||Sep 29, 1988||Jun 13, 1989||International Business Machines Corporation||Current-controlling circuit|
|US4924113 *||Jul 18, 1988||May 8, 1990||Harris Semiconductor Patents, Inc.||Transistor base current compensation circuitry|
|US4975631 *||Dec 20, 1989||Dec 4, 1990||Nec Corporation||Constant current source circuit|
|US5059890 *||Dec 6, 1989||Oct 22, 1991||Fujitsu Limited||Constant current source circuit|
|US5086238 *||Nov 5, 1990||Feb 4, 1992||Hitachi, Ltd.||Semiconductor supply incorporating internal power supply for compensating for deviation in operating condition and fabrication process conditions|
|US5165054 *||Dec 18, 1990||Nov 17, 1992||Synaptics, Incorporated||Circuits for linear conversion between currents and voltages|
|US5491443 *||Jan 21, 1994||Feb 13, 1996||Delco Electronics Corporation||Very low-input capacitance self-biased CMOS buffer amplifier|
|US5703497 *||Jul 25, 1996||Dec 30, 1997||Integrated Device Technology, Inc.||Current source responsive to supply voltage variations|
|US7404157 *||Dec 25, 2003||Jul 22, 2008||Nec Corporation||Evaluation device and circuit design method used for the same|
|US20060107241 *||Dec 25, 2003||May 18, 2006||Nec Corporation||Evaluation device and circuit design method used for the same|
|US20090128231 *||Jan 14, 2009||May 21, 2009||Samsung Electronics Co., Ltd.||Circuits for generating reference current and bias voltages, and bias circuit using the same|
|U.S. Classification||323/316, 330/288, 327/427, 327/535|
|Jun 18, 1982||AS||Assignment|
Owner name: KABUSHIKI KAISHA DAINI SEIKOSHA 31-1, KAMEIDO 6-CH
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST.;ASSIGNORS:KOJIMA, YOSHIKAZU;KAMIYA, MASAAKI;REEL/FRAME:004001/0867
Effective date: 19820507
|May 30, 1986||FPAY||Fee payment|
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|Apr 9, 1990||FPAY||Fee payment|
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|May 16, 1994||FPAY||Fee payment|
Year of fee payment: 12