|Publication number||US4442365 A|
|Application number||US 06/326,925|
|Publication date||Apr 10, 1984|
|Filing date||Dec 2, 1981|
|Priority date||Dec 2, 1980|
|Publication number||06326925, 326925, US 4442365 A, US 4442365A, US-A-4442365, US4442365 A, US4442365A|
|Original Assignee||Nippon Electric Co., Ltd.|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (5), Referenced by (6), Classifications (9), Legal Events (4)|
|External Links: USPTO, USPTO Assignment, Espacenet|
The present invention relates to a latch circuit formed of semiconductor elements, and more particularly to an integrated latch circuit making use of insulated gate field effect transistors.
At present, among 64 K MOS dynamic RAM's, those of 5 V single power supply type are principally used. In these MCS dynamic RAM's, it becomes necessary to make provision such that at every circuit portion a utilization efficiency of a power supply should be raised as high as possible in comparison to the conventional 4 K or 16 K MOS dynamic RAM's of 12 V power supply type. Consequently, with regard to a circuit construction employed in each circuit block, it is required that a precharge potential should be a power supply level or a level that is lower than a power supply level by a threshold voltage of IGFET and that even in the case of receiving a differential input and amplifying it, the input level should be raised as high as possible, and therefore, varieties of the circuit construction would be limited. A latch circuit is one of the circuit constructions widely used in memory circuits and other MOS integrated circuits and is employed as an address buffer, data-in buffer, output buffer and the like. In a prior art latch circuit, an intermediate potential level between an input higher level and an input lower level is internally generated as a reference potential and a differentially amplified output is further amplified logically through a buffer circuit to obtain true and complementary outputs.
Therefore, the prior art latch circuit necessitates a reference voltage generator which should be met with troublesome requirements such as elimination of its dependency upon the threshold voltage of the transistor employed and suppression of the effect thereon, of variation in a power supply voltage. Further, the differentially amplified output per se cannot be used as the latch output and amplification should be performed through two stages of circuits, which requires an additional clock signals.
It is one object of the present invention to provide a latch circuit stably operative without a reference voltage.
It is another object of the present invention to provide a latch circuit operable at a high speed and with a smaller number of control signals.
It is still another object of the present invention to provide a circuit which can latch a difference signal having a large amplitude in response to an input signal by making use of a single activation timing signal.
It is still another object of the present invention is to provide an improved address inverter buffer circuit.
According to the present invention, there is provided a latch circuit comprising a first to a fourth nodes, means for supplying the first node with a logic signal, first charging means for operatively charging said second node at a first potential, means for operatively discharging a potential at the third node to the second potential, second charging means for operatively charging the fourth node at the first potential, first control means for operatively enabling the first and second charging means and the discharging means during a first period, first transfer means for operatively transferring a potential at the second node to the third node, second control means for operatively enabling the first transfer means during a second period after the first period, inverter transistor means responsive to a level of the logic signal at the first node for discharging a potential of the second node to a second potential when the logic signal level is in the first potential and retaining the first potential of the second node when the logic signal level is in the second potential and a second transfer means coupled between the fourth and the first nodes and responsive to the second potential at the third node for discharging the potential of the fourth node to the second potential. A true and a complement signals of a large amplitude are established at the fourth and second nodes.
According to more detailed aspect of the present invention, there is provided a semiconductor circuit comprising a first capacitor having its one terminal connected to a first node, precharge means for the first node; a first insulated gate field effect transistor (hereinafter abbreviated as IGFET) having its drain connected to the first node, its gate connected to a first activation clock and its source connected to a second node, a second IGFET having its drain connected to the second node, its gate connected to a first TLL-level input and its source connected to a first power supply, reset means for the second mode, a second capacitor having its one terminal connected to a third node, precharge means for the third node, a third IGFET having its drain connected to the third node, its gate connected to the second node and its source connected to the first power supply, a fourth IGFET having its drain connected to the third node, its gate connected to the first activation clock and its source connected to a fourth node, reset means for the fourth node, a fifth IGFET having its drain connected to the second node, its gate connected to the fourth node and its source connected to a fifth node, and precharge means for the fifth node. In response to activation of the first activation clock, the first TTL-level input at that moment is latched, and a difference signal having a large amplitude can be generated at the third node and the fifth node such that the third node may take a higher potential when the latched input is at a higher level but the fifth node may take a higher potential when the latched input is at a lower level.
The above-mentioned and other features and objects of the present invention will become more apparent by reference to the following description of preferred embodiments of the invention taken in conjunction with the accompanying drawings.
FIG. 1 is a circuit diagram of a prior art latch circuit used as an address inverter buffer in a dynamic RAM.
FIG. 2 is a waveform diagram of clock signals used in the RAM including the circuit of FIG. 1.
FIG. 3 shows a latch circuit according to a first embodiment of the present invention.
FIG. 4 shows a latch circuit according to a second embodiment of the present invention.
FIG. 5 shows changes in potential levels at principal nodes of the circuit shown in FIG. 3.
FIG. 6 is a circuit diagram of an address inverter buffer employing a latch circuit according to the present invention.
FIG. 7 is a waveform diagram of clock signals used in the circuit of FIG. 6.
FIGS. 8A to 8G show drive timing generator circuits for clock signals shown in FIG. 7.
Throughout the following specification, description will be made by employing MOS transistors (hereinafter called MOST's) which are representative ones of the insulated gate field effect transistor, and especially by employing N-channel MOST's. Therefore, it is assumed that a higher level is a logic "1" level and a lower level is a logic "0" level. However, with regard to circuit constructions, the situation is essentially identical even if P-channel MOST's are employed.
Referring to FIGS. 1 and 2, the prior art latch circuit used as an address inverter buffer in the conventional dynamic RAM of 5 V single power supply type will be further described in more detail. The latch circuit, as shown in FIG. 1, includes a pre-amplifier stage PA, a reference voltage generator circuit RG, and a buffer stage B. The pre-amplifier stage PA is activated by a clock signal φa, which is anticipated by other timing signals P and AL and primarily started by an external clock input φ received by the RAM. Referring also to FIG. 2, when the external clock input φ at a TTL level changes from a higher level to a lower level, the RAM circuit enters from a reset precharge state into an active state. A timing signal φ is generated in response to φ and another timing signal φ0 which primarily defines an active state is also generated in response to φ. Then two signals φ and φ0 are not employed in the latch circuit of FIG. 1. In response to the rise of the timing signal φ0, a timing signal AL falls to the ground potential. This timing AL is a timing signal for latching an address input signal and a reference potential. Before its fall to the ground potential, AL makes MOST's Q1, Q2 and Q13 conductive and an address signal at the address input terminal 1 is latched at a node 2 via MOST's Q1 and Q2 and a reference potential generated at a node 8 by the reference voltage generator RG is latched at a node 7 via a MOST Q13. MOST's Q14 to Q19 form a reference voltage generator circuit RG, and since the rated values of the minimum of the higher level and the maximum of the lower level of the address input signal are 2.2 V and 0.8 V, respectively, the center of the reference level is set at 1.5 V. Also, by the precharge timing P, MOST's Q3, Q11, Q22, Q26, and Q34 are made conductive and nodes 3, 6, 9, 11 and 12 are precharged to a high level. After the fall of AL, the precharge timing P is lowered to the ground potential, and an activation timing signal φ1 rises up to a high level. In response to the timing φ1, the pre-amplifier stage PA of the address inverter buffer starts to operate. Although a node 10 rises in level by φ1 applied via a MOST Q20, MOST's Q21 and Q23 become conducting at the same time and immediately lower the level of the node 13 to the ground potential. Therefore, the node 13 is raised in potential in a spike form as synchronized with the rise of φ1. Boot-strap capacitors C2F and C7F have equal capacitances, so that the nodes 2 and 7 rise in level in a spike form by the same amount. On the other hand, in response to the timing φ1, the nodes 4 and 5 tend to rise via MOST's Q5 and Q9, respectively. While, gate levels of MOST's Q6 and Q10 keep the potential difference between the address signal potential and reference potential, and thus a difference arises in a current capacity between MOST's Q6 and Q10. Accordingly, a drain node of a MOST (Q6 or Q10) having a lower gate level rises higher than the other. When the address input signal is at a lower level, the node 4 is raised earlier, but when it is at a higher level, the node 5 is raised earlier. When one of the nodes 4 and 5, for instance, the node 4 in case of the address input signal at a lower level, exceeds a threshold voltage, MOST's Q8 and Q12 become conducting to suppress the node 5 to a lower level and to discharge the node 6 for making the MOST Q9 non-conducting, and after all the node 5 is lowered to the ground potential. On the other hand, the node 4 rises while following the timing φ1 owing to a self boot effect of the MOST Q5. In this way, an output of a preamplifier PA appears at the nodes 4 and 5. It is desirable to use this output as an address output to be supplied to a decoder of NOR-gate type. However, it is impossible. Assuming that the minimum reference potential (1.5 V) is set at the center of the difference between the gate levels (2.2 V and 0.8 V) of the MOST's Q6 and Q10, the difference between the minimum reference potential and the gate level becomes 2.2-1.5=0.7 V or 1.5-0.8=0.7 V. Since the reference potential would normally vary by about 0.3 V per 1 V of the VDD level, the difference may possibly become 0.4 V or less. As the difference is reduced, the rise of the output on the lower level side becomes large, and hence in the case of a NOR-gate type decoder, there occurs a risk that even a selected decoder is also discharged and is thus deemed to be unselected. Therefore, normally the output of the preamplifier PA is supplied to an additional one stage of a buffer B, through which address outputs are derived. A short time after the rise of the timing φ1, a timing φ2 is generated to rise up to the VDD level, and thereby the buffer section B starts to operate. Since the nodes 11 and 12 have been precharged to conduct MOST's Q28 and Q32, nodes 13 and 14 which serve as address outputs tend to rise, respectively, by φ2 via MOST's Q28 and Q32. The operation of the preamplifier PA has been already completed at this time, and therefore one of the gate levels of the MOST's Q29 and Q33 is held at the ground potential, while the other is held at the VDD level or a level close to the VDD level. For example, when the address input is at a lower level, the node 4 is held at the VDD level, while the node 5 is held at the ground potential, and therefore MOST's Q33 and Q37 are conducting to discharge the node 12 and to make the node 14 at the ground potential. On the other hand, the node 13 rises up to the VDD level, following the timing φ2 via MOST Q28. As a result, the output of the buffer stage B, that is, a complementary output A' of high level and a true output A' of the ground potential, can be attained at the nodes 14 and 13. After the output of the buffer stage B is thus established, a timing φi is generated to terminate the timing φ1 and restore the timing AL. This is because the timing φ1 makes a D.C. current flow via MOST's Q9 and Q10 in the preamplifier PA when the address input is at a high level and the MOST Q10 is conducting by the reference potential supplied to its gate. In order to prevent this D.C. current from flowing over the entire active period, the timing φ1 is lowered to the ground potential by the activation timing φi which is generated at a later moment.
The address inverter buffer circuit shown in FIG. 1 which operates in the above-described manner has been generally employed in the conventional dynamic RAM of 5 V single power supply type, but it has the following problems:
(1) Since an address input is latched at the node 2 via the MOST's Q1 and Q2, it is difficult to guarantee an input setup time such as 0 mS, -5 mS or -10 mS.
(2) A reference voltage generator circuit RG is necessarily employed, which is associated with troublesome requirements such as elimination of dependency upon a threshold voltage of MOST's and suppression of dependency upon the VDD power supply level.
(3) Since an address input and a reference level are in themselves applied to the latch nodes 2 and 7 of the preamplifier PA and the difference therebetween is not amplified, there is a risk that at the preamplifier outputs 4 and 5 the potential on the lower level side may be possibly raised higher than the threshold voltage, and therefore, they are in themselves hardly used as address outputs.
(4) To solve the difficulty of the item (3) above, two stage construction consisting of the preamplifier stage PA and the buffer stage B is necessitated, and two activation clocks (φ1, φ2) become necessary.
(5) When an address input is at a higher level a D.C. current path is established from the timing φ1 through the MOST's Q9 and Q10 and it is necessary to lower the timing φ1 to the ground potential in the latter half of the active period. However, when an address input is at a lower level there is no such problem. Therefore, it is said that this preamplifier is asymmetric in operations in response to an input higher level and an input lower level.
Now referring to FIG. 3, a latch circuit according to the present invention will be described. The latch circuit of this embodiment comprises MOST's Q101, Q102 and Q103 connected in series at nodes 101 and 102 between two terminals of a power supply, VDD and the ground, a MOST Q104 connected between the node 102 and the ground, a MOST Q105 connected between the node 102 and a node 105, MOST's Q106 and Q107 connected in series at a node 103 between VDD and the ground, MOST's Q108 and Q109 connected in series at a node 104 between the node 103 and the ground, a MOST Q110 connected between VDD and the node 105, a capacitor C1A connected between the node 101 and the ground, and a capacitor C3A between the node 103 and the ground. The gates of the MOST's Q101, Q104, Q105, Q109 and Q110 are connected to a terminal of a precharge timing signal Pos. The gates of the MOST's Q102 and Q108 are connected to a terminal of an activation timing signal φ0. The gates of the MOST's Q105 and Q107 are connected to the nodes 104 and 102, respectively. An input terminal is connected to the gate of the MOST Q103 and the nodes 103 and 105 serve as true and complementary output terminals of the latch circuit. These output terminals may be connected to MOST's Q111 and Q112 of a drive circuit,
The operation of this latch circuit will be explained with reference to FIG. 4 which shows changes in potential levels at the nodes 101 to 105. The waveforms on the left-hand side in FIG. 4 are in case of a high level input, while those on the right-hand side in case of a low level input. In this example, a precharge timing Pos of one-shot type is used, which precharges the nodes 101, 103 and 105 to a (VDD -threshold voltage) level and resets the nodes 102 and 104 to the ground potential when it is made at a high level (VDD) during a reset/precharge period. Thereafter, it shifts to a low level in the same reset/precharge period. Accordingly, at the time point when the circuit enters an activation period, the nodes 101 to 105 maintain their respective levels under a high-impedance state. When the activation timing φ0 rises, MOST's Q102 and Q108 become conducting and electric charges at the nodes 101 and 103 is respectively transferred to the nodes 102 and 104. In the case where the input is at a low level, all these nodes 101 to 104 are brought to a level (VDD -threshold voltage-α) that is a little lower than the (VDD -threshold voltage) level. For the purpose of making this resultant level as high as possible, it is necessary to make the nodes 101 and 103 have far larger capacitances than the nodes 102 and 104, respectively. The additional capacitors C1A and C3A are provided for that purpose. When the MOST Q107 becomes conducting in response to potential rise at the node 102, the node 103 is discharged through the MOST Q107 and the node 104 is also discharged through MOST Q108, the node 103 and MOST Q107, both brought to the ground potential. During this active period, a MOST Q105 turns from a saturated state to a non-conducting state, and hence the node 105 is maintained at the precharge potential, that is, at the (VDD -threshold voltage) level. This is because after the nodes 102 and 104 have been simultaneously raised to nearly the same level in response to the rise of the timing φ0, the node 104 falls to the ground potential. Thus the low and the high levels of the nodes 103 and 105 are established as latched outputs. Accordingly, a large difference signal between the ground potential at the node 103 and the (VDD -threshold voltage) level at the node 105, can be obtained. Under this state, even if the input should change from a lower level to a higher level, provided that the change has occurred after the MOSt Q105 became non-conducting, it influences nothing upon the nodes 103 and 105. Therefore, this moment when the MOST Q105 becomes non-conducting, would determine an input hold time.
On the other hand, in the case where the input is at a higher level, when the activation timing φ0 rises, MOST's Q102 and Q103 become conducting and the node 102 held at the ground potential is connected to the ground potential, the node 101 being discharged and brought to the ground potential. At the same time, the MOST Q108 becomes conducting and the nodes 103 and 104 shift to the (VDD -threshold voltage-α) level, and since the MOST Q107 is kept non-conducting, no further change would occur. When the MOST Q105 becomes conducting in response to the rise of the level at the node 104, the node 105 is discharged through the MOST's Q105 and Q103, and hence it is brought to the ground potential. Accordingly, in this case also, the output node 103 takes the (VDD -threshold voltage-α) level, while the other output node 105 takes the ground potential, and thus a large difference in the output signals can be obtained. It is only necessary to maintain the input at a higher level until the node 105 is discharged, and this moment determines an input hold time. Therefore, a latched output having a large amplitude in response to an input can be obtained between the nodes 103 and 105 by making use of a single activation clock φ0.
In the latch circuit shown in FIG. 3, the latch operation in the case that the input is at a lower level is relatively slow. Another embodiment of the present invention shown in FIG. 5 provides a latch circuit in which the latch operation is made faster. The latch circuit of FIG. 5 comprises, in addition to the latch circuit of FIG. 3, MOST Q113 and Q114 connected in series at a node 106 between VDD and the ground, and MOST Q115 connected between the node 102 and the MOST Q103 in series with the MOST's Q101, Q102 and Q103. The gates of the MOST's Q114 and Q115 are connected to the nodes 102 and 106, respectively. The gate of the MOST Q113 is supplied with the precharge timing Pos. In the case that the input is at a lower level, when the activation φ0 rises, electric charge at the node 101 is transferred to the node 102. When the node 102 exceeds the threshold voltage, the MOST Q114 becomes conducting, and hence the node 106 is discharged and brought to the ground potential. Then, the MOST Q115 becomes non-conducting, and the node 102 is completely isolated from the input while it is maintained at the (VDD -threshold voltage-α) level. Accordingly, the input can be latched substantially at the moment when the node 102 rises. Thus, this latch circuit operates faster than that shown in FIG. 3. The subsequent operations are not different from those of the circuit of in FIG. 3. In the case that input is at a higher level also, the operations of this circuit are quite the same as the operations of the circuit of FIG. 3 except for the fact that the electric charge at the node 101 is discharged through a path consisting of MOST's Q102, Q115 and Q103, while the electric charge at the node 105 is discharged through a path consisting of MOST's Q105, Q115 and Q103.
A practical application of the latch amplifier circuit according to the present invention is shown in FIG. 6, as applied to an address inverter buffer circuit. The buffer circuit of FIG. 6 comprises the latch circuit of the embodiment of FIG. 5 consisting of the MOST's Q101 to Q115 and the capacitors C1A and C3A and further having precharge transistors Q101', Q104', Q106', Q109', Q110, and Q113 additionally connected in parallel with the respective MOST's Q101, Q104, Q106, Q109, Q110, and Q113. The added precharge MOST's receives at their gates a precharge timing P. The buffer circuit further comprises a drive circuit having MOST's Q120 to Q138 in addition to the MOST's Q111 and Q112 whose gates are connected respectively to the output nodes 105 and 103 of the latch circuit.
Referring also to FIG. 7, the nodes 101, 103, 105 and 106 of the latch circuit and nodes 108 and 113 of the drive circuit are precharged by the one-shot precharge timing Pos and a recharge timing P. Also, by these timings, the nodes 102 and 104 of the latch circuit and nodes 109, 110, 111 and 112 of the drive circuit are discharged to the ground level. In response to the rise of the activation timing signal φ0, the operation of the latch circuit starts. If the address input signal at the gate of the MOST Q103 is of a low level, the node 103 turns to the ground potential and the node 105 takes the level of VDD -(threshold level), and vice versa if the address input signal is of a high level, as described in the foregoing with respect to the latch circuits of FIGS. 3 and 5. Thus, latch amplification operation is completed and a large difference in level is established between the output nodes 103 and 105. Then, the timing signal φ1, rises until it reaches the VDD level, enabling the operation of the drive circuit. If the address input has been at a low level, the latch output node 103 is kept at a level close to the ground potential and 105 at a level of VDD -(threshold voltage), and therefore the MOST Q112 of the drive circuit is nonconducting and Q111 conducting. Accordingly, the timing signal φ1, raises a level of the node 112 via a MOST Q135 which is conducting by the precharged level at its gate node 113, while the conducting MOST Q111 suppresses the node 109 to a lower level. The raised level at the node 112 turns MOST's Q122 and Q125 to conducting, thereby discharging the node 108, turning "OFF" MOST Q123, and bringing the node 109 to the ground potential. It also turns "ON" MOST Q132, that is a level of a node 111. As a result, the node 111, a complementary output A', reaches a level of VDD minus threshold voltage and makes MOST Q128 conducting to hold the node 110, a true output A', at the ground potential. After completion of the operation of the address inverter buffer, a timing signal φi is generated and applied to MOST's Q124 and Q136 to discharge the latch output nodes 103 and 105, in order to prevent such an undesirable situation from occurring that if an address input hold time is too short, the node 103 or the node 105 may be insufficiently discharged not to reach a complete low level.
The problems (1) through (5) enumerated in connection to the prior art latch circuit used as an address inverter buffer of FIG. 1 are resolved in the address inverter buffer of FIG. 6, in the following manner:
(1) Since an address input is directly applied to the gate of the MOST Q103, the problem of the setup time would not arise.
(2) A reference voltage generator is unnecessary.
(3) In response to an address input, a large difference in level is established between the nodes 103 and 105 only by application of the early activation timing which is solely used in the prior art circuit to generate the superfluous timing AL.
(4) An additional activation timing φ2 is not needed.
(5) No D.C. current patch exists.
The timing signals employed in the address inverter buffer of FIG. 6 and illustrated in FIG. 7 are generated by the timing generator circuits shown in FIGS. 8(A) to 8(G), in the manner described below.
When an external input clock φ of a TTL-level applied to a timing generator circuit of FIG. 8(A) changes from a high level to a low level, a MOST Q141 becomes non-conducting and a first stage timing φ rises up to the VDD level through a MOST Q140 driven in an unsaturated region. In a source-follower consisting of a MOST Q142, the level of a node 201, a timing φ0, follows the rise of the gate level φ and rises up to a level of VDD -(threshold voltage). During the period of the external input clock φ being at a high level, i.e., the reset precharge period, a D.C. current flows via MOST's Q140 and Q141. This current should be suppressed at a rated value as a stand-by current, and accordingly the current capacity for the MOST Q140 is made small. Therefore, the timing signal φ is satisfactorily used only for a first timing activation and precharge. While the timing φ1 has a larger margin to a load capacity to be driven, but its use is restricted because of its high level equal to the (VDD minus threshold voltage) level. In the latch amplifier circuit of the present invention, the timing φ0 merely serves to transfer electric charges at the nodes 101 and 103, and therefore it can be used without any difficulty.
In response to the rise of the timing φ0, a timing P0 turns to a low level by the generator circuit of FIG. 8(B), because the current capacity of a MOST Q146 is far larger than that of a MOST Q145.
At the same time, a MOST Q154 of the generator circuit of FIG. 8(C) receiving φ0 at its gate becomes conducting, and a node 202 which has been held at a high impedance and at a far higher level than VDD is discharged, while a MOST Q158 becomes conducting in response to a source-follower level of a MOST Q149 whose gate is supplied with the timing φ. As a result a timing P is lowered to the ground potential. In response to fall of the timing φ0, the timing P0 shifts to a high level, and subsequently the timing P shifts to a high level.
The timing Pi is generated by delaying the timing P, as shown in FIG. 8(D). The one-shot precharge timing Pos is held at the ground potential during the active period, and raised by the rise of the timing P applied to a MOST Q161 of the generator circuit of FIG. 8(E) and turned to the ground potential by the rise of the delayed timing Pi. In the reset/precharge period after the timing Pos has fallen, the nodes 101 to 106 in the latch circuit of FIGS. 5 and 6 are floating and may be affected by other nodes. Therefore, in the latch circuit of FIG. 6, these nodes are fixed by the timing Po to the respective levels via MOST's Q101', Q104', Q106', Q109', Q110' and Q113' connected in parallel thereto. Current capacities of these MOST's are small to such extent that they may not adversely affect the circuit operation.
The timings φ, φo, Po and P are applied to the φ1, generator circuit of FIG. 8(F). A node 203 is charged via a MOST Q167 by the timing φ. When the timing φo has risen and a node 204 has been discharged, a node 205 is raised via a MOST Q168 and the node 29 which is then at a high impedance state is raised far higher than the VDD level by a boot-strap capacitor C29 F. Then, the timing φ1, rises up to the VDD level. The timing φi is generated by delaying the rise of the timing φ, via the genrator circuit of FIG. 8(G).
In the drive circuit shown in FIG. 6, the nodes 109 and 112 are not directly used as true and complementary address outputs A' and A', for the purpose of reducing the load of the timing signal φ1. Where the timing signal φ1, has a sufficient driving capacity, these nodes 109 and 112 can be used as address output terminals. Since a large difference in level appears between the latch output nodes 103 and 105, the raising of the output on the low level side would not cause any problem.
As described above, the present invention provides a latch circuit producing a true and complementary pair of latched and amplified output signals without any necessity of a special latch timing signal nor a reference voltage. The latch circuit can be directly connected to a buffer without an additional timing signal to form an address inverter buffer circuit. In that case, a setup time as well as a hold time of an address input are also shortened as compared to the prior art circuit, and it can operate at a higher speed than the prior art circuit because of unnecessity of an additional activation timing. The latch circuit of the present invention may be also used in data-in buffer and data-output buffer of memory devices and in other and in other application where a true and complementary pair of latched and amplified signals are required.
|Cited Patent||Filing date||Publication date||Applicant||Title|
|US3641360 *||Jun 30, 1969||Feb 8, 1972||Ibm||Dynamic shift/store register|
|US3832578 *||Jun 13, 1973||Aug 27, 1974||Hitachi Ltd||Static flip-flop circuit|
|US3846643 *||Jun 29, 1973||Nov 5, 1974||Ibm||Delayless transistor latch circuit|
|US4149099 *||Sep 9, 1977||Apr 10, 1979||Nippon Electric Co., Ltd.||Amplifier circuit for obtaining true and complementary output signals from an input signal|
|US4216389 *||Sep 25, 1978||Aug 5, 1980||Motorola, Inc.||Bus driver/latch with second stage stack input|
|Citing Patent||Filing date||Publication date||Applicant||Title|
|US4559608 *||Jan 21, 1983||Dec 17, 1985||Harris Corporation||Arithmetic logic unit|
|US4680482 *||Jul 16, 1986||Jul 14, 1987||Nec Corporation||Inverter for use in binary counter|
|US4785206 *||Jul 7, 1986||Nov 15, 1988||Nec Corporation||Signal input circuit utilizing flip-flop circuit|
|US4931675 *||Nov 25, 1986||Jun 5, 1990||Kabushiki Kaisha Toshiba||Semiconductor sense amplifier|
|US4937479 *||Jan 30, 1989||Jun 26, 1990||Nec Corporation||Data latch circuit with improved data write control function|
|US4952826 *||Jun 3, 1988||Aug 28, 1990||Nec Corporation||Signal input circuit utilizing flip-flop circuit|
|International Classification||H03K19/096, H03K19/0185, G11C11/408, H03K3/356|
|Cooperative Classification||H03K3/356026, H03K3/356078|
|European Classification||H03K3/356D1, H03K3/356E2|
|Dec 2, 1981||AS||Assignment|
Owner name: NIPPON ELECTRIC CO., LTD. 33-1, SHIBA GOCHOME, MIN
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST.;ASSIGNOR:NAGAMI, AKIRA;REEL/FRAME:003963/0658
Effective date: 19811201
|Oct 1, 1987||FPAY||Fee payment|
Year of fee payment: 4
|Sep 30, 1991||FPAY||Fee payment|
Year of fee payment: 8
|Sep 5, 1995||FPAY||Fee payment|
Year of fee payment: 12