|Publication number||US5001724 A|
|Application number||US 07/297,357|
|Publication date||Mar 19, 1991|
|Filing date||Jan 13, 1989|
|Priority date||Jan 13, 1989|
|Also published as||CA2003463A1, CA2003463C, DE69020589D1, DE69020589T2, DE69033478D1, DE69033478T2, EP0378405A2, EP0378405A3, EP0378405B1, EP0651259A1, EP0651259B1|
|Publication number||07297357, 297357, US 5001724 A, US 5001724A, US-A-5001724, US5001724 A, US5001724A|
|Inventors||Raymond A. Birgenheier, Richard P. Ryan|
|Original Assignee||Hewlett-Packard Company|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (3), Referenced by (55), Classifications (17), Legal Events (7)|
|External Links: USPTO, USPTO Assignment, Espacenet|
The present invention relates generally to digital radios and, more specifically, to measuring the phase and amplitude errors in a continuous-phase-modulated signal.
Presently a number of manufacturers manufacture and market radios for use in communications, such as digital cellular radios and the like. Typically each manufacturer provides its own specifications for its products. Traditionally the accuracy of these specifications has been measured using many separate, possibly indirect methods. Phase accuracy of the transmitted signal, for example, typically is indirectly determined by measuring spurious signals, phase noise, the modulation index, frequency settling speed, carrier frequency and data clock frequency. Further, amplitude measurements present special problems because the amplitude versus time profile must be synchronized to the data typically utilizing external equipment.
It has been proposed that a standardized mobile digital radio system be implemented throughout Europe. Such a radio system would require that all components such as transmitters and receivers for example, be manufactured to standard specifications measured by a common method. A group known as the Group Speciale Mobile (GSM) has proposed a measurement technique to measure the accuracy of the modulation process of the transmitted signal. In the proposed measurement technique, a sampled measurement of the transmitted phase trajectory is obtained. This measurement is compared with the mathematically computed ideal phase trajectory to determine the phase difference between the transmitted signal and the ideal signal. The regression line of the phase difference thus determined provides an indication of the frequency error and the regression line is subtracted from the phase difference to give the phase error. Utilization of a standard method such as this would simplify the testing and manufacture of radios. An individual manufacturer would then only need to insure that the standardized overall phase error specifications were met rather than several interrelated specifications.
The present invention provides a method and apparatus for computing the ideal phase trajectory of a transmitted signal to be used in the above described GSM standard phase error measurement method. According to the principles of the present invention a transmitted signal is mixed with a local oscillator signal to provide an intermediate frequency (IF) signal having a relatively low frequency which is then filtered and sampled by an analog-to-digital convertor (ADC). The digitized samples of the IF signal are then filtered in a digital low pass filter, such as a linear-phase finite impulse response (FIR) filter to eliminate the IF signal harmonics without distorting the phase modulation of the transmitted signal. An FIR digital filter is less complex and less expensive than an equivalent analog filter required to perform this filtering operation.
The transmitted signal phase trajectory and amplitude profile are calculated from the filtered IF signal samples. A Hilbert transformer is utilized to create two component signals that are in phase-quadrature with each other. The signal phase trajectory is provided by calculating the arctangent of the quadrature signals and the amplitude is calculated as the square root of the sum of the squares of the quadrature signals.
The signal phase trajectory is then utilized to detect the data and determine the data clock phase. Detection of the data could be accomplished utilizing a Viterbi decoder or, in the case of a high signal-to-noise ratio (SNR) and low inter-symbol-interference (ISI) signal, by differentiating the phase trajectory. Differentiation of the phase trajectory provides the instantaneous frequency of the signal from which the carrier frequency may be subtracted to provide the frequency deviation of the signal. The instants of time at which the frequency deviation passes through zero are then used in a least squares algorithm to estimate the data clock phase. An accurate estimation of the data clock is critical to the measurement of phase errors.
The zero crossing of the frequency deviation function are also used to detect the data. Synchronization of the data is accomplished utilizing a correlation scheme between the detected data and a known portion of the data sequence such as a preamble. The synchronization information is then used to find the time interval of interest in the measurement operation. The synchronization information is also used to synchronize the amplitude versus time profile with the data clock.
Utilizing the data clock phase, the detected data sequence and the time interval of interest, a digital signal synthesizer mathematically generates the ideal phase trajectory corresponding in the transmitted signal. The ideal phase trajectory thus generated is subtracted from the previously measured phase trajectory of the transmitted signal to provide a signal phase difference versus time measurement. A linear regression analysis performed on the phase difference versus time measurement provides an estimate of the frequency error as well as the instantaneous phase error.
FIG. 1 is a flow chart illustrating a first embodiment of a method for measuring the phase error of a transmitted signal according to the principles of the present invention.
FIG. 2 is a conceptual block diagram of an apparatus for measuring the phase error of a transmitted signal according to the method shown in FIG. 1;
FIG. 3 is a flow chart of a method for measuring the received amplitude and the phase error of a transmitted signal according to the principles of the present invention;
FIGS. 4, 5 and 6 are functional block diagrams illustrating three different techniques for converting an IF signal to in-phase and quadrature-phase signals;
FIG. 7 is a frequency plot illustrating a typical frequency deviation function for an GMSK.3 modulated signal;
FIG. 8 is a plot illustrating the error in the detected zero crossings of the frequency deviation plot shown in FIG. 7;
FIG. 9a is a plot showing the phase pulse response for minimum shift-key modulation;
FIG. 9b is a plot showing the phase pulse response for Gaussian minimum shift-key modulation;
FIG. 10 is a plot showing the instantaneous phase difference and linear regression curve;
FIG. 11 is a plot showing instantaneous measured phase error versus bit number;
FIG. 12 is a plot showing measured pulse amplitude;
FIG. 13 is a plot showing an expanded view of the rise time of the pulse shown in FIG. 12; and
FIG. 14 is a plot showing an expanded view of the fall time of the pulse shown in FIG. 12.
Referring now to FIG. 1, a flow chart illustrating a first preferred embodiment of a method for measuring the phase error of a continuous-phase-modulated RF signal is shown. A modulated RF signal generated by a transmitter is received and converted to digital form by a digitizer circuit 1. The digitized signal is then converted or transformed into its component in-phase and quadrature-phase signals by a transformation circuit (such as shown in FIGS. 4, 5 and 6) and the transmitted signal amplitude and phase functions are computed by a calculator 3 from the component signals. Utilizing a known synchronization signal 9, which may comprise a known sequence of data bits, a preamble or midamble for example, the bit sequence representing the transmitted data is synchronized, block 4 from the phase and amplitude functions to provide the transmitter data clock and a test data interval. A data detector 5 detects the data bit sequence and provides the three signals, transmitter data clock, test data interval and the data bit sequence to a synthesizer block 7 to synthesize or mathematically calculate an ideal phase function corresponding to the transmitted signal. The data detector 5 may be implemented as a maximum likelihood sequence estimator utilizing the Viterbi algorithm. The measured phase function (i.e., the transmitted signal phase) is subtracted from the ideal phase function thus synthesized in block 7 to provide a phase difference. A linear regression in block 8 of the phase difference then provides the frequency error, the slope of the regression line 101, and the phase error, curve 102 (as shown in FIG. 10).
Referring now to FIG. 2, a conceptual block diagram of an apparatus for measuring the phase error and phase amplitude of a continuous-phase-modulated RF signal is shown. The modulated RF signal is received by a receiver 20 and coupled to a down conversion mixer circuit 11. This mixer circuit receives a local oscillator signal on line 12 generated by the local oscillator 13 and a test signal on line 15 to provide an intermediate frequency (IF) signal having a substantially lower frequency than that of the test signal, in the present embodiment the IF frequency is preferably 700 KHz. The IF signal is filtered in an analog anti-aliasing filter 17 to remove local oscillator and RF signal feed through and spurious signals. The filtered IF signal is coupled to a digitizer 19 to convert the analog IF signal to a discrete-time data sequence at a high sample rate, preferable at 2.8 million samples per second (Msps). An HP70700A digitizer manufactured by Hewlett-Packard Company may be used for this purpose or the digitizer 19 may be implemented by an ADC sampling at a high rate as shown in FIGS. 4, 5 and 6. After conversion to an IF signal having a frequency of approximately 700 KHz, the test signal test can be represented as
y(t)=A(t) cos [(ω0 +Δφ)t+φ(t;a)+φ0 ](1)
A(t) is the received signal amplitude;
ω0 =2 π(700 KHz) is the nominal IF signal frequency;
Δω is the frequency uncertainty;
φ(t;a) is the received signal phase modulation function;
and φ0 is an unknown offset phase.
As given here only φ(t;a) is a function of the data sequence a; however, in general A(t) may also be a function of a.
A transmitted RF signal or the IF signal down converted from the RF transmitted signal defined by equation (1) typically will be received in bursts having a duty cycle of 0.125 and being approximately 0.5 milliseconds (ms) in duration.
A(t) and φ(t;a) are, respectively, the amplitude modulation and phase modulation of the received signal (i.e., the transmitted signal) which will be different than the ideal modulation of the transmitted signal. The present method determines the difference between the values of the received signal functions A(t) and φ(t;a) and the ideal values of these functions.
The digitizer 19 converts the IF signal defined by equation (1) to a sequence of discrete time samples. If the sampling points are given as t=kTS, k=0, 1, 2, . . . . where TS is the time period between samples, and if we define Ω0 =ω0 TS and ΔΩ=ωTS, then the sequence of samples can be written as
y[k]=A[k]cos [(Ω0 +ΔΩ)k+φ(k;a)+φ0 ](2)
k=0, 1, 2, . . . .
Quantized values of equation (2) provide the sequence of binary numbers coupled to the digital signal processor 21 for implementation of the present method.
The outputs of the digital signal processor 21, phase error, frequency error and the amplitude profile are coupled to various display means, such as a cathode ray tube (CRT) 22 and a printer 18. The display means include the required circuity to format the display of the information provided by the digital signal processor 21. Typically, the phase, frequency and amplitude information are plotted versus time with the time interval defined by the number of data bits contained in a transmitted signal burst. FIGS. 10 and 11 are examples of phase difference and frequency error and phase error plots while FIGS. 12, 13 and 14 are plots of the transmitted signal amplitude profile.
FIG. 3 is a flow chart illustrating a second preferred embodiment of the method according to the principles of the present invention for determining the received RF signal amplitude, A[k], and the difference between the measured phase modulation, φ(k;a), of the received RF signal and the ideal phase modulation, φ(k;a). The modulation functions have been discretized by replacing "t" with kTS, k=0, 1, 2, . . . .
The first step in the flow diagram is to pass the digital IF samples through a low-pass digital filter 23. The low-pass digital filter 23 would preferably be a finite impulse response (FIR) filter that would have a linear phase response to avoid distortion of the phase modulation of the signal passed by the filter 23. The purpose of the low-pass filter 23 is to eliminate the harmonics of the 700 kHz IF signal. An FIR digital filter can perform this job with relative ease and with less cost than an analog filter which otherwise would be required.
After the initial low-pass filtering, the signal is converted to two component signals that are in phase quadrature with each other. Three different techniques are proposed as possible methods for producing the quadrature signals.
Referring now to FIG. 4, a first method of conversion to in-phase, I[k], and quadrature-phase, Q[k], (I-Q conversion) signals utilizes a Hilbert transformer 31. An RF signal is down converted to an IF signal by mixing with a local oscillator signal in mixer 39. The resulting IF signal is coupled to an ADC 35 via band pass filter 37. The filtered IF signal is converted to a digital signal by a high-sampling rate ADC 35 which is clocked by the sample signal on line 36. The Hilbert transformer 31 comprises a filter with a constant magnitude response and a phase response of -90 degrees for positive frequencies and +90 degrees for negative frequencies. An approximation to the Hilbert transformer 31 can be realized with a anti-symmetric FIR filter 31 that has an ideal phase response and an amplitude response that is nearly ideal over the range of frequencies of the signal. Delay line 33 compensates the in-phase signal for time delays introduced into the quadrature-phase signal by the FIR filter 31.
Referring now to FIG. 5, a second method of I-Q signal decomposition involves mixing the digitized IF signal with quadrature signals at mixers 41 and 43 and passing the low-frequency components through low-pass filter 45 and 47, respectfully. If the signal given by equation (2) is multiplied by 2 cos(Ω0 k) and -2 sin (Ω0 k), and the double frequency terms rejected by low-pass filtering, then the outputs of the low-pass filters are
I[k]=A[k] cos [ΔΩk+φ(k;a)+φ1 ]
Q[k]=A[k]sin [ΔΩk+φ(k;a)+φ1 ]; k=0, 1, 2, . . . . (3)
Equations (3) represents the desired I-Q signals.
The digital implementation of the I-Q mixing method illustrated in FIG. 5 has a significant advantage over a corresponding analog implementation in terms of the precise quadrature phase and amplitude balance that can be maintained. Precise balance of the quadrature signals is a critical requirement for this method.
Referring now also to FIG. 6, I-Q signal decomposition involves the utilization of a Hilbert transformer 51, delay line 49 and four mixers 53, 55, 57 and 59. This configuration approximates two single-sideband mixers that are in phase-quadrature. The advantage of this method over that shown in FIG. 5 is the elimination of the low-pass filters 45 and 47 which are not required because the double frequency terms are cancelled by the single-sideband mixers.
All three techniques described above will allow decimation of the I[k] and Q[k] samples by a factor of four or more to allow efficient processing of I[k] and Q[k]. An advantage of the low-pass filtering shown in FIG. 5 is a reduction in ADC quantization noise introduced by the digitizer 19.
After I[k] and Q[k] are produced, amplitude and phase functions are computed and output on lines 24 and 26, respectively. The amplitude function is given as
A[k]=SQRT[I2 [k]+Q2 [k]]
k=0, 1, 2, . . . , K (4)
and the phase function is given as
k=0, 1, 2, . . . , K (5)
K+1 is the number of samples in a burst, for example, if the duration of a burst is 0.5 milliseconds and the sampling rate is 2800 Ksps, then K=1400.
The phase samples given by equation (5) are passed through a differentiator to produce samples of the frequency versus time function. The differentiator 25 would preferably be an anti-symmetric FIR digital filter that has a linear magnitude response and a 90░ phase shift over the range of frequencies of the test signal. Like the Hilbert transformer 31, the differentiator 25 is a well-known digital filter that is easily and accurately implemented in digital hardware.
Referring now also to FIGS. 7 and 8, a typical frequency deviation function for GMSK.3 modulation which is a modulation scheme proposed in Europe for digital mobile radios is shown. In FIG. 7, (f-fc)Tb is the frequency deviation from the signal carrier (IF) frequency, fc, normalized by the bit rate fb= 1/Tb where Tb is the bit interval. The frequency deviation is shown for 36 bits in FIG. 7. A positive value of frequency deviation over a bit interval represents one binary state and a negative value the other binary state. The frequency function shown in FIG. 7 represents the bit sequence
or the complement of this sequence.
From FIG. 7, it can be seen that the frequency deviation passes through zero approximately at multiples of Tb as shown in FIG. 8. From FIGS. 7 and 8, it can be seen that if the bit pattern is known, then errors in the zero-crossings from multiples of Tb are predictable. For example, if bit 10 is followed by bit 11, then the zero-crossing between bit 10 an bit 11 will have an error of -0.0142Tb. The error in the zero-crossing between bit 00 and bit 10 will be 0.0142Tb and the error in zero-crossing between bit 11 and bit 00 will be approximately zero, etc.
The output of the differentiator 25 is not a continuous time function as shown in FIG. 7 but is actual samples (values) of the frequency function. For example, if the bit rate is 270 kbps and the sampling rate is 2.8 Msps, then there would be 10.37 samples per bit.
Referring again to FIG. 3, following the differentiator 25, the IF frequency is subtracted (block 27) from the frequency function to produce the frequency deviation function as presented in FIG. 7. The next step, block 29, is to detect the zero-crossing from which the received data sequence is detected as illustrated by bit sequence (6). Since discrete time samples of frequency deviation are available, the zero-crossings are detected using an interpolation algorithm. From the detected data sequence, a correction is made, block 31, to compensate for the difference in zero-crossings from multiples of Tb . These compensated zero-crossings provide the data used to establish a data clock synchronized to the transmitter (not shown) data clock.
In block 33, the period and phase of the transmitter data clock must be estimated very accurately to minimize errors in the measured phase error. For example, an error of 1 per cent in the data clock phase will result in a phase measurement error as large as 0.9 degrees which may not be acceptable. Even though measured zero-crossings are compensated, measurement noise may result in an unreliable data clock unless the data clock is estimated in an optimal manner. The transmitter data clock may be represented as
tk =kT+b, k=0, 1, 2, . . . (7)
where T is the transmitter data clock period and b is the unknown data clock phase. The a priori clock period T is known within a specified tolerance of T. The objective is to obtain estimates T/ and b/ of T and b from the measured zero-crossings.
Suppose si, i=1, 2, . . . , N are the measured and compensated zero-crossings of the frequency deviation function. An estimate of the zero-crossings spaced by multiples of T/ can be written as
si =ki T+b (8)
and ε1 is a time reference which may be a zero-crossing near the center of the signal burst. Values of T/ and b/ are obtained such that the mean-square error between the sets si and s/ i, i=1, 2, . . . , N given by ##EQU1## is minimized. The resulting estimates are ##EQU2##
The receiver data clock synchronized to the transmitter data clock is given as
tk=kT+b; k+ 0, 1, 2, . . . . (13)
If the clock period T is known a priori with sufficient accuracy for the required measurement, or it is required that the measurement include the measurement of phase errors attributable to inaccuracies in T, T would not be estimated. In this case T/ =T in equations (12) and (13) and only the data clock phase is estimated as given by equation (12). The next step, block 35, is to synchronize bit patterns to establish the active time interval of a signal burst over which the phase and amplitude errors are determined and displayed. If a synchronizing pattern such as a preamble or midamble is available, i.e., included in the transmitted signal burst, then the leading and trailing edges of the envelope of the burst obtained from A[k] as given by equation (4) are used to establish the range over which the preamble or midamble may exist. A discrete-time cross-correlation of the detected bit pattern with the known synchronizing pattern is performed to align the two patterns and establish the active interval. If a synchronizing pattern does not exist, then the active interval of the test is centered between the leading and trailing edges of the envelope of the burst.
Knowledge of the clock phase and period, the data sequence and the time interval of interest provide the information needed to mathematically compute the ideal amplitude and phase modulating functions A[k] and φ[k;a]. These computed functions are then compared at block 38 with the corresponding measured values of amplitude and phase to obtain measurements of amplitude and phase errors.
By way of example, synthesis, block 37, of the phase function for continuous-phase-modulated signals (CPM) will be considered here.
The phase function for CPM can be written as ##EQU3## where
a=(. . . , a-1 a0,a 1,a 2, . . . )
with a1 =▒1, ▒3, . . . , ▒(2M-1)
is the data sequence. For binary modulation M=1 and
hi is the modulation index which in general may be a cyclic function of time. For many common modulations such as minimum shift-key (MSK) and Gaussian minimum shift-key (GMSK), h=1/2 (constant). q(t) is the phase pulse-shape function which has the property that ##EQU4## where L is a positive integer. The type of modulation is determined by q(t). Phase pulse response curves for MSK and GMSK.3, L=5 are plotted in FIGS. 9a and 9b, respectively.
After the ideal phase function φ[k;a] is synthesized, it is subtracted from the measurement phase function
to produce the phase difference given as ##EQU5## The phase error is defined as
ε.sub.φ [k]=φ[k;a]-φ[k;a] (18)
i.e. the difference between the received and synthesized ideal phase functions, so that the phase difference is
Θ.sub.φ [k]=ΔΩk+ε.sub.φ [k]+φ1
k=1, 2, . . . , K (19)
ΔΩ is the frequency error and φ1 is the unknown offset phase.
The phase difference Θ.sub.φ [k], has a linear term ΔΩk with slope ΔΩQ and a constant term φ1, that can be estimated by fitting the K values given by equation (19) to a linear regression curve
Θ.sub.φ [k]=ΔΩk+φ1 (20)
The difference between equations (19) and (20) given as
ε.sub.φ [k]=ε.sub.φ[k]+ (ΔΩ-ΔΩ)k+(φ1 -φ1)
k=1, 2, . . . , K (21)
along with statistics of ε.sub.φ [k] is the desired output of the method.
Referring now also to FIGS. 10, 11, 12, 13 and 14, the phase error and other information determined by the above described method is plotted. In FIG. 10, the measured phase difference on a bit-by-bit basis is plotted versus time as curve 103. Curve 103 shows the difference in phase between the ideal phase function and the transmitted phase function for each data bit in a signal burst. Curve 101 is the linear regression of the phase difference plotted versus the data bit number for a data burst. The slope of the linear regression curve 101 represents the frequency error of the transmitted signal. In FIG. 11, curve 111 is a plot of the instantaneous phase error versus time (bit number) for the data bits in a signal burst and represents the instantaneous phase error of the transmitted signal when compared to the ideal signal. FIGS. 12, 13 and 14 are a plot of the measured signal amplitude versus bit number for a signal burst. Curve 121 is the amplitude of the signal burst. Curves 123 and 125 are the upper and lower bounds allowed for the amplitude. Curve 127 is an expanded plot of the rise time of the transmitted signal amplitude and curve 129 is an expanded plot of the fall time of the transmitted amplitude.
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|WO2002087083A1 *||Apr 18, 2002||Oct 31, 2002||Draper Lab Charles S||Digital method and system for determining the instantaneous phase and amplitude of a vibratory accelerometer and other sensors|
|U.S. Classification||375/226, 375/371, 455/226.1, 329/304|
|International Classification||H04L27/20, G01R23/20, G01R29/26, G01R29/06, G01R25/00|
|Cooperative Classification||H04L27/2017, G01R25/00, G01R23/20, G01R29/26|
|European Classification||G01R23/20, H04L27/20C1N, G01R25/00, G01R29/26|
|Mar 13, 1989||AS||Assignment|
Owner name: HEWLETT-PACKARD COMPANY, CALIFORNIA
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST.;ASSIGNORS:BIRGENHEIER, RAYMOND A.;RYAN, RICHARD P.;REEL/FRAME:005043/0252
Effective date: 19890307
|Sep 1, 1994||FPAY||Fee payment|
Year of fee payment: 4
|Sep 21, 1998||FPAY||Fee payment|
Year of fee payment: 8
|Apr 28, 2000||AS||Assignment|
|Jun 15, 2000||AS||Assignment|
|Sep 18, 2002||FPAY||Fee payment|
Year of fee payment: 12
|Oct 2, 2002||REMI||Maintenance fee reminder mailed|