Search Images Maps Play YouTube News Gmail Drive More »
Sign in
Screen reader users: click this link for accessible mode. Accessible mode has the same essential features but works better with your reader.

Patents

  1. Advanced Patent Search
Publication numberUS5363064 A
Publication typeGrant
Application numberUS 08/050,621
Publication dateNov 8, 1994
Filing dateApr 22, 1993
Priority dateApr 24, 1992
Fee statusLapsed
Also published asDE69305912D1, DE69305912T2, EP0568880A1, EP0568880B1
Publication number050621, 08050621, US 5363064 A, US 5363064A, US-A-5363064, US5363064 A, US5363064A
InventorsYasuki Mikamura
Original AssigneeSumitomo Electric Industries, Ltd.
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Preamplifier for optical communication having a gain control circuit
US 5363064 A
Abstract
A phase-inverting amplifier includes an input-stage FET, a load thereof and a gain control circuit. The gain control circuit is provided in parallel with the load, and reduces an effective load resistance to lower an open-loop gain of the phase-inverting amplifier when a current flowing through the load exceeds a predetermined value. The gain control circuit is typically a FET whose gate is biased at a constant voltage. A feedback resistor is provided in a negative feedback path of the phase-inverting amplifier. A bypass circuit is provided in parallel with the feedback resistor, and reduces an effective feedback resistance when a feedback quantity exceeds a predetermined value.
Images(7)
Previous page
Next page
Claims(12)
What is claimed is:
1. A preamplifier for optical communication that amplifies an input signal produced by a photodetector, comprising:
a phase-inverting amplifier including an input-state FET and a load thereof, for amplifying the input signal;
gain control means for increasing a bandwidth for the preamplifier so that the preamplifier bandwidth remains wider than a bandwidth of an optical receiver which includes the preamplifier, the increase being achieved by reducing an effective load resistance to lower an open-loop gain of the phase-inverting amplifier when a current flowing through the load exceeds a first predetermined value;
a feedback resistor provided in a negative feedback path of the phase-inverting amplifier; and
bypass means for reducing an effective feedback resistance when a feedback quantity exceeds a second predetermined value.
2. The preamplifier of claim 1, wherein the gain control means comprises a gain-control FET, a source and a drain of the gain-control FET being connected to respective terminals of the load, and a gate of the gain-control FET being biased at a constant voltage.
3. The preamplifier of claim 1, wherein the gain control means comprises a diode provided in parallel with the load.
4. The preamplifier of claim 1, wherein the bypass means comprises a diode which is provided in parallel with the feedback resistor.
5. A preamplifier for optical communication that amplifies an input signal produced by a photodetector, comprising:
a phase-inverting amplifier including an input-state FET and a load thereof, for amplifying the input signal;
gain control means for reducing an effective load and resistance to lower an open-loop gain of the phase-inverting amplifier when a current flowing through the load exceeds a first predetermined value;
a feedback resistor provided in a negative feedback path of the phase-inverting amplifier; and
bypass means for reducing an effective feedback resistance when a feedback quantity exceeds a second predetermined value, wherein the bypass means comprises an enhancement-type FET, a drain and a gate of the enhancement-type FET being connected to one terminal of the feedback resistor and a source of the enhancement-type FET being connected to the other terminal of the feedback resistor.
6. A preamplifier for optical communication that amplifies an input signal produced by a photodetector, comprising:
a phase-inverting amplifier including an input-state FET and a load thereof, for amplifying the input signal;
gain control means for reducing an effective load resistance to lower an open-loop gain of the phase-inverting amplifier when a current flowing through the load exceeds a first predetermined value;
a feedback resistor provided in a negative feedback path of the phase-inverting amplifier; and
bypass means for reducing an effective feedback resistance when a feedback quantity exceeds a second predetermined value, wherein the bypass means comprises a depletion-type FET, a drain and a source of the depletion-type FET being connected to respective terminals of the feedback resistor, and a gate of the depletion-type FET being biased at a constant voltage.
7. A preamplifier for optical communication that amplifies an input signal produced by a photodetector, comprising:
a phase-inverting amplifier including an input-state FET and a load thereof, for amplifying the input signal;
gain control means for reducing an effective load resistance to lower an open-loop gain of the phase-inverting amplifier when a current flowing through the load exceeds a first predetermined value;
a feedback resistor provided in a negative feedback path of the phase-inverting amplifier; and
bypass means for reducing an effective feedback resistance when a feedback quantity exceeds a second predetermined value, wherein the bypass means comprises a depletion-type FET, a drain and a source of the depletion-type FET being connected to respective terminals of the feedback resistor, and a gate of the depletion-type FET being controlled by an output signal of the phase-inverting amplifier.
8. A preamplifier for optical communication that amplifies an input signal produced by a photodetector, comprising:
a phase-inverting amplifier including an input-state FET and a load thereof, for amplifying the input signal;
voltage dividing means for dividing an output signal of the phase-inverting amplifier;
gain control means for reducing an effective load resistance to lower an open-loop gain of the phase-inverting amplifier when a current flowing through the load exceeds a first predetermined value;
a feedback resistor provided in a negative feedback path of the, phase-inverting amplifier; and
bypass means for reducing an effective feedback resistance when a feedback quantity exceeds a second predetermined value, wherein the bypass means comprises a depletion-type FET, a drain and a source of the depletion-type FET being connected to respective terminals of the feedback resistor, and a gate of the depletion-type FET being controlled by the divided output signal of the voltage dividing means.
9. The preamplifier of claim 8, wherein the voltage dividing means comprises a FET for source-following the output signal, and resistors for dividing the source-followered output signal.
10. The preamplifier of claim 8, wherein the voltage dividing means comprises a FET for source-following the output signal, and a second depletion-type FET, a gate and a source of the second depletion-type FET being connected to each other and a drain of the second depletion-type FET being connected to a source of the source-following FET.
11. The preamplifier of claim 9, wherein the voltage dividing means comprises a second depletion-type FET provided in parallel with one of the resistors, a gate and a source of the second depletion-type FET being connected to each other, and the divided output signal appearing at a drain of the second depletion-type FET.
12. The preamplifier of claim 1, wherein FETs used in the phase-inverting amplifier and the bypass means are GaAs MESFETs.
Description
BACKGROUND OF THE INVENTION

The present invention relates to a preamplifier for use in optical communication receivers.

In general, a transimpedance-type preamplifier (FIG. 10(a)) and a high-impedance-type preamplifier (FIG. 10(b)) are employed in optical communication receivers. For example, these circuits are described in K. Ogawa, "Considerations for Optical Receiver Design," IEEE Journal on Selected Areas in Communications, Vol. SAC-1, No. 3, 1983, pp. 524-532.

Conventionally, repeaters used in a trunk system of a telephone network, which is the main application field of the optical communication, are required to be highly sensitive to amplify weak signals that have been subjected to attenuation through long-distance transmission while maintaining their waveforms.

On the other hand, in recent years, the application fields of the optical communication have expanded, for instance, to the data communication and subscriber systems. To be applicable to such a variety of communication systems, i.e., to accommodate various light sources, wide-range transmission distance, signal attenuation in optical fibers and communication network topology, etc., the development of an optical communication receiver is now desired which, in addition to being highly sensitive, can deal with optical signals having a wide dynamic range.

In the above circumstances, the present inventor proposed a preamplifier as shown in FIG. 11 (see, for instance, "Wide Dynamic Range GaAs Preamplifier IC for Lightwave Transmission," Autumn National Conference of the Institute of Electronics, Information and Communication Engineers, Presentation No. B-743, 1990). As shown in FIG. 11, a preamplifier A has the transimpedance-type basic constitution, and is incorporated in an optical receiver B that corresponds to the front-end portion of a repeater. The preamplifier A produces an output signal VOUT by amplifying an input signal VIN produced by impedance-conversion of a photocurrent that is an output of a photodetector PD receiving an optical signal hυ from a light transmission line. A bypass circuit C is provided to improve the dynamic range. That is, a cascade connection of a transimpedance-type phase-inverting amplifier 1 having a feedback resistor rf and an output buffer circuit 2 is provided between the input and output terminals of the preamplifier A. The source and drain of a field-effect transistor 3 of the bypass circuit C are connected to the respective terminals of the feedback resistor rf. A voltage in proportion to an output level of the output buffer circuit 2 is applied to the gate of the transistor 3 via a level shift circuit 4.

When the input signal VIN having an excessively large amplitude input signal is input to the preamplifier A, the field effect transistor 3 turns on in response to a variation of the output signal VOUT. Therefore, an effective feedback resistance RF (parallel resistance of the feedback resistance rf and the transistor 3) decreases to improve the dynamic range.

Referring to FIG. 12, a specific example of the transimpedance-type preamplifier A of FIG. 11 is described in detail. The preamplifier of FIG. 12 consists of compound semiconductor field-effect transistors (hereinafter referred to as FETs) T1 -T8 such as GaAs MESFETs, level-shift diode groups d1 and d2, resistors r1 and r2 and a feedback resistor rf, and operates on a single supply voltage VDD. The phase-inverting preamplifier 1 is formed by the FETs T1 -T4 and the diode groups d1 and d2, the output buffer circuit 2 is formed by the FETs T5 and T6, and the level shift circuit 4 is formed by the FET T7 and the resistors r1 and r2. The feedback resistor rf and the FET T8 in FIG. 12 correspond to the resistor rf and the FET 3 in FIG. 10, respectively. The threshold voltage VT8 of the FET T8 is -0.5 V, and the bias setting is so made that the gate-source voltage VGS of the FET T8 is lower than -0.5 V and no current flows between the source and drain when no signal or a very small signal (e.g., smaller than -20 dBm) is input.

As the amplitude of the input signal VIN increases in response to the rise of the input optical signal intensity, the voltage level of the output signal VOUT decreases as a result of the phase-inverting amplification and both of the gate voltage VG and the source voltage VS of the FET T8 decrease. Since the source voltage Vs drops more than the gate voltage VG, the gate-source voltage VGS increases. As a result, when the input optical signal intensity exceeds a certain value (e.g., -10 dBm), the FET T8 turns on to reduce the transimpedance. Even if the amplitude of the input signal VIN is further increased, the reduction of the transimpedance causes clipping of voltage variations within the preamplifier. Therefore, the respective FETs T1 -T7 are not so biased as to work in the non-saturation region, which means the increase of the maximum allowable input level (maximum allowable amplitude of the input signal VIN). That is, the preamplifier of FIGS. 11 and 12 can increase the dynamic range by raising the maximum allowable input level.

On the other hand, in the preamplifier described above, the feedback resistor rf needs to have a large value to lower the minimum sensible light intensity within the dynamic range. That is, since the thermal noise <iRF 2 > decreases as the feedback resistance rf increases (see equation (1) below), the minimum sensible light intensity can be lowered. ##EQU1## where RF is the effective feedback resistance between the input and output of the amplifier 1, W is a frequency bandwidth of the preamplifier, T is the temperature and k is the Boltzmann constant.

However, as is seen from equation (2) belows simply increasing the effective feedback resistance RF causes a problem that a frequency bandwidth ωC of the optical receiver B is reduced. ##EQU2## where G is an open-loop gain of the amplifier 1, CT is an input capacitance, and RF is the effective feedback resistance.

Therefore, as is understood from equation (2), the open-loop gain G of the amplifier 1 needs to be increased or the input capacitance needs to be decreased to provide the large effective resistance RF (for the improvement of the minimum sensible light intensity) and secure the sufficiently wide bandwidth ωC of the optical receiver B.

The present inventor proposed a preamplifier shown in FIG. 13 which can increase the open-loop gain G (see, for instance, "High Gain and Broadband GaAs Preamplifier IC's for High Speed Optical Receivers," Autumn National Conference of the Institute of Electronics, Information and Communication Engineers, Presentation No. B-744, 1990). The open-loop gain is increased by employing a current-injection-type circuit using FETs having two different threshold voltages. The preamplifier of FIG. 13 has the basic constitution in which a FET TIN for current injection is provided between the drain of the input-stage FET T1 and the supply voltage VDD of the preamplifier of FIG. 12, and a FET TIS is added to isolate the FET TIN from the FET T2 serving as a load. The feedback resistor rf is provided between the input and output terminals.

In the preamplifier of FIG. 13, the open-loop gain G is increased by current injection from the current injection FET TIN to the input-stage FET T1, to a large value of about 33 dB. However, according to the general principle of constant gain-bandwidth product, the bandwidth ωC decreases, to about 600 MHz, compared to the case of not incorporating the measure for increasing the gain G (e.g., the preamplifier of FIG. 12).

To cope with this problem, a preamplifier of FIG. 14 was developed by combining the advantages of the first conventional preamplifier of FIGS. 11 and 12 which can increase the maximum allowable input level by incorporation of the bypass circuit C and the second conventional preamplifier of FIG. 13 which can lower the minimum sensible light intensity.

However, this simple combination could not provide an optimum preamplifier because the following problems actually occurred. In the preamplifier of FIG. 14, the bypass circuit is formed by adding a switching diode dS (corresponding to the FET T8 in FIG. 12) to the preamplifier of FIG. 13 having the current injection FET TIN. If the frequency characteristic of the preamplifier has a first-order pole, a transimpedance transfer function ZT (s) of an optical receiver including it has a second-order pole and is therefore expressed by equation (3) where ωC is an angular frequency bandwidth of the optical receiver, GO is a d.c. open-loop gain of the preamplifier, ωh is an angular frequency bandwidth of the preamplifier, RF is an effective feedback resistance (i.e., a parallel resistance of the feedback resistor rf and the switching diode dS), and CT is an input capacitance of the preamplifier. ##EQU3##

The problems occur in the following manner. When an excessively large input signal VIN comes in, the bypass circuit operates to reduce the effective feedback resistance RF, which means decrease of ζ in equation (3). Since the decrease of ζ means increase of the feedback quantity, the bandwidth ωC of the optical receiver increases to become close to the upper limit of the bandwidth ωh of the preamplifier as the input signal VIN further increases to further reduce ζ, as seen from equation (2). In this situation, a condition ωC ≧ωh is more likely to be established than in the first conventional case, because, as described above, the bandwidth ωh of the preamplifier whose gain is increased by the current injection is narrower than the first conventional preamplifier of FIG. 12. As a result, as is understood from equation (3), the optical receiver exhibits a peaking-type transimpedance characteristic as shown in FIG. 15(a) and its operation becomes unstable.

With the peaking-type characteristic, a rectangular NRZ (non-return-to-zero) optical signal will cause ringing or oscillation (FIG. 15(b)), so that the output signal VOUT will not assume a waveform faithful to the input signal waveform. In an actual measurement (see FIG. 16), an output signal VOUT having a waveform distortion associated with an oscillation phenomenon was observed in response to a NRZ input optical signal having an average light intensity of -10 dBm and a pulse rate of 622 Mbit/sec.

In summary, when the input signal VIN is not too large, the bandwidth ωh of the preamplifier is sufficiently larger than the bandwidth ωC of the optical receiver (ωhC) and the stable operation is obtained. However, when the amplitude of the input signal VIN becomes too large, ωC increases to become close to ωh or to exceed it, in which case the optical receiver becomes unstable and exhibits, for instance, an oscillation.

SUMMARY OF THE INVENTION

The present invention has been made in view of the above problems in the art, and has an object of providing a preamplifier for optical communication which can increase the frequency bandwidth and the dynamic range at the same time.

According to the invention, a preamplifier for optical communication that amplifies an input signal produced by a photodetector, comprises:

a phase-inverting amplifier including an input-stage FET and a load thereof, for amplifying the input signal;

gain control means for reducing an effective load resistance to lower an open-loop gain of the phase-inverting amplifier when a current flowing through the load exceeds a first predetermined value;

a feedback resistor provided in a negative feedback path of the phase-inverting amplifier; and

bypass means for reducing an effective feedback resistance when a feedback quantity exceeds a second predetermined value.

With the above constitution, when the input signal is small, the open-loop gain of the phase-inverting amplifier is large and the effective feedback resistance is large, so that the preamplifier exhibits a higher light-receiving sensitivity.

On the other hand, when the input signal exceeds a predetermined amplitude, the effective feedback resistance is reduced to increase the maximum allowable input level. Although the effective load resistance is reduced and the open-loop gain of the phase-inverting amplifier is thereby lowered, the bandwidth of the phase-inverting amplifier is increased to always assure a condition that the bandwidth of the phase-inverting amplifier is larger than the bandwidth of the optical receiver. As a result, undesired phenomena such as oscillation and ringing can be prevented.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram showing a generally concept of a preamplifier according to the present invention;

FIG. 2 is a graph showing operation of the preamplifier of FIG. 1;

FIG. 3 is a circuit diagram showing a preamplifier according to a first embodiment of the invention;

FIG. 4 is a circuit diagram showing a preamplifier according to a second embodiment of the invention;

FIG. 5 is a circuit diagram showing a preamplifier according to a third embodiment of the invention;

FIG. 6 is a circuit diagram showing a preamplifier according to a fourth embodiment of the invention;

FIG. 7 is a circuit diagram showing a preamplifier according to a fifth embodiment of the invention;

FIG. 8 is a partial circuit diagram showing a preamplifier according to a sixth embodiment of the invention;

FIG. 9 is a partial circuit diagram showing a preamplifier according to a modification of the preamplifier of FIG. 1;

FIGS. 10(a) and 10(b) are circuit diagrams showing two general types of preamplifiers;

FIG. 11 is a circuit diagram showing general constitution of a first conventional preamplifier;

FIG. 12 is a circuit diagram showing specific constitution of the preamplifier of FIG. 11;

FIG. 13 is a circuit diagram showing a second conventional preamplifier;

FIG. 14 is a circuit diagram showing a third conventional preamplifier;

FIGS. 15(a) and 15(b) are graphs showing problems of the preamplifier of FIG. 14;

FIG. 16 is a waveform showing a problem of the preamplifier of FIG. 14; and

FIG. 17 is a waveform of an output signal of the preamplifier of FIG. 1.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 is a circuit diagram showing the general constitution of a preamplifier according to the invention. An optical signal h υ transmitted through an optical transmission line is detected by a photodetector PD such as a photodiode, and a resultant photocurrent is impedance-converted to an input signal VIN, which is applied to the gate of a FET Q1. The source of the FET Q1 is connected to a low supply voltage VSS and its drain is connected to a high supply voltage VDD via a load resistor ZL having a fixed value. A gain control means is provided in parallel with the load resistor ZL. More specifically, the gain control means consists of a FET QZL whose source and drain are connected to the respective terminals of the load resistor ZL, and a d.c. voltage VBB for biasing the gate of the FET QZL. A first-stage amplifier 100 is constituted of the FETs Q1 and QZL, load resistor ZL and d.c. voltage VBB. A signal Sx appearing at the drain X of the FET Q1 is shifted to a proper bias level by a level shift circuit 101, and fed back to the gate of the FET Q1 via a feedback resistor rf . A bypass circuit 103 is provided in parallel with the feedback resistor rf. An output buffer circuit 102 power-amplifies the output of the level shift circuit 101 to produce an output signal VOUT. The resistance of the bypass circuit 103 changes in accordance with the output level of the level shift circuit 101 to automatically adjust an effective feedback resistance RF (parallel resistance of the feedback resistor rf and the bypass circuit 103).

With additional reference to FIG. 2, the operation of the preamplifier of FIG. 1 is described below. Since the first-stage amplifier 100 is a phase-inverting amplifier, the input signal VIN is amplified, with phase inversion, to produce the signal SX at the drain of the FET Q1.

When the input signal VIN has a small amplitude, a d.c. variation of the output SR of the level shift circuit 101 is small. Therefore, the bypass circuit 103 is kept open, and the effective feedback resistance RF is equal to the feedback resistance rf. With the large feedback resistance rf, a sufficiently high light-receiving sensitivity is obtained when the input signal VIN is very small or has a small amplitude.

When the input signal VIN exceeds a predetermined amplitude VINC and the output SR of the level shift circuit 101 thereby exceeds a preset threshold of the bypass circuit 103, the internal resistance of the bypass circuit 103 decreases, so that the effective feedback resistance RF becomes smaller than the feedback resistance rf . Therefore, even when a large input signal VIN comes in, voltage variations in the preamplifier are clipped to prevent the internal active devices from being biased to operate in the non-saturation region. As a result, the maximum allowable input level can be increased.

As the amplitude of the input signal VIN increases, the voltage at the drain X of the FET Q1 decreases. The source voltage of the FET QZL decreases accordingly and its gate-source voltage VGS increases. Therefore, when the input signal VIN exceeds the predetermined amplitude and the gate-source voltage VGS of the FET QZL exceeds a threshold voltage VTZL, the FET QZL turns on to gradually reduce an effective load resistance RZL (parallel resistance of the load resistance ZL and the FET QZL). As a result, when a large amplitude input signal VIN comes in, the open-loop gain of the first-stage amplifier 100 decreases but its bandwidth ωh increases, so that undesired phenomena such as oscillation and ringing can be prevented. That is, in equation (4), the parameter ζ does not decrease because the increase of the bandwidth ζh of the amplifier 100 compensates for the decrease of the effective feedback resistance RF. Even if the bandwidth ωC of the optical receiver increases according to equation (2), the bandwidth ωh of the preamplifier increases to always satisfy the relationship ωCh to thereby assure stable operation of the optical receiver.

In summary, when the input signal VIN has an amplitude smaller than the predetermined value, the effective feedback resistance RF is kept large to lower the minimum sensible light intensity. When the input signal VIN takes a large amplitude, the effective feedback resistance RF decreases to increase the maximum allowable input level, and the bandwidth ωh of the amplifier 100 increases to prevent unstable operation such as oscillation.

A measurement was performed by applying, to the preamplifier of FIG. 1, the same NRZ rectangular input signal VIN as applied to the conventional preamplifier of FIG. 12. As shown in FIG. 17, a waveform of the output signal VOUT has no oscillation and is faithful to the input signal VIN, and is clearly improved from the waveform of FIG. 16.

Referring to FIG. 3, a first specific embodiment is described below. A preamplifier of this embodiment consists of a first-stage amplifier formed by FETs Q1, QIS, QIN, Q2 and QZL and bias-setting diodes D1 and D2, a level shift circuit formed by FETs Q3 and Q4 and diodes D3 and D4, an output buffer circuit formed by FETs Q5 and Q6, a diode DS as a bypass circuit, and a feedback resistor rf having a high resistance.

The input signal VIN is applied to the gate of the FET Q1. The bias-setting diodes D1 and D2 are connected to the source of the FET Q1, and the FET Q2 serving as a load is connected to the drain of the FET Q1 via the FET QIS. Further, the FET QIN for current injection is connected to the drain of the FET Q1, and the FET QZL, whose gate is biased at a predetermined voltage VBB, is connected to the load FET Q2. The source terminal X of the FET Q2 is connected to the gate of the FET Q3. A signal SR after the level shifting by the diodes D3 and D4 is power-amplified by the FET Q5 having the source-follower connection to become an output signal VOUT. The signal SR is also fed back to the gate of the FET Q1 via the feedback resistor rf and the diode DS.

In this embodiment, when the input signal VIN has a small amplitude, the FET QZL is in an off-state, because it is so biased that its gate-source voltage VGS is lower than the threshold voltage VTZL. Therefore, the effective load resistance RZL for the FET Q1 takes a large value determined by the FET Q2. A resultant large open-loop gain contributes to the improvement of the minimum sensible light intensity. Since the cathode-anode voltage of the diode DS does not reach its on-voltage, the effective feedback resistance RF is equal to the feedback resistance rf, which also contributes to the improvement of the minimum sensible light intensity.

On the other hand, when the input signal VIN is large, the voltage at the source terminal X of the FET Q2 take a low value and the gate-source voltage VGS of the FET QZL is higher than its threshold voltage VTZL. Therefore, the FET QZL is in an on-state, and the effective load resistance RZL for the FET Q1 takes a smaller value. Further, since the diode D5 is in an on-state, the effective feedback resistance RF is smaller than the feedback resistance rf . Therefore, as in the case of FIG. 1, while the open-loop gain of the preamplifier decreases, its bandwidth ωh increases. As a result, the maximum allowable input level is increased without causing an unstable operation such as an oscillation.

Referring to FIG. 4, a second specific embodiment is described below. The parts in FIG. 4 that are the same as or equivalent to those in FIG. 3 are given the same reference symbols. The second embodiment is different from the first embodiment in that instead of the diode DS as the bypass circuit, the drain and source of an enhancement-type FET QDS whose gate and drain are short-circuited are connected to the respective terminals of the feedback resistor rf. The FET has the same functions as the diode DS in FIG. 3. That is, the FET QDS is off when the input signal VIN is small, and is on and serves to reduce the effective feedback resistance RF when the input signal VIN is larger than the predetermined value. Therefore, like the first embodiment, the second embodiment can provide a preamplifier that has a wide dynamic range and operates stably.

Referring to FIG. 5, a third specific embodiment is described below. The parts in FIG. 5 that are the same as or equivalent to those in FIG. 3 are given the same reference symbols. The third embodiment is different from the first embodiment in that instead of the diode D5 serving as the bypass circuit, a depletion-type FET QF is provided in parallel with the feedback resistor rf . That is, the drain of the FET QF is connected to the gate of the input-side FET Q1, its source is connected to the drain of the FET Q4 of the level shift circuit, and its gate is grounded. When the input signal VIN is small, the FET QF is in an off state, because its gate-source voltage is lower than the threshold voltage. When the VIN exceeds the predetermined value and the gate-source voltage exceeds the threshold voltage, the FET QF is turned on. Therefore, as in the case of the first embodiment, the minimum sensible light intensity is lowered and the maximum allowable input level is increased. That is, the third embodiment can also provide a preamplifier that has a wide dynamic range and operates stably.

Referring to FIG. 6, a fourth specific embodiment is described below. The parts in FIG. 6 that are the same as or equivalent to those in FIG. 5 are given the same reference symbols. The fourth embodiment is different from the third embodiment in that a voltage divider formed by a FET Q7 and resistors R1 and R2 is connected to the output terminal where the output signal VOUT appears. The gate bias of the depletion-type FET QF is set by a divided voltage VG of the output signal VOUT. That is, the gate of the FET Q7 is connected to the output terminal, its drain is connected to the supply voltage VDD, and its source is grounded via the resistors R1 and R2. The divided voltage VG produced by the resistors R1 and R2 is applied to the gate of the FET QF. The resistances of the resistors R1 and R2 are so set that the gate bias voltage of the FET QF is always lower than its source bias voltage. Since, as in the case of the third embodiment, the FET QF is automatically switched between the on-state and off-state in accordance with the amplitude of the input signal VIN, the minimum sensible light intensity is lowered and the maximum allowable input level is increased. Thus, the preamplifier has a wide dynamic range and operates stably.

In general, due to variations in their manufacturing process, semiconductor devices have variations in their characteristics. Therefore, in the third embodiment of FIG. 5 where the gate of the FET QF is grounded, the condition of its switching between the on-state and off-state varies with its threshold voltage. In contrast, in the fourth embodiment of FIG. 6 where the divided voltage VG by the resistors R1 and R2 is applied to the gate of the FET QF, variations in the related devices originating from their manufacturing process cancel out each other, so that the switching condition of the FET QF can be stabilized.

Referring to FIG. 7, a fifth specific embodiment is described below. The parts is FIG. 7 that are the same as or equivalent to those in FIG. 6 are given the same reference symbols. The fifth embodiment is different from the fourth embodiment in the following points. The gate of a FET Q9 is connected to the output terminal where the output voltage VOUT appears, and the drain of a FET Q10 is connected to the source of the FET Q9. The gate and source of the FET Q10 are short-circuited and grounded. A voltage appearing at the drain of the FET Q10 serves as a gate bias for the FET QF .

Since the FET Q10 operates merely as a resistor, the same function as the fourth embodiment can be obtained with smaller devices than the case of the voltage division by resistors.

FIG. 8 is a partial circuit diagram showing a sixth specific embodiment. This embodiment is different from the fourth embodiment (FIG. 6) only in that an FET Q11 is added in parallel with the resistor R2. By employing the FET Q11 that is of the same type as the bypass FET QF, characteristics variations of the FETs Q11 and QF due to a temperature change or variations in their manufacturing process cancel out each other.

FIG. 9 is a partial circuit diagram showing a modification of the general circuit diagram of FIG. 1. In this modification, the gain control means is a series connection of diodes D5 -D7. This simplified gain control means has the function similar to that in FIG. 1. The number of diodes may be set suitably to provide a desired gain control function.

Further, in the general circuit diagram of FIG. 1, the supply voltage VBB may be changed in accordance with the input signal VIN.

As described above, according to the invention, the increase of the frequency bandwidth of the internal amplifier enables the provision of the preamplifier that has a wide dynamic range and operates stably. As a result, it becomes possible to provide optical communication receivers which are highly sensitive and capable of receiving wide-dynamic-range optical signals, and which therefore can accommodate various light sources, wide-range transmission distance, signal attenuation in optical fibers and communication network topology, etc. The invention can suitably applied to preamplifiers using GaAs MESFETs, whereby high-speed optical communication receivers can be realized.

Patent Citations
Cited PatentFiling datePublication dateApplicantTitle
US4218613 *Oct 26, 1978Aug 19, 1980Ernst Leitz Wetzlar GmbhAmplifier for electric signals
US4620321 *May 15, 1984Oct 28, 1986Stc PlcOptical fibre receiver
US4623786 *Nov 7, 1984Nov 18, 1986At&T Bell LaboratoriesTransimpedance amplifier with overload protection
US4808810 *Sep 23, 1987Feb 28, 1989At&T And Philips Telecommunications B.V.Preamplifier for an optical receiver
US5025226 *Mar 2, 1990Jun 18, 1991Triquint Semiconductor, Inc.High-gain low-noise amplifier
US5030925 *Mar 15, 1990Jul 9, 1991Triquint Semiconductor, Inc.Transimpedance amplifier
US5216386 *Dec 20, 1991Jun 1, 1993Honeywell Inc.Transimpedance amplifier
EP0382373A2 *Jan 24, 1990Aug 16, 1990Hewlett-Packard CompanyNonlinear noninverting transimpedance amplifier
Non-Patent Citations
Reference
1IEEE Journal on selected areas in communications, vol. SAC. 1 No. 3. Apr., "Consideration for Optical Receivers Design," Kinichiro Ogawa, Member IEEE. pp. 524-532.
2 *IEEE Journal on selected areas in communications, vol. SAC. 1 No. 3. Apr., Consideration for Optical Receivers Design, Kinichiro Ogawa, Member IEEE. pp. 524 532.
3 *Presentation at Autumn National conference of the Institute of Electronics, Information and Communication Engineers 1990, S. Inano et al, B 744, High Gain and Broadband GaAs Preamplifer IC s for High Speed Optical Receivers.
4Presentation at Autumn National conference of the Institute of Electronics, Information and Communication Engineers 1990, S. Inano et al, B-744, High Gain and Broadband GaAs Preamplifer IC's for High Speed Optical Receivers.
5 *Presentation at Autumn National Conference of the Institute of Electronics, Information and Communication Engineers 1990, Yasuki Mikamura et al, B 743, Wide Dynamic Range GaAs Preamplifier IC for Lightwave Transmission.
6Presentation at Autumn National Conference of the Institute of Electronics, Information and Communication Engineers 1990, Yasuki Mikamura et al, B-743, Wide Dynamic Range GaAs Preamplifier IC for Lightwave Transmission.
Referenced by
Citing PatentFiling datePublication dateApplicantTitle
US5455705 *Mar 14, 1994Oct 3, 1995Analog Devices, Inc.Transimpedance amplifier for optical receiver
US5525929 *Nov 22, 1994Jun 11, 1996Nec CorporationTransimpedance amplifier circuit with feedback and load resistor variable circuits
US5532471 *Dec 21, 1994Jul 2, 1996At&T Corp.Optical transimpedance amplifier with high dynamic range
US5539196 *Mar 28, 1995Jul 23, 1996Canon Kabushiki KaishaPhoto-electric conversion apparatus with gain controllable amplifiers
US5572074 *Jun 6, 1995Nov 5, 1996Rockwell International CorporationCompact photosensor circuit having automatic intensity range control
US5625181 *Aug 10, 1994Apr 29, 1997Fujitsu LimitedLight-receipt system with current bias circuit and pre-amplifier for use in optical digital communication
US5646573 *Feb 28, 1995Jul 8, 1997Anadigics, Inc.Automatic gain-control transimpedence amplifier
US5708392 *Feb 16, 1996Jan 13, 1998Maxim Integrated Products, Inc.Method and apparatus for providing limiting transimpedance amplification
US5790295 *Aug 28, 1995Aug 4, 1998Apple Computer, Inc.Circuit for receiving infrared signals
US5821814 *Dec 17, 1996Oct 13, 1998Mitsubishi Denki Kabushiki KaishaNegative feedback amplifier
US5933265 *Apr 22, 1998Aug 3, 1999Sdl, Inc.Optical receiver module for an optical communication transmission system
US6052030 *May 4, 1998Apr 18, 2000Lucent Technologies Inc.Low voltage variable gain amplifier with feedback
US6057736 *Aug 18, 1998May 2, 2000Electronics And Telecommunications Research InstituteGain controlled amplifier
US6127886 *Oct 30, 1997Oct 3, 2000The Whitaker CorporationSwitched amplifying device
US6396614 *Oct 26, 1998May 28, 2002Nec CorporationOptical signal receiving circuit and method for receiving optical signal
US6614312 *Mar 22, 2002Sep 2, 2003Agilent Technologies, Inc.Low noise amplifier and imaging element using same
US6665013 *May 3, 1999Dec 16, 2003California Institute Of TechnologyActive pixel sensor having intra-pixel charge transfer with analog-to-digital converter
US6915083 *Dec 15, 1998Jul 5, 2005Zilog, Inc.Signal receiver having wide band amplification capability
US7053992Mar 4, 2004May 30, 2006Meade Instruments CorporationRangefinder and method for collecting calibration data
US7212749Mar 15, 2005May 1, 2007Zilog, Inc.Signal receiver having wide band amplification capability
US7358818 *Oct 20, 2004Apr 15, 2008Sumitomo Electric Industries, Ltd.Optical receiver for an optical communication
US7414707May 3, 2006Aug 19, 2008Meade Instruments CorporationRangefinder and method for collecting calibration data
US7508497Nov 23, 2004Mar 24, 2009Meade Instruments CorporationRangefinder with reduced noise receiver
CN100542013CNov 16, 2005Sep 16, 2009松下电器产业株式会社Light receiving amplification circuit
EP0817373A2 *Feb 7, 1997Jan 7, 1998Mitsubishi Denki Kabushiki KaishaNegative feedback amplifier
EP0880822A1 *Dec 24, 1996Dec 2, 1998Maxim Integrated ProductsMethod and apparatus providing limiting transimpedance amplification
EP1469595A2 *Feb 7, 1997Oct 20, 2004Mitsubishi Denki Kabushiki KaishaNegative feedback amplifier
WO1997030515A1 *Dec 24, 1996Aug 21, 1997Maxim Integrated ProductsMethod and apparatus providing limiting transimpedance amplification
WO2003084055A2 *Mar 24, 2003Oct 9, 2003Agilent Technologies IncLow noise amplifier and imaging element using same
Classifications
U.S. Classification330/308, 250/214.0AG
International ClassificationH03F3/08, H03G3/30
Cooperative ClassificationH03G3/3084, H03F1/342, H03F3/082
European ClassificationH03F3/08B, H03G3/30F, H03F1/34B
Legal Events
DateCodeEventDescription
Jan 7, 2003FPExpired due to failure to pay maintenance fee
Effective date: 20021108
Nov 8, 2002LAPSLapse for failure to pay maintenance fees
May 28, 2002REMIMaintenance fee reminder mailed
Apr 27, 1998FPAYFee payment
Year of fee payment: 4
Apr 22, 1993ASAssignment
Owner name: SUMITOMO ELECTRIC INDUSTRIES, LTD., JAPAN
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:MIKAMURA, YASUKI;REEL/FRAME:006536/0089
Effective date: 19930316