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Publication numberUS5574506 A
Publication typeGrant
Application numberUS 08/478,052
Publication dateNov 12, 1996
Filing dateJun 7, 1995
Priority dateSep 7, 1989
Fee statusLapsed
Also published asEP0696403A1, EP0696403A4, US5504532, US5729296, WO1994026065A1
Publication number08478052, 478052, US 5574506 A, US 5574506A, US-A-5574506, US5574506 A, US5574506A
InventorsCharles W. Rhodes
Original AssigneeAdvanced Television Test Center, Inc.
Export CitationBiBTeX, EndNote, RefMan
External Links: USPTO, USPTO Assignment, Espacenet
Bi-directional television format digital signal converter with improved luminance signal-to-noise ratio
US 5574506 A
Abstract
A television signal format converter is provided for converting without loss of picture information a television format to or from a given recorder/player format. An interface converts between RGB and luminance/chrominance inputs and between analog and digital inputs. The interface couples a source television format to a plurality of pairs of memories. A clock and control circuit controls addressing of the memories for reading and writing so that conversion is performed with an improved luminance signal-to-noise ratio between a source television format and the format required for a given high definition digital video tape recorder or any other comparable recorder.
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Claims(23)
What is claimed is:
1. An apparatus for bi-directionally converting a television signal between a first television signal format and a second television signal format, the first television signal format having a first field rate and the second television signal format having a second field rate, the apparatus comprising:
a memory for storing data, the data stored in an arrangement representing both the first television signal format and the second television signal format;
first means, responsive to the first field rate, for generating a first synchronizing signal;
second means, responsive to the second field rate, for generating a second synchronizing signal;
dummy data generating means, responsive to the first field rate and the second field rate, for generating dummy data samples for storage with the data stored in memory; and
control means, coupled to the memory and responsive to the first synchronizing signal and the second synchronizing signal, for storing data in the memory in synchronism with the first synchronizing signal and for reading data from the memory in synchronism with the second synchronizing signal when the first television signal format is being converted to the second television signal format, and for storing data in the memory in synchronism with the second synchronizing signal and for reading data from the memory in synchronism with the first synchronizing signal when the second television signal format is being converted to the first television signal format.
2. The apparatus according to claim 1 further comprising a sampling circuit responsive to the first means for sampling the television signal of the first television signal format in synchronism with the first field rate to produce data stored in the memory when the first television signal format is being converted to the second television signal format, the data related to an amplitude of the television signal of the first television signal format.
3. The apparatus according to claim 2, wherein the dummy data samples include data related to an amplitude of the first television signal.
4. The apparatus according to claim 1 further comprising format reassembly means, coupled to the memory means and responsive to the first means and the second means, for reassembling the first television signal format from the data read from the memory when the second television signal format is being converted to the first television signal format.
5. The apparatus according to claim 1, wherein the field rate of the first television signal format is 50 fields per second, and the field rate of the second television signal format is 60 fields per second.
6. The apparatus according to claim 1, wherein the second television signal format is compatible with a digital tape recorder.
7. A method of bi-directionally converting a television signal between a first television signal format and a second television signal format, the first television signal format having a first field rate and the second television signal format having a second field rate, the method comprising the steps of:
generating a first synchronizing signal in response to the first field rate;
generating a second synchronizing signal in response to the second field rate;
generating dummy data samples in response to the first and second field rates;
storing first data samples in a memory in synchronism with the first synchronizing signal and reading the first data samples from the memory in synchronism with the second synchronizing signal when the first television signal format is being converted to the second television signal format in an arrangement representing both the first television signal format and the second television signal format; and
storing the first data samples in the memory in synchronism with the second synchronizing signal and reading the first data samples from the memory in synchronism with the first synchronizing signal when the second television signal format is being converted to the first television signal format.
8. The method according to claim 7 further comprising the step of storing second data samples in the memory with the first data samples in synchronism with the first synchronizing signal and reading the first data samples and the second data samples from the memory in synchronism with the second synchronizing signal when the first television signal format is being converted to the second television signal format.
9. The method according to claim 8, wherein the second data samples are formed from the first data samples.
10. The method according to claim 7 further comprising the step of storing first data samples and second data samples in the memory in synchronism with the second synchronizing signal and reading the first data samples and the second data samples from the memory when the second television signal format is being converted to the first television signal format, the second data samples being combined with the first data samples when read from the memory to form amplitude data of the television signal of the first television signal format.
11. An apparatus for converting a first television signal format to a second television signal format, the first television signal format having a first field rate and the second television signal format having a second field rate, the apparatus comprising:
a memory for storing data, the data stored in an arrangement representing both the first television signal format and the second television signal format;
first means, responsive to the first field rate, for generating a first synchronizing signal;
second means, responsive to the second field rate, for generating a second synchronizing signal;
dummy data generating means, responsive to the first field rate and the second field rate, for generating dummy data samples for storage with the data stored in memory; and
control means, coupled to the memory and responsive to the first synchronizing signal and the second synchronizing signal, for storing data in the memory in synchronism with the first synchronizing signal and for reading data from the memory in synchronism with the second synchronizing signal when the first television signal format is being converted to the second television signal format.
12. The apparatus according to claim 11 further comprising a sampling circuit responsive to the first means for sampling the first television signal format in synchronism with the first field rate to produce the data stored in the memory, the data related to an amplitude of the television signal of the first television signal format.
13. The apparatus according to claim 12, wherein the dummy data samples include data related to an amplitude of the television signal of the first television signal format.
14. The apparatus according to claim 11, wherein the field rate of the first television signal format is 50 fields per second, and the field rate of the second television signal format is 60 fields per second.
15. The apparatus according to claim 11, wherein the second television signal format is compatible with a digital tape recorder.
16. The apparatus according to claim 11 further comprising format assembly means, coupled to the memory means and responsive to the first means and the second means, for assembling the second television signal format from the data read from the memory.
17. The apparatus according to claim 16 wherein the format assembly means includes a digital-to-analog converter.
18. The apparatus according to claim 11, wherein the field rate of the first television signal format is 60 fields per second, and the field rate of the second television signal format is 50 fields per second.
19. The apparatus according to claim 11, wherein the first television signal format is compatible with a digital tape recorder.
20. A method of converting a first television signal format to a second television signal format, the first television signal format having a first field rate and the second television signal format having a second field rate, the method comprising the steps of:
generating a first synchronizing signal in response to the first field rate;
generating a second synchronizing signal in response to the second field rate;
generating dummy data samples in response to the first and second field rates;
storing first data samples in a memory in synchronism with the first synchronizing signal and reading the first data samples from the memory in synchronism with the second synchronizing signal when the first television signal format is being converted to the second television signal format, the first data samples related to an amplitude of the television signal of the first television signal format and stored in the memory in an arrangement representing both the first television signal format and the second television signal format.
21. The method according to claim 20 further comprising the step of storing second data samples in the memory with the first data samples in synchronism with the first synchronizing signal and reading the first data samples and the second data samples from the memory in synchronism with the second synchronizing signal.
22. The method according to claim 21, wherein the second data samples are formed from the first data samples.
23. The method according to claim 20 further comprising the step of storing second data samples in the memory with the fast data samples in synchronism with the first synchronizing signal and reading the first data samples and the second data samples from the memory in synchronism with the second synchronizing signal, the second data samples being combined with the fast data samples when read from the memory to form amplitude data of the second television signal format.
Description
REFERENCE TO THE PARENT APPLICATION

This application is a division of application Ser. No. 08/053,230, filed Apr. 28, 1993 now U.S. Pat. No. 5,488,424, which is a continuation-in-part application of application Ser. No. 07/404,190, filed Sep. 7, 1989 which is now U.S. Pat. No. 5,280,397.

BACKGROUND OF THE INVENTION

1. Technical Field

This invention relates to high definition television and to a method and apparatus for converting without loss of picture information a television format to and from a given recorder/player format, and more particularly to a method and apparatus for converting a first television format to and from a second television format with an improved luminance signal-to-noise ratio.

2. Related Information

Prior to the introduction in the United States and Japan of the existing NTSC standard television signal format and the introduction in Europe of the PAL and SECAM formats, there was considerable discussion over which broadcast black and white format to choose. Some broadcast formats were desirable because they exhibited less RF interference when transmitted. Other broadcast formats were desirable because they delivered a television picture having a sharper or less choppy image. Discussion again occurred, upon the introduction of color television, over which color signal broadcast format to choose and whether the chosen color format should be compatible with the broadcast black and white television format. In the United States today, the same issues are again being raised over the choice of a second generation television format--commonly known as high definition television (HDTV) or advanced television (ATV). Recent issues include discussion concerning which format is best suited for conventional over-the-air broadcast as well as best suited for transmission over new mediums such as fiber optic cable, coaxial cable, telephonic, broadcast satellite, and pre-recorded mediums. The recent issues include discussion of whether the new HDTV format should be compatible with the present NTSC format. While the Federal Communications Commission of the United States would like the new format to be NTSC compatible, there is considerable influence world-wide to adopt a totally new standard. The discussion further includes the issue of whether a proposed format will be adversely affected during broadcast or transmission on the new mediums and whether such a proposed format will subjectively provide an adequate picture. Many formats have been proposed for adoption by proponents both inside and outside the United States.

The Advanced Television Test Center was established in Alexandria, Va., as a neutral test center for evaluation and comparison of proposed HDTV formats. Besides testing the proposed HDTV formats for immunity to electrical interference and other impairments and testing for NTSC compatibility, the proposed HDTV formats are shown to viewers for subjective evaluation. Viewers compare, by psycho-physical testing, the proposed HDTV formats and rate them as to their characteristics with and without transmission impairments. In order to perform testing on all proposed formats, the television test center has a need for a uniform video tape recorder which can record all proposed source formats.

Such a uniform video source may be provided by a video tape recorder capable of recording any one of all proposed HDTV formats and playing back in the recorded HDTV format. Several digital video tape recorders (DVTR), for example, the Sony HDD-1000 and the Hitachi DVTR, are capable of recording or playing back a HDTV signal. Either the Sony or the Hitachi video tape recorder is capable of recording 1920 bytes of luminance data at 74.25 megabytes per second and two chrominance data components comprising another 1920 bytes per line at 74.25 megabytes per second as packets of data. Digital data packets are recorded at a rate of 1035 lines per frame with each frame having 2 fields. The HDD-1000 has a field rate of 60 Hz at which 74.25 megabytes per second are recorded as mentioned above. The HDD-1000 can also be used at a field rate of 59.94 Hz at which 74.175 megabytes per second are recorded. The field rate of 59.94 Hz is actually 60(1000/1001) Hz. While a DVTR is capable of recording and playing back in accordance with the above-described input/output specifications, a requirement remains to provide an interface for such a recorder allowing other source formats to be recorded uniformly without the introduction of losses.

Prior systems have been proposed for recording at least one format on a single video tape recorder. For example, U.S. Pat. No. 4,549,224, issued to Nakamura et al., provides an apparatus for recognizing either an NTSC or a PAL/SECAM format and generating an appropriate recording frequency dependent on the recognized format. The Nakamura system is incapable of recording proposed high definition television formats. Furthermore, the Nakamura system does not perform signal conversion for recording or playback in a desired format.

Systems are also known for providing conversion between television formats. Many systems, however, are incapable of converting television formats without loss or alteration of picture quality. At the Advanced Television Test Center, conversion between formats must occur without alteration of picture quality or loss of information content. All known conversion systems use interpolation techniques and approximation algorithms for this type of conversion. For instance, U.S. Pat. No. 4,587,556, issued to Collins, discloses a television standard converter for converting conventional PAL and NTSC signals using weighing factors and interpolation. Furthermore, U.S. Pat. No. 4,276,565, issued to Dalton et al., converts conventional television formats using interpolation. U.S. Pat. No. 4,658,284, issued to Kawamura et al., is capable of downsizing a 625 line PAL format to a 525 line format for printing on a color printer. Interpolation is used for the downsizing conversion. Conversion between conventional formats is also performed in U.S. Pat. No. 4,661,862 issued to Thompson, wherein data reduction is performed by deletion and in U.S. Pat. No. 4,283,736, issued to Morio et at., wherein conversion by discarding or repeating information signals is performed. Such schemes entail loss in picture quality or content. Interpolation itself is a faltering function. Some information must be lost and therefore such schemes cannot be truly bi-directional.

Other systems such as that disclosed in U.S. Pat. No. 4,743,958, issued to Bannister et al., convert conventional encoded NTSC, PAL, SECAM and analog RGB, YUV to separate chrominance and luminance signals for input to a special effects device. Bannister et al. performs the conversion using filters for processing the signals. U.S. Pat. No. 4,463,387, issued to Hashimoto et al., processes video data both before recording and after playback for quality improvement. No conversion is performed.

Systems for adapting an input to a VCR include U.S. Pat. No. 4,597,020, issued to Wilkinson, wherein a video signal is shuffled before recording and unshuffled upon playback to disburse bursts of errors. U.S. Pat. No. 4,530,048, issued to Proper, adapts a VCR for computer memory backup storage. Proper concerns interfacing a VCR to avoid VCR information dropouts, a problem of no concern for digital video recording. U.S. Pat. No. 4,638,380, issued to Wilkinson et al., discloses a multiple head video tape recorder with switching and interpolation to remove the effects of a failed head.

Yet another system disclosed in U.S. Pat. No. 4,797,746 to Ashcraft is directed to a system for converting digital image information in a storage format to a format such as a digital scan converter format or standard television format. The disclosed system includes a high bit rate interface unit (HBRI) comprising an input controller, an image buffer RAM, an output controller and a system controller. The HBRI receives stored data from a storage device, such as a tape unit or an optical disk, and based on commands output from the system controller, converts the received data to another format. The system controller commands control the input controller as to the location in the image buffer RAM where each data point or bit is read. The output controller is responsive to the commands of the system controller for reading out each data point from the image buffer RAM so that a standard digital scan converter format or television format is produced.

U.S. Pat. No. 4,577,240 to Hedberg et al. relates to a system capable of on-line acquisition, processing and display of radiographs and for off-line recording, retrieval, review, image processing and archival storage of images. The disclosed system includes word formatter circuits which convert incoming digital data having a width of between 6 and 10 bits into a sequence of 5 separate data streams for recording into a centralized library. The word formatter circuits are used in playback to convert the recorded data for display by HDTV direct imaging or fluoroscopy to produce from quality images.

U.S. Pat. No. 4,651,208 was issued to Charles Rhodes, the inventor of the present application, and was assigned to an assignee other than the assignee of the present application. The patent discloses conversion between widescreen and non-widescreen television transmissions using input/output multiplexers for line conversion in line memory pairs. Picture lines are clocked into and out of a memory to change the aspect ratio of a picture from a large to a small picture aspect ratio. The process is not bi-directional because side panel pixels are discarded when a wide aspect ratio image, such as 16:9, is converted to a less wide aspect ratio image, such as 4:3. Afterwards, the reduced aspect ratio image cannot be reconverted back to the original wide aspect ratio image.

None of the above systems provide a uniform video source for equally comparing proposed high definition television formats. Known interpolators and filters unfortunately reproduce and convert high definition television signals with the sacrifice of picture information. While the invention disclosed in aforementioned U.S. Pat. No. 4,651,208 to Rhodes relates to a method and apparatus for converting widescreen television signals for display on non-widescreen television displays, the process is irreversible and does not facilitate video recording.

SUMMARY OF THE INVENTION

An important object of the present invention is to provide a recorder for recording a television format without loss of picture information.

Another object of the present invention is to provide a player for playing a television format without loss of picture information.

Another object of the present invention is to provide a recorder/player for playing back a television format without loss of picture information.

Another object of the present invention is to provide a converter for converting a television format to digital luminance and chrominance data signals for recording on a digital video recorder without loss of picture information.

Another object of the present invention is to provide a conversion circuit for playback of digital luminance and chrominance signals in a desired television format without loss of picture information.

A further object of the present invention is to provide a converter for allowing a commercially available digital video tape recorder to record and playback a television format.

Another object of the present invention is to provide a converter for storing data from a source television format in a memory in synchronism with the frame rate of the source format and reading the data from the memory in synchronism with the frame rate of a target format.

Another object of the present invention is to provide a converter which includes a plurality of spare lines of data with lines of data read from a memory to generate a complete field of a television signal format.

Another object of the present invention is to provide a converter for bi-directionally converting a television signal between a first television format and a second television signal format with an improved luminance signal-to-noise ratio.

The present invention solves the above-mentioned problems by providing a converter for converting without loss of picture information any high definition television format to and from a given recorder/player format. The format converter of the present invention provides an interface for converting between RGB and luminance/chrominance inputs and between analog and digital inputs. The interface couples any proposed high definition television format to a plurality of memory pairs. A clock and control circuit controls addressing of the memories for reading and writing so that conversion is performed between any high definition television format and the format required for the Sony HDD-1000 or Hitachi digital video tape recorder or any other comparable recorder.

In particular, during a record mode of operation, involving both a converter according to the present invention and a digital video tape recorder, the converter is responsive to horizontal and vertical drive synchronizing signals provided by a source format. The source format signal is digitized and stored in a memory at one rate and read from the memory at the rate of operation of the DVTR. An advantage of the present invention is that the size of the memory may be simultaneously limited and the clocking and controlling of the converter memory facilitated without any loss in the proponent's signal regardless of the input signal format. Furthermore, the reading of converter memory and writing of data into the DVTR are controlled and synchronized by the control signals provided by the source format.

During a playback mode of operation, the converter, according to the present invention, becomes the source of controlling the DVTR and the output to the broadcaster. Nevertheless, the same limited size memory may be employed with the same advantages as during the record mode.

In the event one of the proposed HDTV formats other than those formats based upon the format for which commercially available recorders are designed is adopted by the Federal Communications Commission in the United States, broadcasters will be able to employ the present invention with presently available commercial DVTRs (such as the Sony FIDD-1000 or Hitachi DVTR) to record and playback HDTV pictures and audio--not having to await development of a new HDTV DVTR specifically constructed for the adopted format at an uncertain delay and possibly greater realized cost.

These and other objects and features of the present invention will become evident from the following detailed description of the invention when read in conjunction with the appended drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block schematic diagram of a television test center for testing and comparing proposed HDTV source formats.

FIG. 2 is a block schematic diagram of components for recording a television signal format on a given digital video recorder.

FIG. 3 is a block schematic diagram of components for playing back a signal on a digital video player in a television signal format.

FIG. 4 is a detailed block schematic diagram of circuitry for converting an RGB television signal to digital luminance and chrominance inputs for a given digital tape recorder.

FIG. 5 is a detailed block schematic diagram of components for converting, to an RGB signal, digital luminance and chrominance outputs from a given digital tape player.

FIGS. 6(a) and (b) are record mode and playback mode generic block schematic diagrams, respectively, for clock and control oscillators.

FIG. 7 illustrates seven lines of a first exemplary source television format included in five lines of a digital luminance signal for recording on the digital recorder.

FIG. 8 illustrates seven lines of a first exemplary source television format included in five lines of a digital chrominance signal of the digital recorder.

FIG. 9 illustrates how samples of lines of fields of a first exemplary television format are placed in the lines of fields of the digital video recorder.

FIG. 10 illustrates how samples of the lines of fields of a second exemplary source television format are placed in the lines of fields of the digital video recorder.

FIGS. 11(a) and (b) are flowcharts describing the structuring of converter parameters for a source format.

FIGS. 12(a) and (b) are record mode and playback mode block schematic diagrams, respectively, for clock and control oscillators for a fast exemplary source format.

FIGS. 13(a) and (b) are record mode and playback mode block schematic diagrams, respectively, for clock and control oscillators for a second exemplary source format.

FIGS. 14(a) and (b) are record mode and playback mode block schematic diagrams, respectively, for a clock and control oscillator for a third exemplary source format.

FIG. 15 illustrates how lines of fields of a fourth exemplary source format are arranged in the lines of fields of a digital video recorder.

FIG. 16 is a block schematic diagram of components for recording a fourth exemplary source format on a given digital video recorder with an improved luminance S/N ratio.

FIG. 17 is a detailed block schematic diagram of one embodiment of circuitry for multiplexer circuit 442 of FIG. 16.

FIG. 18 is a detailed block schematic diagram of one embodiment of circuitry for a field processor configured to store the fourth exemplary source format on a given digital video recorder with an improved luminance S/N ratio.

FIG. 19 is a block schematic diagram of components for playing back a signal on a digital video recorder having an improved luminance S/N ratio in the fourth exemplary source format.

FIG. 20 is a detailed block schematic diagram of one embodiment of circuitry for a field processor configured to playback a signal on a digital video recorder having an improved luminance S/N ratio in the fourth exemplary source format.

FIG. 21 is a detailed block schematic diagram of one embodiment of circuitry for field multiplex circuit 590 of FIG. 20.

FIGS. 22(a) and (b) are record mode and playback mode block schematic diagrams, respectively, for a clock and control oscillator for a fourth exemplary source format.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 illustrates a television test center utilizing the multiple HDTV format/recorder players 110 and 150, including converter 160 of the present invention. Signal sources for high-definition television are provided by high-definition digital video tape recorder (DVTR) 110, PIXAR 111 and test signal block 112. Signal sources for test audio are provided by program audio block 113 and test signal audio block 114. Signal sources for the conventional NTSC format are provided by program source block 115, video test signal block 116 and audio test signal block 117. The outputs of the signal sources are directed to test bed 120, including RF test bed 121, satellite and microwave test bed 122 and cable test bed 123. At test bed 120, the television signals are subjected to impairments including noise conditions, multipath, airplane flutter, two channel signal level and time differentials and interference. An impaired television signal is then sent from test bed 120 to displays 130 for psycho-physical testing by viewers 131. An actual or impaired signal output from test bed 120 is also sent to processing equipment 140 for NTSC compatibility tests. The actual or impaired signal output from the test bed can be recorded on high definition television digital video tape recorder (DVTR) 150. Measurement of the HDTV signal can also be made by spectrum analyzer 141 and advanced television audio and visual measurement equipment 142. Digital video tape recorder 150 can also playback directly to high definition television displays 130 for psycho-physical testing.

The present invention provides converters 160 used in conjunction with high definition digital video tape recorders (DVTR) 110 and 150. DVTR 110 plays and DVTR 150 records in only one advanced television format. DVTR 110 is preferably a Sony HDD-1000 or Hitachi digital recorder/player having predetermined input/output specifications substantially in accordance with Japanese high definition broadcast standards; however, the principles of the present invention may be likewise applied to other digital signal recorders having other input/output specifications. Important specifications for the Sony HDD-1000 are illustrated in Table I. Important specifications for the Hitachi DVTR are illustrated in Table II.

                                  TABLE I__________________________________________________________________________PARAMETERS              SPECIFICATIONS__________________________________________________________________________  CODED SIGNALS Y, PB, PR                   THESE SIGNALS ARE OBTAINED FROM GAMMA                   PRE-CORRECTED  OR G, B, R            SIGNALS  NUMBER OF SAMPLES PER Y 2200 G 2200  TOTAL LINE:           P 1100 B 2200  G, B, R OR LUMINANCE SIGNAL                   P 1100 R 2200  EACH COLOR-DIFFERENCE  SIGNAL (PB, PR)  SAMPLING STRUCTURE    ORTHOGONAL LINE, FIELD AND FRAME REPETITIVE.  G, B, R OR LUMINANCE SIGNAL                   THE G, B, R SAMPLING STRUCTURES TO BE COINCIDENT                   AND  EACH COLOR-DIFFERENCE COINCIDENT ALSO WITH THE LUMINANCE SAMPLING                   STRUCTURE  SIGNAL                OF THE Y, PB, PR SYSTEM.                   PB AND PR SAMPLES COSITED WITH ODD                   (1st, 3rd, 5th, ETC.) Y SAMPLES IN EACH LINE.  SAMPLING FREQUENCY:   Y 74.25 MHz                           G 74.25 MHz  G, B, R OR LUMINANCE SIGNAL                   P 37.125 MHz                           B 74.25 MHz  EACH COLOR-DIFFERENCE P 37.125 MHz                           R 74.25 MHz  SIGNAL                (1)                   THE TOLERANCE FOR THE SAMPLING FREQUENCIES                   SHOULD                   COINCIDE WITH THE TOLERANCE FOR THE LINE                   FREQUENCY                   OF THE RELEVANT 1125/60 HDTV STANDARD  FORM OF CODING        UNIFORMLY QUANTIZED PCM, AT LEAST 8 BITS PER                   SAMPLE.  NUMBER OF SAMPLES PER Y 1920  G 1920  DIGITAL ACTIVE LINE:  P 960   B 1920  G, B, R OR LUMINANCE SIGNAL                   P 960   R 1920  EACH COLOR-DIFFERENCE  ANALOG-TO-DIGITAL HORIZONTAL  TIMING RELATIONSHIP:  FROM END OF DIGITAL ACTIVE                   (2)  LINE TO THE HORIZONTAL SYNC                   88 LUMINANCE CLOCK PERIODS  TIMING REFERENCE  CORRESPONDENCE BETWEEN VIDEO  SIGNAL LEVELS AND THE 8 MOST  SIGNIFICANT BITS (MSB) OF THE  QUANTIZATION LEVEL FOR EACH  SAMPLE:  SCALE                 0 TO 255  G, B, R OR LUMINANCE SIGNAL                   220 QUANTIZATION LEVELS WITH THE BLACK LEVEL                   CORRESPONDING TO LEVEL 16 AND THE PEAK WHITE                   LEVEL                   CORRESPONDING TO LEVEL 235.  EACH COLOR-DIFFERENCE THE SIGNAL LEVEL MAY OCCASIONALLY EXCURSE BEYOND  SIGNAL                LEVEL 235.                   225 QUANTIZATION LEVELS IN THE CENTER PART OF                   THE                   QUANTIZATION SCALE WITH ZERO SIGNAL                   CORRESPONDING                   TO LEVEL 128                   THE SIGNAL LEVEL MAY OCCASIONALLY EXCURSE BEYOND                   LEVEL 16 AND 240.  CODE-WORD USAGE FOR THE 8 MOST                   CODE-WORDS CORRESPONDING TO QUANTIZATION LEVELS  SIGNIFCANT BITS (MSB) 0 AND 255 ARE EXCLUSIVELY FOR SYNCHRONIZATION.                   LEVELS 1 TO 254 ARE AVAILABLE FOR__________________________________________________________________________                   VIDEO.

                                  TABLE II__________________________________________________________________________SPECIFICATIONS__________________________________________________________________________GENERAL TAPE TYPE:       1 INCH METAL PARTICLE TAPERECORDING/PLAYBACK TIME: 63 MIN (11 78" TOOL)                    96 MIN. (14 0" TOOL)POWER REQUIREMENTS:      AC                    50/60 HzPOWER CONSUMPTION:TAPE TRANSPORT UNIT:     950 W APPROX.SIGNAL PROCESSOR UNIT:   1300 W APPROX.DIMENSIONS (WHD):TAPE TRANSPORT UNIT:     480  844  745 cms                    (18 7/8  33 1/4  29 5/16 IN)SIGNAL PROCESSOR UNIT:   480  730  615 cms                    (18 7/8  26 3/4  24 3/16 IN)WEIGHT:TAPE TRANSPORT UNIT:     120 kg APPROX. (365 lbs.)SIGNAL PROCESSOR UNIT:   110 kg APPROX. (243 lbs.)AMBIENT TEMPERATURE:     15-35 C.VIDEO SAMPLING FREQUENCYY:                       74.25 MHzρBR :                    37.125 MHzQUANTIZATION:            8 BITS/SAMPLEFREQUENCY BANDWIDTH:Y:                       30 MHz  15 dBρBR :                    15 MHz  15 dBS/K:                     5-6 dBK FACTOR:                LESS THAN 1 (2T PULSE)DIGITAL AUDIO (CH1-CH2)SAMPLING FREQUENCY:      48 MHzQUANTIZATION:            18 BITS/SAMPLEFREQUENCY BANDWIDTH:     20 Hz-20 MHz                    0.5 dB                    10 dBDYNAMIC RANGE:           GREATER THAN 90 dBCROSSTALK:               LESS THAN - 80 dB                    (1 MHz BETWEEN CHANNELS)EMPHASIS:T1:                      50 μS (ON/OFF PERMITTED)T2:                      15 μS (ON/OFF PERMITTED)ANALOG AUDIO (CUE TRACK)FREQUENCY RESPONSE:      100 Hz-12 KHz  3 dBS/K:                     42 dB                    (3% DISTORTION LEVEL)DISTORTION               LESS THAN 3%                    (1 MHz OPERATING LEVEL)INPUT/OUTPUT SIGNALS1. VIDEO INPUT(1) ANALOG SIGNAL:       R, G, B/Y, ρBR SWITCHABLE                    10 V.sub.ρ-ρ  75 OHMS 1 CIRCUIT                    (VIDEO 0.7 V.sub.ρ-ρ, SYNC                    +0.3 V.sub.ρ-ρ)(2) DIGITAL SIGNALY:                       8-BIT PARALLEL (74.25 MHz)ρBA MULTIPLEX 8-BIT PARALLEL                    (74.25 MHz)CLOCK:                   74.25 MHz2. AUDIO INPUT(1) ANALOG SIGNAL (LINE) +4 dB 100 OHMS/                    100K OHMS SWITCHABLE                    BALANCED 8 CIRCUITS(2) ANALOG SIGNAL (CUE)  +4 dB 100 OHMS                    BALANCED 8 CIRCUITS(3) DIGITAL SIGNALCORRECOMMENDATION 647)   XLR  4/O-SUB  1                    SWITCHABLE3. SYNC INPUT:TRI-LEVEL BIPOLAR PULSE SYNC SIGNAL:                    +0.3 V.sub.ρ-ρ  75 OHMS                    1 CIRCUIT4. SMPTE TIME CODE INPUT 2.4 V.sub.ρ-ρ  800 OHMS                    1 CIRCUIT5. VIDEO OUTPUT(1) ANALOG SIGNAL:       R, G, B/Y, ρBR SWITCHABLE                    1.0 V.sub.ρ-ρ  75 OHMS                    2 CIRCUITS                    SWITCHABLE                    (VIDEO 0.7 V.sub.ρ-ρ,                    SYNC + 0.3 V.sub.ρ-ρ)(2) DIGITAL SIGNALY:                       8-BIT PARALLEL                    (74.25 MHz)ρBR MULTIPLEX 8-BIT PARALLEL                    (74.25 MHz)CLOCK:                   74.25 MHz6. AUDIO OUTPUT(1) ANALOG SIGNAL (LINE):                    +4 dB LOW IMPEDENCE                    BALANCED 8 CIRCUITS(2) ANALOG SIGNAL (CUE): +4 dB LOW IMPEDENCE                    BALANCED 1 CIRCUIT(3) DIGITAL SIGNAL(COR RECOMMENDATION 547) XLR  4/O-SUB  1                    SWITCHABLE7. SYNC OUTPUTTRI-LEVEL BIPOLAR PULSE SYNC SIGNAL:                    + 0.3 V.sub.ρ-ρ 75 OHMS                    2 CIRCUITS8. SMPTE TIME CODEOUTPUT:                  2.4 V.sub.ρ-ρ LOW IMPEDENCE                    600 OHMS BALANCED                    1 CIRCUIT9. MONITOR OUTPUT(1) WAVEFORM MONITOR OUTPUT SIGNAL                    1.0 V.sub.ρ-ρ 75 OHMS                    1 CIRCUIT(2) VIDEO MONITOR OUTPUT SIGNAL                    1.0 V.sub.ρ-ρ 75 OHMS                    1 CIRCUIT(3) MONITOR SYNC OUTPUT SIGNAL(TRI-LEVEL):             +0.3 V.sub.ρ-ρ 75 OHMS                    2 CIRCUITS(4) AUDIO MAIN SIGNALMONITOR OUTPUT (FULL):(CHOSEN FROM DA1-DA8 & CUE)                    +4 dB LOW IMPEDENCE                    BALANCED 1 CIRCUIT10. SYNC OUTPUT (525/IO):                    0.3 V.sub.ρ-ρ 75 OHMS                    1 CIRCUIT__________________________________________________________________________

Converter 160, according to the present invention, converts an advanced television or high definition television signal for recording or playback on DVTRs 110 and 150. Converter 160 will be discussed in conjunction with FIGS. 2 and 3 followed by a detailed discussion in conjunction with FIGS. 4-10.

FIG. 2 illustrates converter 160 adapted for recording a television format from HDTV source 210 on digital video recorder 150. HDTV source 210 can be a high definition television camera or other known source such as a specialized high definition television video player. Preferably, RGB (red, green, blue) signals, a FORMAT V-DRIVE IN (vertical) synchronization signal and a FORMAT H-DRIVE IN (horizontal) synchronization signal are provided by HDTV source 210 (FIG. 6(a)). Matrix 211 converts the RGB signals to luminance signal Y and chrominance color difference signals R-Y and B-Y. Matrix 211 is preferably constructed using resistors that combine the RGB signal according to the following equation:

Y=0.731G+0.212R+0.087B

The chrominance signals are found as R-Y and B-Y using resistors and phase inverters. Other chrominance signals can alternatively be converted such as I and Q associated with the video signal. Clock and control circuit 230 clocks analog to digital converter 212 at a frequency fs and 0.5 fs where fs is the sampling clock frequency. The luminance signal is digitized at the sampling frequency fs and the chrominance signals are digitized at the sampling frequency 0.5 fs. Because the two chrominance signals will later be combined into a single chrominance signal, the chrominance signals are digitized at half the rate of the luminance signal.

It is conceivable that HDTV source 210 could provide signals in either analog or digital, RGB, or chrominance and luminance components. Thus, depending upon the particular output of the HDTV source, matrix 211 and analog to digital converter 212 may not be required.

Clock and control circuit 230 also provides control signals to conversion memory 240. Data indicating the high definition television format input by HDTV source 210 for recording on digital video recorder 150 is input to clock and control circuit 230 for programming the converter to provide control signals for the input HDTV format. This data can be input by an operator manually selecting the format to be employed or the format converter can be built for one specific format, for example, any format adopted by the United States. Based on the input HDTV format type and the FORMAT H-DRIVE IN and FORMAT V-DRIVE IN synchronization signals, clock and control circuit 230 outputs read, write and timing signals to conversion memory 240, thus instructing conversion memory 240 to convert the HDTV input format to a format suitable for recording on digital video recorder 150. Conversion memory control 230 controls conversion of the input HDTV format to luminance and chrominance data signals in accordance with FIGS. 9 and 10 which map data into memory 240.

FIG. 3 illustrates converter 160 for converting the output of DVTR 110 in playback mode, to an HDTV format displayed, for example, on display monitor 130. In playback mode, converter 160 controls the operation of the DVTR and provides synchronizing information with the output of the converted video signal. Display 130 can be any output device such as a monitor, a cathode ray robe, liquid crystal display, projection screen, video cassette recorder, or other output, storage, conversion or transmission device. Conversion memory 340 converts the digital signals in accordance with control signals from clock and control circuit 230. Digital to analog converter 312 and matrix 311 perform the reverse functions of matrix 211 and analog to digital converter 212. Further details of conversion memory 340 and digital to analog converter 312 will be discussed subsequently in conjunction with FIG. 5. Further details of clock and control circuit 230 will be discussed in conjunction with FIG. 8.

FIG. 4 illustrates in greater detail the components of FIG. 2. FIG. 4 particularly illustrates matrix 411, anti-aliasing filters 421-423, the individual components of analog to digital converters 424-426, memories 431-436 and multiplexer 440 during record mode. Matrix circuit 411 converts an analog RGB input signal to luminance and color difference signal outputs. Anti-aliasing low pass filters 421, 422 and 423 as well as analog-to-digital converters 424, 425 and 426 are shown connected to the outputs of matrix 411 B-Y, R-Y and Y, respectively. Write clock control signal 401 from controller 230 clocks luminance analog-to-digital converter 426 at fs and chrominance analog-to-digital converters 424 and 425 at 0.5 fs via divide-by-two flip-flop 402.

Two memories are connected to the output of each analog-to-digital converter. Memories 431 and 432 are connected to the output of B-Y chrominance analog-to-digital converter 424 under read/write control 403 by controller 230. Likewise, memories 433 and 434 are connected to the output of R-Y chrominance analog-to-digital converter 425 and memories 435 and 436 are connected to the output of luminance analog-to-digital converter 426 under read/write control 403 by controller 230. Read/write control 403 from controller 230 controls alternatively read and write functions for the first and second memories connected to each analog-to-digital converter. For example, memory 431 alternatively reads and writes with respect to memory 432 under control of read/write control signal 403. Read/write control signal 403 connects directly to the first memory, for example, and read/write control signal 403 inverted by inverter 404 connects to the second memory. Consequently, under control of read/write control signal 403 from controller 230, digital video information can be written into the one memory and simultaneously read out of the other memory. Controller 230 controls addressing of video data written into and read out of the memories thereby performing a desired conversion between formats such as the conversion exemplified by FIGS. 7, 8, 9, and 10. Further details of the read/write clocking and control will later be described in conjunction with FIGS. 6(a) and 6(b).

Multiplexer 440 combines the outputs of memories 431, 432, 433 and 434 to yield a single chrominance signal. In particular, for the Sony HDD-1000 or Hitachi DVTR examples, the output of multiplexer 440 is an 8 bit, byte-interleaved B-Y/R-Y chrominance data bit stream of 1920 bytes per line at a 59.94 Hz field rate. The field rate of 59.94 Hz is actually 60(1000/1001) Hz. Multiplexer 440 is controlled by controller 230 using a 74.175 MHz control signal to match the exemplary HDD-1000 74.175 megabyte per second data rate at a 59.94 Hz field rate. Because the chrominance memory outputs are multiplexed together, each chrominance memory only needs to be half the size of each luminance memory. The outputs of memories 435 and 436 are thus sequentially read to provide a 8-bit luminance data bit stream of 1920 bytes per line and at a 59.94 Hz field rate.

According to the Nyquist criterion, no spectral component of any signal should exceed one-half the digitizing clock frequencies fs for luminance and 0.5 fs for chrominance. Therefore, low pass filters 421-423 shown in FIG. 4 are used in conjunction with analog-to-digital converters 424-426, respectively. The low pass filters can also be called anti-aliasing filters. Low pass filter 423 on the luminance component preferably has a -3 dB drop off at approximately 0.4 fs. Low pass filters 421 and 422 on the two chrominance components should have a -3 dB drop off at approximately 0.2 fs. The filters should not cut off extremely rapidly at the Nyquist frequency of 0.5 fs and 0.25 fs respectively.

FIG. 5 illustrates in greater detail the components of FIG. 3. FIG. 5 particularly illustrates conversion memory 340 (memories 531-536), digital to analog converter 312 (converters 524-526) and matrix 311 (matrix 511) during playback mode. Read/write control 503 from controller 230 controls memory pairs for alternating read and write. Luminance data from, for example, digital video player 110 is alternately written in memories 535 and 536. The chrominance signal from digital video player 110 is de-multiplexed by de-multiplexer 540 using the 74.175 MHz control signal from control circuit 230 for the 59.94 Hz field rate. De-multiplexer 540 splits the chrominance signal into a B-Y signal stored in memories 531 and 532 and a R-Y signal stored in memories 533 and 534. Controller 230 controls the reading and writing from the memories to perform conversion back to the original HDTV format. The conversion controlled by controller 230 is performed in response to a desired input HDTV format programmed into the controller similarly as the control discussed above in conjunction with FIG. 4.

After conversion to the digital HDTV format, the outputs of the memories are respectively fed to digital-to-analog converters 524, 525 and 526. The outputs of the above-mentioned digital-to-analog converters are respectively connected to low pass filter 523 having negligible response at 0.5 fs (about 38 MHz) and low pass filters 522, 521 having negligible response at 0.25 fs (about 19 MHz). The outputs of low pass filters 521, 522 and 523 feed through matrix 511 to produce an RGB high definition television output.

FIGS. 6(a) and 6(b) illustrate the clocking and control circuitry 230 to derive READ CONTROL, READ MEMORY RESET, WRITE CONTROL, WRITE MEMORY RESET and DVTR OUT synchronization control signals. FIG. 6(a) derives the signals for the record mode based on H-DRIVE IN and V-DRIVE IN from a broadcaster or proposed format source. In the record mode, the proposed format sampling rate fs oscillator 931 is locked to the broadcaster or proposed format source via phase locked loop PLL2 940. The DVTR is locked at DVTR frequency 74.175 MHz for the HDD-1000 by oscillator 930 locked to oscillator 931 via phase locked loop PLL1 941. FIG. 6(b) derives the signals for the playback mode wherein DVTR oscillator 930 drives the entire system like the broadcaster or proposed format source drove the entire system in the record mode. DVTR oscillator 930 drives the DVTR via DVTR OUT at 74.175 MHz for the HDD-1000. DVTR oscillator 930 also controls the sampling frequency fs oscillator 931 via phase locked loop PLL3 944.

FIG. 6(a) illustrates a generic depiction of the components for control in the record mode. Oscillator 931 is locked to FORMAT H-DRIVE IN via phase locked loop PLL2 940. Divide-by 967 divides down the output of oscillator 931 by p for a comparison in phase locked loop comparator 940 with the FORMAT H-DRIVE IN signal. The output of PLL2 comparator 940 trims the frequency of crystal oscillator VCXO 931. The output of crystal oscillator VCXO 931 and the output of DVTR crystal oscillator VCXO 930 are compared in phase locked loop PLL1 comparator 941. The output of phase locked loop PLL1 comparator 941 trims the frequency of crystal VCXO 930. The output of oscillator 931 is divided by q in divide-by 966 and the output of oscillator 930 is divided by r in divide-by 960 before comparison in phase locked loop PLL1 comparator 941.

The output of crystal VCXO 930 divided by t in divide-by 972 and divided by s in divide-by 973 is combined by NAND gate 970 and set by flip-flop 980 to yield the READ MEMORY RESET synchronization control signal. The FORMAT H-DRIVE IN and FORMAT V-DRIVE IN are combined by NAND gate 971 and set by flip-flop 981 to yield the WRITE MEMORY RESET synchronization control signal.

The DVTR OUT synchronization control signal of FIG. 6(a) drives the DVTR. The READ CONTROL and WRITE CONTROL synchronization control signals clock memory addresses for i.e. luminance read memory 435 and luminance write memory 436, respectively. Memories 431-436 are addressed by address counters (not shown). The address counters are programmed to count up to the number of necessary storage spaces required for conversion. The READ MEMORY RESET and WRITE MEMORY RESET synchronization control signals reset i.e. luminance read memory 435 and luminance write memory 436, respectively. When a memory is reset by a READ or a WRITE MEMORY RESET synchronization control signal, the signal's associated memory resets to a reference storage space--such as the upper-left comer of the frame.

VCXO oscillators 931 and 930 are preferably crystal oscillators. A crystal oscillator has a crystal ground to produce a very stable and accurate frequency. The oscillation frequency of the crystal can be controlled by a voltage controlled capacitor (varicap) or similar reactive means. Such an accurate crystal oscillator is an ideal oscillator for accurate phase locked loop (PLL) frequency control. As an alternative to crystal VCXO oscillators, microprocessor control oscillators can be used in the event a microprocessor controlled oscillator is available having a frequency stability comparable to that of a crystal VCXO.

FIG. 6(b) illustrates a generic depiction of the components for control in the playback mode. Sampling frequency fs crystal oscillator VCXO 931 and DVTR crystal oscillator VCXO 930 are compared in phase locked loop comparator PLL3 944 to control the frequency of oscillator 931. The output of phase locked loop PLL3 comparator 944 trims the frequency of oscillator 931. The output of oscillator 931 is divided by q in divide-by 966 and the output of oscillator 930 is divided by r in divide-by 960 before comparison in phase locked loop comparator 944.

The DVTR H-DRIVE IN and DVTR V-DRIVE IN are combined in NAND gate 974 and set by flip-flop 982 to yield the WRITE MEMORY RESET synchronization control signal. The output of crystal VCXO 931 divided by 2p in divide-by 976 and divided by n in divide-by 977 is combined in NAND gate 975 and set by flip-flop 983 to yield the READ MEMORY RESET synchronization control signal.

The DVTR OUT synchronization control signal of FIG. 6(b) drives the DVTR. The READ CONTROL and WRITE CONTROL synchronization control signals clock memory addresses for i.e. luminance read memory 535 and luminance write memory 536, respectively. Memories 531-536 are addressed by address counters. The address counters are programmed to count up to the number of necessary storage spaces required for conversion. The READ MEMORY RESET and WRITE MEMORY RESET synchronization control signals reset i.e. luminance read memory 535 and luminance write memory 536, respectively. When a memory is reset by a READ or a WRITE MEMORY RESET synchronization control signal, the signal's associated memory resets to a reference storage space--such as the upper-left comer of the frame.

In the above, the addressed memory can be a random access memory in size dependent upon the number of DVTR lines necessary for a complete sequence of chrominance and luminance lines. Instead of a random access memory, the memory can be a sequential address memory. The READ and the WRITE MEMORY RESET pulses from the NAND gates can be used to reset the sequential access memories to their first memory locations. When using a sequential access memory, the dummy samples illustrated in FIGS. 7 and 8 preferably should be inserted between lines during the time of the horizontal interval. Therefore, from FIG. 7, dummy sample 612 should be inserted between lines 610 and 611 at 612a; dummy sample 614 should be inserted between lines 611 and 613 at 614a; dummy sample 617 should be inserted between either/or lines 613 and 615 or lines 615 and 616 at 617a and/or 617b; dummy sample 619 should be inserted between lines 616 and 618 at 619a; and dummy sample 621 should be inserted between dummy samples 618 and 620 at 621a. By inserting the dummy samples between lines, the dummy samples can be stored in the sequential access memory during the horizontal interval. When using a random access memory, the dummy samples do not need to be placed at the end of a line for they can be randomly accessed or even skipped over by the addressing circuitry at any point in time.

Among the proposed HDTV formats, four exemplary source formats are described herein. These exemplary source formats are examples for teaching the present invention. They are in no way preferred over each other or over any other source format. The first exemplary source format is representable as 1370 luminance (Y) samples per line of video signal and 685 chrominance (R-Y and B-Y) samples per line of video signal. Proper digital representation for the first exemplary source format may require 1370 samples. This format has an active line time of approximately 17.8 microseconds and a RGB signal bandwidth is 28.9 MHz. Accordingly, the minimum number of samples per active line to accurately reproduce the luminance signal is 2.528.917.8 or 1286 samples. The factor 2.5 allows for practical Nyquist frequency filter rolloff, corresponding to the 1/0.4 factor discussed elsewhere in this text.

In accordance with FIGS. 7 and 8, for the fast source HDTV format, the memories 435 and 436 are written into and read out of in a manner corresponding to the 1920 samples per line permitted by the exemplary HDD-1000 DVTR. An entire 1370 sample first luminance line 610, a fast portion of 548 samples of a second luminance line 611 and 2 dummy samples 612 make up DVTR line #1, to be stored in memory 435 or 436. A second portion of 822 samples of the second luminance line 611, a first portion of 1096 samples of a third luminance line 613 and 2 dummy samples 614 make up DVTR line #2. A second portion of 274 samples of the third luminance line 613, an entire 1370 sample fourth luminance line 615 and a first portion of 274 samples of a fifth luminance line 616 and 2 dummy samples 617 make up DVTR line #3. A second portion of 1096 samples of the fifth luminance line 616, a first portion of 822 samples of a sixth luminance line 620 and 2 dummy samples make up DVTR line #4. A second portion of 548 samples of the sixth luminance line 618, an entire 1370 sample seventh luminance line and 2 dummy samples make up DVTR line #5. DVTR line #6 begins with a next entire 1370 sample line, thus repeating the above sequence of seven luminance lines every five DVTR lines. The partitioning of luminance lines and the number of dummy samples are chosen to adapt a given number of luminance lines into a given number of DVTR lines and thus have a predictable repeating pattern, thus minimizing memory size.

Alternatively, the dummy samples may be located between lines, especially in the instance of a sequential access memory. Thus, dummy samples 612 may be located at location 612a, dummy samples 614 at location 614a, and so on during the time of the horizontal blanking interval.

FIG. 8 illustrates an 685 sample R-Y or B-Y chrominance line 710 corresponding to 1370 sample luminance line 610 of the first source format illustrated in FIG. 7. FIG. 8 illustrates the conversion of source lines to DVTR lines output of either memories 433 and 434 for B-Y chrominance lines and memories 431 and 432 for R-Y chrominance lines. Because the R-Y and B-Y chrominance lines are multiplexed together by multiplexer 440, the number of samples of each chrominance source line (685 samples) are half that of each luminance source line (1370 samples). Thus, partitioning of the chrominance source lines and the use of dummy samples in FIG. 8 is similar to FIG. 7.

As described in connection with FIG. 7, dummy samples may be inserted between lines at locations 712a, 714a, 717a, 717b, 719a, 721a and so on.

Depending on the-source format recorded, any number of dummy samples, including zero dummy samples per line, are possible when the partitioning of source lines is chosen. The dummy samples are not necessary for conveying information but may convey useful information by their use as additional luminance and chrominance data, parity bits, as synchronization bits, as memory row identification bits or the like. For example, the dummy samples can be used to indicate the beginning of each new frame. In the event the DVTR should drop data or lose sync, the dummy sample would permit quick correction of the error. Dummy samples can also be used to indicate the beginning of a new frame when the converted format is interlaced at i=2, for example, for correction in the event of an error.

FIG. 9 illustrates how the seven luminance or chrominance source lines for the first exemplary source format are fit into five DVTR lines. DVTR 110, for example, of FIG. 3 dictates the DVTR lines and display 130, for example, dictates the source lines. Note that after a predetermined number of source or DVTR lines, the source and DVTR line pattern repeats.

FIG. 10 illustrates a second exemplary source format having 2055 samples per line. Note that after a predetermined number of fourteen source lines and fifteen DVTR lines, the pattern repeats. The exemplary source format of FIG. 10 also uses 2 dummy samples per DVTR tape line.

FIG. 11(a) is a flowchart illustrating the following method steps 1-11 according to a first embodiment and FIG. 11(b) is a flowchart illustrating the following method steps 1-11 according to a second embodiment. Any proposed format can be realized in a programmable converter as described below:

DIGITAL VIDEO TAPE RECORDER (DVTR) VARIABLES

Given for a DVTR:

1) the number of total lines per frame (NDT),

2) the number of active lines per frame (NDa),

3) the number of total luminance samples (bytes) per line (NDTL),

4) the number of active luminance samples (bytes) per line (NDL),

5) a DVTR field rate (fields per second) equal to the proposed format field rate (fvd =fv),

6) the interlace characteristic of the DVTR [iD 1:1=1 field/frame 1:2=2 fields/frame 1:3=3 fields/frame . . . ], and

7) fsD number of active luminance samples per second

SOURCE FORMAT VARIABLES

Given for a source format:

a) the number of total lines per frame (NT),

b) the number of active lines per frame (Na),

c) a source format field rate (fields per second) equal to the DVTR field rate (fv =fvd),

d) the interlace characteristic of the source format [i 1:1=1 field/frame 1:2=2 fields/frame 1:3=3 fields/frame . . . ], and

e) the minimum number of luminance samples (bytes) per line (NL),

STEPS FOR DERIVING THE NUMBER OF DUMMY SAMPLES ACCORDING TO A FIRST EMBODIMENT

Step #1,

Determine the number of active lines in a field of the DVTR and the number of active lines in a field of the source format.

NDa field =NDa /iD

Na field =Na /i

Step #2,

Determine the number of active luminance samples (bytes) per field for the DVTR

B=NDL NDa field

Step #3,

Determine the number of active luminance samples of the source format samples (bytes) to arrange on a DVTR active line (Nc)

Nc =B/Na field

The result, Nc, will not always be an integer. Nc must be an integer because a digital sample (byte) can only be an integer.

Step #4,

If Nc is not an integer, decrement NDL by 1 and go to step #2.

When Nc becomes an integer, the number of times NDL is decremented, ndec, equals the number of dummy samples that will be used on each line.

Preferably, the integer Nc will be an even number as Nc /2 will be the number of chrominance samples which preferably is an integer. Even numbers will be easier to divide by two in order to clock the 0.5 fs chrominance analog to digital and digital to analog converters. However, if an odd integer Nc is used, the circuitry will tolerate dropping one sample at the edge of every other line. Anti-aliasing low pass falters 421, 422, 521, 522 will also minimize any effects of the dropped half cycle.

Nc thus equals the number of active luminance samples (bytes) 610 of a source active format luminance (Y) line placed on a DVTR line. The number of active samples (bytes) 710 of a source active format chrominance (R-Y) line or (B-Y) line placed on a DVTR line thus equals >Nc because the chrominance component is divided in half by multiplexers 440 and 540 for the chrominance lines. The line arrangements of FIGS. 7 and 8 can thus be determined from the calculated Nc samples (bytes) placed among the repeated lines. The Nc samples are placed among the lines with ndec dummy samples at the end of each line. Eventually a pattern will repeat over and over after a number of lines (five lines for the first exemplary source format, fourteen lines for the second exemplary source format and five lines for the third exemplary source format).

STEPS FOR DERIVING THE NUMBER OF DUMMY SAMPLES ACCORDING TO A SECOND EMBODIMENT

Step #1,

Determine the number of active lines in a field of the DVTR and the number of active lines in a field of the source format.

NDa field =NDa /iD

Na field =Na /i

Step #1A,

Determine the ratio (R) of active lines in a field of the DVTR to active lines in a field of the source format.

R=NDa field /Na field

Step #2,

Find a ratio (j/k) of integers that closely approximate but do not exceed R. The ratio should be selected so that the number of stored lines j is manageable.

The source active format lines will be stored in j lines of memory, each line of memory being NDL samples (bytes) in length. As j directly expresses the size of the memory, it is desirable to minimize the value of j. The smaller the ratio j/k, the lesser the cost of the memory. A value for j of at least about 16 is desirable for inexpensive digital construction.

Step #3,

Determine the number of active luminance samples (bytes) per active DVTR line.

Nc =j NDL /k

If Nc is not an integer, some number d of dummy samples must be added to the real active luminance bytes to comprise a total of NDL active luminance samples (bytes) per active line to be recorded.

Step #4,

If Nc is not an integer, round down to find the largest integer (less than b) which when multiplied by k/j yields an integer i.

When Nc becomes an integer, the number of dummy samples d that will be used on each line is d=NDL -i.

Now that the memory arrangement and number of dummy samples for each DVTR line has been determined by either of the above embodiments, how to provide oscillators for generating the clock and control signals and how to control the oscillators will be discussed below.

STEPS FOR DERIVING fs, p, q and r

The active portion of a television line is approximately 85% of the total line time.

Step #5,

Determine the total number of source format luminance samples to be stored per DVTR line

N=Nc /0.85

The result, N, will not always be an integer. In order for digital generation N, must be an integer.

Step #6,

Round N to the nearest integer.

In effect, by rounding the approximated 85% blanking time is slightly varied up or down until the nearest integer for N is arrived at.

For accurate PLL oscillator control, a small number of low prime integers are desired to divide the oscillator outputs before comparison in the PLL comparator. The preferred integers are the remaining factors derived from the above divided N and from the number of total luminance samples per line NDTL.

Step #7,

Factor NDTL.

Step #8,

Factor N. Check for common factors of NDTL and N.

Step #9,

Increment and decrement N and jump to step #7 until the number of common factors of NDTL and N is greatest.

Alternatively, N NT and NDTL NDT can be factored. Then N can be incremented and decremented until the number of common factors of N NT and NDTL NDT is greatest.

Step #10,

Compute the sampling frequency oscillator rate (fs) in samples per second.

fs =N NT fv

The incrementing and decrementing of N should only be performed within reasonable limits in order to provide a frequency for PLL comparison yielding less error than an oscillator PLL driven by dividers of many uncommon factors. If N=NDTL, the two oscillators will be at the same frequency and a PLL is not required (i.e. 900 line, 59.94, 1:1, n=1200). Choosing a value for N may be performed in an iterative process until the designed circuit achieves an acceptable cost and accuracy tradeoff.

Step #11,

Solve the following equation for q and r by first cancelling common factors. ##EQU1## Then take q as the remaining factors of (N NT /i) and r as the remaining factors of (NDTL NDT /iD).

N itself will be designated p for purposes of FIGS. 6(a) and 6(b), 12(a)-14(b), and 16(a) and 16(b). As discussed in conjunction with FIGS. 6(a) and 6(b), 12(a)-14(b), and 16(a) and 16(b), the remaining factors derived from NDTL and N are thus used as divide by factors p, q, and r for accurate frequency comparison by the phase locked loops.

Also, for purposes of FIGS. 6(a) and 6(b), 12(a)-14(b), and 16(a) and 16(b), t=NDTL, s=fsD /fv, and n=2(NT)(p).

THE DVTR VARIABLES DEFINED

The Sony HDD-1000 DVTR and the Hitachi DVTR have the following parameters:

1) NDT =1125 total lines per frame

2) NDa =1035 active lines per frame

3) NDTL =2200 total luminance samples (bytes) per line

4) NDT =1920 active luminance samples (bytes) per line

5) fv =59.94 fields per second (or 60--operator selectable.)

6) iD =2 fields per frame

7) fsD =74.175 MHz

Other DVTRs can also be used.

STRUCTURING THE CONVERTER PARAMETERS FOR A FIRST EXEMPLARY SOURCE FORMAT ACCORDING TO THE FIRST EMBODIMENT

A first exemplary source format has the following parameters:

a) NT =787.5 total lines per frame

b) Na =720 active lines per frame

c) fv =59.94 fields per second

d) i=1 field per frame

Step #1, NDa field =NDa /iD =1035 active lines per frame/2 fields per frame=517.5 active lines per field, Na field =Na /i=720 active lines per field/1 field per frame=720 active lines per field.

Note that the HDD-1000 DVTR records 1035 active lines per frame as 517 active lines in a first field and 518 active lines in a second field. Nevertheless, NDa field is calculated as 517.5 active lines per field for determining converter parameters according to the present invention. Well-known control circuitry is then used to store the correct number of active lines in the appropriate DVTR field as required by the HDD-1000 DVTR.

Step #2, B=NDL NDa field =(1920 active luminance samples (bytes) per line) (517.5 active lines per field)=993,600 active luminance samples (bytes) per field.

Step #3, Nc =B/Na field =(993,600 active luminance samples (bytes) per field)/(720 active lines per field)=1380 active luminance samples (bytes) per active line.

Step #4, Nc is thus an integer and ndec =0 dummy samples.

Step #5, N=Nc /0.85=(1380 active luminance samples (bytes) per active line)/0.85=1623.529412 total luminance samples (bytes) per active DVTR line.

Step #6, N=1624 total luminance samples (bytes) per active DVTR line, rounded to the nearest integer.

Step #7, NDTL =2200=2225511.

Step #8, N=1624=222729. 1624 factors poorly and has few common factors with NDTL.

Step #9, increment and decrement N until the number of common factors of NDTL and N is greatest. Take the larger factors of 2200, like 11, 5 and 2. Look for numbers close to 1624 (but not 1624) that have factors with 11, 5, 5 and 2 in them. 550=11552. 2200/550=4. 4-1=3. 115523=1650. Therefore, an N of 1650 would work. Also, if factors with 11, 5 and 5 were tried instead, 275=1155, 2200/275=8, 8-1=7 (seven is not a lowest prime to settle at just yet), 8-2=6=32, 11523=1650. Therefore, N=1650 is used.

Step #10, fs =N NT fv =(1650 total samples (bytes) per DVTR line) (787.5 total lines per frame) (59.94 fields per second)=77.88461538 MHz.

Step #11, ##EQU2##

The product of the remaining factors of (NDTL, NDt /iD), 5, 2 and 2, equals 20 and is designated r. The product of the uncommon factors of N, 3 and 7, equals 21 and is designated q. N itself, 1650, is designated p. The control circuitry can be built for the first source format by conforming the circuitry of FIGS. 6(a) and 6(b) with the above calculated variables. The control circuitry of the preferred embodiment for the first exemplary source format as illustrated in FIGS. 12(a) and 12(b) is bunt as calculated according to the second embodiment below. In the second embodiment, when j/k is selected as 5/7 instead of 23/32, a smaller memory size is achieved having two dummy samples. Dummy samples may be necessary when converting an interlaced format and/or for alleviating data errors. In a high volume commercial application, the variables determined by the first and second embodiments may need to be varied in order to achieve the lowest necessary memory capacity for the lowest cost per converter. When designing such a converter, an iterative design approach to using the first and second embodiments is necessary for the best tradeoff.

STRUCTURING THE CONVERTER PARAMETERS FOR A FIRST EXEMPLARY SOURCE FORMAT ACCORDING TO THE SECOND EMBODIMENT

Step #1, NDa field=NDa /id =1035 active lines per frame/2 fields per frame=517.5 active lines per field, Na field =Na /i=720 active lines per field/1 field per frame=720 lines per field.

Step #1A, R=j/k=NDa field /Na field =517.5/720=0.71875

Step #2, j/k=23/32, select a smaller ratio j/k that approximates 23/32 so that the memory size j is at most 16 (design choice for this example). Let j/k=5/7.

Step #3, Nc =(j/k)NDL =(5/7) (1920 active luminance samples (bytes) per line)=1371.428571 active luminance samples (bytes) per line.

Step #4, Nc is not an integer, round down Nc to 1371, (7/5) (1371)=1919.4 which is not an integer, round down Nc to 1370, (7/5) (1370)=1918 which is an integer. Therefore, Nc =1370 and d=1920-1918=2 dummy bytes per line.

Step #5, N=NC /0.85=(1370 active luminance samples (bytes) per line)/0.85=1611.764706 total luminance samples (bytes) per DVTR line.

Step #6, N=1612 total samples (bytes) per DVTR line, rounded to the nearest integer.

Step #7, NDTL =2200=2225511.

Step #8, N=1612=221331. 1624 factors poorly and has few common factors with NDTL.

Step #9, increment and decrement N until the number of common factors of NDTL and N is greatest. Take the larger factors of 2200, like 11, 5 and 2. Look for numbers close to 1612 (but not 1612) that have factors with 11, 5, 5 and 2 in them. 550=11552. 1612/550=2.93. 115523=1650. 1650 is not closest. Try numbers that have 522 in them. 522=20. 2200/20=100. 1612/20=80.6. 52281=1620. Therefore, N=1620 is used.

Step #10, fs =N NT fv =(1620 total samples (bytes) per DVTR line) (787.5 total lines per frame) (59.94 fields per second)=76.468455 MHz.

Step #11, ##EQU3##

The product of the remaining factors of (NDTL NDT /iD), 5, 5, 2 and 11, equals 550 and is designated r. The product of the uncommon factors of N, 3, 3, 3, 3 and 7, equals 567 is designated q. N itself, 1620, is designated p. t=NDTL =2200, s=fsD /fv =1237487, 2p=3240, n=2(NT)(p)=2(787.5)(1620)=2551500. The control circuitry can be built for the first source format by conforming the circuitry of FIGS. 6(a) and 6(b) with the above calculated variables. FIGS. 12(a) and 12(b) show such circuitry connected per the above calculated values.

STRUCTURING THE CONVERTER PARAMETERS FOR A SECOND EXEMPLARY SOURCE FORMAT ACCORDING TO THE FIRST EMBODIMENT

A second exemplary source format has the following parameters:

a) NT =525 total lines per frame

b) Na =483 active lines per frame

c) fv =59.94 fields per second

d) i=1 field per frame

Step #1, NDa field =NDa /iD =1035 active lines per frame/2 fields per frame=517.5 active lines per field, Na field =Na /i=483 active lines per field/1 field per frame=483 active lines per field.

Step #2, B=NDL NDa field =(1920 active luminance samples (bytes) per active line) (517.5 active lines per field)=993,600 active luminance samples (bytes) per field.

Step #3, Nc =B/Na field =(993,600 active luminance samples (bytes) per field)/(483 active lines per field)=2057.142857 active luminance samples (bytes) per active line.

Step #4, Nc is not an integer. Decrement NDL by 1, NDL =1919 active luminance samples (bytes) per line and ndec =1. GO back to step #2.

Step #2, repeated, B=NDL NDa field =(1919 active luminance samples (bytes) per active line) (517.5 active lines per field)=993,082.5 active luminance samples (bytes) per field.

Step #3, repeated, Nc =B/Na field =(993,082.5 active luminance samples (bytes) per field)/(483 active lines per field)=2056.071429 active luminance samples (bytes) per active line.

Step #4, repeated, Nc is not an integer. Again decrement NDL by 1, NDL =1918 active luminance samples (bytes) per active line and ndec =2. Go back to step #2.

Step #2, repeated again, B=NDL NDa field =(1918 active luminance samples (bytes) per line)(517.5 active lines per field)=992,565 active luminance samples (bytes) per field.

Step #3, repeated again, Nc =B/Na field =(992,565 active luminance samples (bytes) per field)/(483 active lines per field)=2055 active luminance samples (bytes) per active line. Therefore, ndec =2 and there will be two dummy samples (bytes). Nc =2055 samples of a source format luminance (Y) line will be placed on the NDL =1920 sample DVTR line by placing 1920 active luminance samples on a first DVTR line and 2055-1920=135 active luminance samples on a second DVTR line. The number of chrominance R-Y placed will be>Nc =1027 samples and chrominance B-Y will be also>Nc =1027 samples (one half a sample is dropped due to an odd Nc).

Step #5, N=Nc /0.85=(2055 active luminance samples (bytes) per active line)/0.85=2417.647059 total luminance samples (bytes) per DVTR line.

Step #6, N=2418 active luminance samples (bytes) per line, rounded to the nearest integer.

Step #7, NDTL =2200=2225511.

Step #8, N=2418=231331.2418 factors poorly and has few common factors with NDTL.

Step #9, increment and decrement N until the number of common factors of NDTL and N is greatest. Take the larger factors of 2200, like 11, 5 and 2. Look for numbers close to 2418 (but not 2418) that have factors with 11, 5, 5 and 2 in them. 550=11552. 2200/550=4. 4+1=5. 115525=2750. Therefore, an N of 2750 would work. But, a number closer to 2418 is preferred. Look for numbers close to 2400 (but not 2400) that have factors with 22255 in them. 200=22255. 2200/200=11. 11-1=10=25. 2225525=2000. 11+1=12=322. 22255322=2400. 2400 is very close to 2418. Therefore N=2400.

Step #10, fs =N NT fv =(2400 total samples (bytes) per DVTR line) (525 total lines per frame) (59.94 fields per second)=75.52447552 MHz.

Step #11, ##EQU4##

The product of the uncommon factors of (NDTL, NDT /iD), 11 and 5, equals 55 and is designated r. The product of the uncommon factors of (N NT /i), 2, 2, 2 and 7, equals 56 and is designated q. N itself, 2400, is designated p. t=NDTL =2200, s=fsD /fv =1237487, 2p=4800, n=2(NT)(p)=2(525)(2400)=2520000. The control circuitry can be built for the second source format by conforming the circuitry of FIGS. 6(a) and 6(b) per the above calculated values. FIGS. 13(a) and 13(b) show such circuitry connected as calculated above except that, to save costs in the preferred embodiment, the values of q and r have been multiplied by ten so that r=550 for both the first and the second exemplary source formats. Multiplying the values of q and r by ten, however, will decrease the frequencies compared at PLL1 941 and PLL3 944 thus decreasing the frequency control stability.

STRUCTURING THE CONVERTER PARAMETERS FOR A SECOND EXEMPLARY SOURCE FORMAT ACCORDING TO THE SECOND EMBODIMENT

Step #1, NDa field =NDa /iD =1035 active lines per frame/2 fields per frame=517.5 active lines per field, Na field =Na /i=483 active lines per field/1 field per frame=483 active lines per field.

Step 1A, R=NDa field /Na field =517.5/483=1.071428571.

Step #2, j/k=15/14. j=15 stored lines is a manageable number.

Step #3, Nc =(j/k) NDL =(15/14) (1920 active luminance samples (bytes) per line)=2057.142857 active luminance samples (bytes) per line.

Step #4, Nc is not an integer. 2057(14/15)=1919.866667. 2056(14/15)=1918.933333. 2055(14/15)=1918. Therefore, Nc =2055 active luminance samples (bytes) per line and d=1920-1918=2 dummy samples per line.

Step #5, N=Nc /0.85=(1918 active luminance samples (bytes) per line)/0.85=2417.647059 total luminance samples (bytes) per DVTR line.

Step #6, N=2418 total samples (bytes) per DVTR line, rounded to the nearest integer.

Step #7, NDTL =2200=2225511.

Step #8, N=2418=231331. 2418 factors poorly and has few common factors with NDTL.

Step #9, increment and decrement N until the number of common factors of NDTL and N is greatest. Take the larger factors of 2200, like 11, 5 and 2. Look for numbers close to 2418 (but not 2418) that have factors with 11, 5, 5 and 2 in them. 550=11552. 2418/550=4.3966. 115524=2200. 115525=2750. But, a number closer to 2418 is preferred. Also, if factors with 11, 5, 2 and 2 were tried instead, 220=11522, 2418/220=10.9909. 1152211=2420. We have few common factors with 11. Look for numbers close to 2418 (but not 2418) that have 2, 2, 2, 2, 2, 3 in them. 222223=96. 2418/96=25.1875. 24=3222, 2222233222.times.=2304. Try for a higher number of common factors. Up 24 to 25. 25=55. We need fives. 22222355=2400.

Step #10, fs =N NT fv =(2400 total samples (bytes) per DVTR line) (525 total lines per frame) (59.94 fields per second)=75.52447552 MHz.

Step #11, ##EQU5##

The product of the common factors of (NDTL, NDT /iD), 11 and 5, equals 55 and is designated r. The product of the uncommon factors of (N NT /i), 2, 2, 2 and 7, equals 56 and is designated q. N itself, 2400, is designated p. t=NDTL =2200, s=fsD /fv =1237487, 2p=4800, n=2(NT)(p)=2(525)(2400)=2520000. The control circuitry can be built for the second source format by conforming the circuitry of FIGS. 6(a) and 6(b) per the above calculated values. FIGS. 13(a) and 13(b) show such circuitry connected as calculated above except that, to save costs in the preferred embodiment, the values of q and r have been multiplied by ten so that r=550 for both the first and the second exemplary source formats. Multiplying the values of q and r by ten, however, will decrease the frequencies compared at PLL1 941 and PLL3 944 thus decreasing frequency control stability.

STRUCTURING THE CONVERTER PARAMETERS FOR A THIRD EXEMPLARY SOURCE FORMAT ACCORDING TO THE FIRST EMBODIMENT

A third exemplary source format has the following parameters:

a) NT =900 total lines per frame

b) Na =828 active lines per frame

c) fv =59.94 fields per second

d) i=1 field per frame

Step #1, NDa field =NDa /iD =1035 active lines per frame/2 fields per frame=517.5 active lines per field, Na field =Na /i=828 active lines per field/1 field per frame=828 active lines per field.

Step #2, B=NDL NDa field =(1920 active luminance samples (bytes) per line) (517.5 active lines per field)=993,600 active luminance samples (bytes) per field.

Step #3, Nc =B/Na field =(993,600 active luminance samples bytes per field)/(828 active lines per field)=1200 active luminance samples (bytes) per active line.

Step #4, Nc is an integer, therefore ndec =0 and there will be no dummy samples (bytes). Nc =1200 active luminance samples of a proposed format luminance (Y) line will be placed on a NDL =1920 active luminance sample DVTR line. The number of chrominance R-Y placed will be>Nc =600 samples and chrominance B-Y will be also>Nc =600 samples.

Step #5, N=Nc /0.85=(1200 active luminance samples (bytes) per line)/0.85=1411.764706 total luminance samples (bytes) per DVTR line.

Step #6, N=1412 total luminance samples (bytes) per DVTR line, rounded to the nearest integer.

Step #7, NDTL =2200=2225511.

Step #8, N=1412=22353. 1412 factors poorly and has few common factors with NDTL.

Step #9, increment and decrement N until the number of common factors of NDTL and N is greatest. Take the larger factors of 2200, like 11, 5 and 2. Look for numbers close to 1412 (but not 2200) that have factors with 11, 5, 5 and 2 in them. 550=11552. 2200/550=4. 4-1=3. 115523=1650. Therefore, an N of 1650 would work. But, a number closer to 1412 is preferred. Also, if factors with 11, 5 and 5 were tried instead, 275=1155, 2200/275=8, 8-1=7 (seven is not a lowest prime to settle at just yet), 8-2=6=32, 115532=1650. If we used 8-3=5 then 11555=1375 which would work better since it is very close to 1412. Therefore, N=1375 is used and in fact yields an fs equal to the DVTR clock, and thus oscillator 931 is not needed for this format. This simplifies the circuitry and eliminates any jitter associated with two phase locked loops in cascade.

Step #10, fs =N NT fv =(1375 total luminance samples (bytes) per DVTR line) (900 total lines per frame) (59.94 fields per second)=74.17582418 MHz.

Step #11, ##EQU6##

There are no remaining factors. Hence q=r=1 and fs =74.17582418 MHz which is the same sampling frequency as the DVTR. Oscillators 930 and 931 can be the same oscillator. N itself, 1375, is designated p. t=NDTL =2200, s=fsD /fv =1237487, 2p=2750, n=2(NT)(p)=2(900)(1375)=2475000. The control circuitry can be built for the second source format by conforming the circuitry of FIGS. 6(a) and 6(b) per the above calculated values. FIGS. 14(a) and 14(b) show such circuitry connected as calculated above.

STRUCTURING THE CONVERTER PARAMETERS FOR A THIRD EXEMPLARY SOURCE FORMAT ACCORDING TO THE SECOND EMBODIMENT

Step #1, NDa field =NDa /iD =1035 active lines per frame/2 fields per frame=517.5 active lines per field, Na field =Na /i=828 active lines per field/1 field per frame=828 active lines per field.

Step 1A, R=NDa field /Na field =517.5/828=0.625.

Step #2, j/k=5/8. j=5 lines is a manageable number.

Step #3, Nc =(j/k) NDL =(5/8)(1920 active luminance samples (bytes) per line)=1200 active luminance samples (bytes) per line.

Step #4, Nc is an integer. Therefore, Nc =1200 active luminance samples (bytes) per line and d=0 dummy samples per line.

Step #5, N=Nc /0.85=(1200 active luminance samples (bytes) per line)/0.85=1411.764706 total luminance samples (bytes) per DVTR line.

Step #6, N=1412 total luminance samples (bytes) per DVTR line, rounded to the nearest integer.

Step #7, NDTL =2200=2225511.

Step #8, N=1412=22353. 1412 factors poorly and has few common factors with NDTL.

Step #9, increment and decrement N until the number of common factors of NDTL and N is greatest. Take the larger factors of 2200, like 11, 5 and 2. Look for numbers close to 1412 (but not 1412) that have factors with 11, 5, 5 and 2 in them. 550=11552. 2200/550=4. 4-1=3. 115523=1650. Therefore, an N of 1650 would work. But, a number closer to 1412 is preferred. Also, if factors with 11, 5 and 5 were tried instead, 275=1155, 2200/275=8, 8-1=7 (seven is not a lowest prime to settle at just yet), 8-2=6=32, 115532=1650. If we used 8-3=5 then 11555=1375 which would work better since it is very close to 1412. Therefore, N=1375 is used and in fact yields an fs equal to the DVTR clock, and thus oscillator 931,932 is not needed for this format.

Step #10, fs =N NT fv =(1375 total samples (bytes) per DVTR line) (900 total lines per frame) (59.94 fields per second)=74.17582418 MHz.

Step #11, ##EQU7##

There are no remaining factors. Hence q=r=1 and fs =74.17582418 MHz which is the same sampling frequency as the DVTR. Oscillators 930 and 931 can be the same oscillator. N itself, 1375, is designated p. t=NDTL =2200, s=fsD /fv =1237487, 2p=2750, n=2(NT)(p)=2(900)(1375)=2475000. The control circuitry can be built for the second source format by conforming the circuitry of FIGS. 6(a) and 6(b) per the above calculated values. FIGS. 14(a) and 14(b) show such circuitry connected as calculated above.

The record mode and playback mode circuits of FIGS. 6(a) and 6(b) can be combined using switches to connect plural components (such as oscillators 931, 932, 933 or 934) according to the desired mode. Furthermore, the circuits of FIGS. 12(a) and 12(b) through 14(a) and 14(b) can be combined with switches to select a format to convert. Furthermore, the dividers can be programmable dividers to aid switching between source formats. The above-mentioned switches and dividers can be substituted with a program controlled processor wherein source format data are entered, parameters are calculated by the above embodiment steps and clock and control signals are generated. It is also important to note that the record/playback circuitry of FIGS. 4 and 5 can share common components. For instance, it is preferred they share the same DVTR and controller. It is also preferred they share the same memories.

A FOURTH EXEMPLARY SOURCE FORMAT AND AN IMPROVEMENT IN LUMINANCE S/N RATIO

A fourth exemplary source format has the following parameters:

a) NT =1250 total lines per frame

b) Na =1152 active lines per frame

c) fv =50 fields per second

d) i=2 fields per frame

The fourth source format, which happens to be the format for European HDTV, has 50 fields per second of interlaced video signal with 625 total lines per field, of which 576 lines are active. The present invention can store this format on a DVTR using a conversion technique which is slightly different from the conversion technique described thus far.

Generally, the conversion technique already described utilizes digital borrowing and carrying of active samples between source format lines to form complete DVTR lines (with any dummy samples, of course). This can be seen in FIG. 7, where DVTR line 1 is shown storing 1370 active samples 610 from line 1 of the first exemplary source format with 548 "borrowed" samples 611 from source line 2. Two dummy samples 612 are stored with these active samples to complete DVTR line 1. In DVTR line 2, 822 "carried" active samples 611 from source format line 2 are stored with 1096 samples 613 from source format line 3 (plus 2 dummy samples 614). DVTR line 3 stores 274 "carried" samples 613 from source format line 3, all 1370 samples 615 of source format line 4, and 274 "borrowed" samples 616 from source format line 5 (plus 2 dummy samples 617). DVTR line 4 stores 1096 "carried" samples 615 from source format line 5 and 822 "borrowed" samples 618 from source format line 6 along with 2 dummy samples 619. DVTR line 5 holds 548 "borrowed" samples 618 from source format line 6 and all 1370 samples 620 of source format line 7, plus 2 dummy samples 621. DVTR line 6 shows the same arrangement of samples as stored in DVTR line 1, indicating that the pattern repeats.

Thus, for the conversion technique of the present invention described thus far, no borrows or carries extend across field boundaries of the source format. That is, the number of samples from a source format line selected to be stored on a DVTR line is such that one complete source format field is stored without borrowing or carrying any samples from an adjacent field.

The conversion technique used by this embodiment of the present invention borrows and carries lines between fields in much the same way as samples are borrowed and carried between lines of the source format. With this approach, 5 fields of this particular source format are conveniently stored within 6 fields of the DVTR format. To facilitate this, the one inactive line per field of this format is treated as active video and recorded. Since the field rate of the DVTR is 1.2 times the field rate of the source format, to store 5 fields of the television format in 6 fields of the DVTR, (576 active lines per source field)/(1.2 DVTR fields per source field)=480 active lines of the television format should be stored in each DVTR field. FIG. 15 shows a sequence of 480 lines from 5 consecutive television format fields being stored in 6 consecutive DVTR fields.

As previously mentioned, one frame of the DVTR has 1035 active lines per frame which are stored as a first field of 517 active lines and a second field of 518 active lines. When the first 480 lines of the fourth proposed format are stored in the first DVTR field, 37 lines of the DVTR field are unused. The next 480 lines of the fourth proposed format, i.e., the remaining 96 lines of the first format field and the first 384 lines of the second field, are stored in the second DVTR field with 38 lines being unused. FIG. 15 shows that as this sequence continues, 192 lines from the second field and 288 lines from the third field of the fourth proposed format are stored in the third DVTR field. The fourth DVTR field holds the last 288 lines from the third field and the first 192 lines from the fourth field of the fourth proposed format. The fifth DVTR field holds the last 384 lines from the fourth field and the first 96 lines from the fifth field of the source format. Finally, the sixth DVTR field holds the remaining 480 lines of the fifth field of the source format.

The unused lines of each DVTR field can be used to carry information for identifying source format fields stored so that the source format can be properly reassembled upon playback. However, the data capacity of these unused lines exceeds requirements for identifying source fields so the available unused lines can be used to improve the signal-to-noise (S/N) ratio of the luminance data. For example, luminance data quantized to 9 bits yields an improvement in S/N ratio of 6 dB over 8-bit luminance data.

To take advantage of this, the available storage in the unused DVTR lines must be considered. The DVTR records 1920 luminance and 1920 chrominance samples (bytes) per line for a total of 3840 bytes per line. Since there are 37 unused lines per each DVTR field (or 38 depending on the DVTR field), there are 142,000 available bytes of unused storage per DVTR field. (The calculation of available storage for a ninth bit of luminance is simplified if the fact that every other DVTR field actually has 38 unused lines is ignored.) This translates into 170,496 unused bytes per source format field, or 1,363,968 bits.

Each source field has 576 active lines with 1920 samples (bytes) of luminance data per active line, yielding a total of 1,105,920 pixels per source field. If each pixel is quantized to 9 bits of luminance data, then 1,105,920 bits of DVTR storage are required for each source field. Since 1,363,968 bits are available in the DVTR format for each source field, a ninth luminance bit can be stored, leaving 258,048 bits, or 32,256 bytes, for source field identification and signalling purposes. These remaining unused bytes can be grouped in any appropriate manner for a given source format. For example, 16,128 bytes can be associated with luminance data and 8064 bytes can be associated with each of Pb and Pr chrominance data. Additional signalling information can be grouped accordingly.

A ninth bit of luminance data can be stored, for example, in the bytes of the unused DVTR field lines at the end of the DVTR field. At playback, each ninth bit is associated with its respective luminance sample. Alternatively, dummy samples containing ninth bit luminance data can be distributed throughout the stored luminance and chrominance data in any appropriate arrangement convenient for reassembly at playback.

FIGS. 16-18 show one embodiment of the present invention for converting a source format to a format compatible with a DVTR and utilizes a ninth luminance bit for improving luminance S/N ratio. FIG. 16 shows the various functional units of this embodiment using the same reference designators as those shown in FIG. 4 where applicable.

A video signal having red, green and blue components is input to matrix 411 to produce corresponding Pr, Pb and Y signal components as outputs. The three outputs of matrix 411 are each filtered by low pass filters 422, 421 and 423, respectively, and then respectively coupled to analog-to-digital converters 425, 424 and 426. Analog-to-digital converters 424 and 425 are both 8-bit converters, while analog-to-digital converter 426 is a 9-bit converter. Analog-to-digital converters 424 and 425 can also have 9-bits of resolution, even though only 8-bits are used. Of course, all three analog-to-digital converters can produce any number of bits greater than 9-bits for economy purposes, even though chrominance data utilizes 8-bits and luminance data utilizes 9-bits. Analog-to-digital converter 426 is clocked at a frequency of fs, while converters 424 and 425 are clocked at 0.5 fs.

The outputs of converters 424-426 are applied to multiplexer circuit 442 to direct the digitized luminance and chrominance data to either odd field processor 450 or even field processor 470 at the field rate of the source format, i.e., 50 Hz for the present embodiment and source format. When the RGB signal applied to matrix 411 is image information from the odd field of the source format, the digitized outputs from converters 424-426 are directed to odd field processor 450. Similarly, when the image information is from the even field of the source format, multiplexer circuit 442 directs the outputs of converters 424-426 to even field processor 470. Multiplexers 440 and 441 select luminance and chrominance data from odd and even field processors 450 and 470 alternately, depending on the current field being processed, for storage by the DVTR. While multiplexer circuit 442 is directing data to odd field processor 450, multiplexers 440 and 441 select even field processor 470 for outputting data to the DVTR. When even field data is being directed to even field processor 470, multiplexers 440 and 441 select odd field processor 450 for outputting data to the DVTR.

FIG. 17 shows details of multiplexer circuit 442. The digitized chrominance outputs of converters 424 and 425 are multiplexed by multiplexer 443 at frequency fs to form a single 8-bit Pr/Pb data stream 446. Demultiplexer 444 directs chrominance data in data stream 446 to either odd or even field processors 450 or 470, respectively, depending on the current field of the source format. Similarly, demultiplexer 445 directs digitized luminance data to odd field processors 450 or even field processor 470.

Details of one embodiment of the field processor 450 are shown in FIG. 18. The details of even field processor 470 are similar and are not specifically discussed or shown in detail. Odd field processor 450 includes shift registers 451 and 452, chrominance field store 456, luminance field store 457, special stores 458 and 459, divide-by-eight counter 454, 8-bit latch 455 and multiplexers 460 and 461. Special stores 458 and 459 are shown in FIG. 18 as separate units for clarity, however, it should be understood, as explained below, they can be combined into a single functional unit since they perform similar functions.

Eight-bit chrominance data is applied to eight 8-stage shift registers 4510 -4517 so that each individual chrominance bit C0 -C7 is applied to a single 8-stage shift register. Simultaneously, nine-bit luminance data is applied to nine 8-stage shift registers 4520 -4528 so that each individual luminance bit Y0 -Y8 is similarly applied to a single 8-stage shift register. Both shift registers 451 and 452 are clocked at the same rate as analog-to-digital converter 426, i.e., fs. Shift registers 451 are first-in/first-out (FIFO) having their individual outputs connected to chrominance field store 456 which holds 480 lines of 1920 eight-bit bytes.

Shift registers 4521 -4528 are also FIFO registers having their respective outputs connected to luminance field store 457 which holds 480 lines of 1920 eight-bit bytes per line. Shift register 4520 is a serial-in/parallel out (SIPO) register having its outputs connected to 8-bit latch 455. Luminance data Y1 -Y8 is clocked through shift registers 452. When all eight registers of SIPO register 4511 are filled, divide-by-eight counter 453 strobes latch 454 to latch bits y0 -y7, the least significant bit of the first eight 9-bit luminance samples. On the next clock cycle of fs, bits Y1 -Y8 of the first luminance sample are stored in luminance field store 457, and the first eight least significant bits (LSB) of the first eight luminance samples, i.e., Y0, is stored in special store 458 as an 8-bit byte. As luminance data continues to be input to shift registers 452, the FIFO registers of shift register 452 outputs luminance samples to luminance field store 457 and the SIPO register 453 accumulates eight LSBs of the luminance samples applied to shift register 452 to form another 8-bit byte. On the eighth count of fs, 8-bit latch 454 latches the output of the SIPO for subsequent storage in special store 458. When special store 458 is filled, the 8-bit bytes of luminance LSBs are directed to special store 459.

Simultaneously, chrominance data is applied to 8-stage shift registers 451, which are FIFO registers. After 8 cycles in shift registers 451, chrominance data is applied to chrominance field store 456. The luminance and chrominance data is applied their respective field stores until both field stores are filled.

When the entire field of the source format has been digitized and input to field stores 456 and 457, each field store will be holding 480 lines of 1920 eight-bit bytes per line, while special stores 458 and 459 each will be holding 30 lines of 1920 eight-bit bytes per line. That is, for every 8 luminance samples stored in luminance field store 457, one 8-bit byte will be stored in either special store 458 or 459.

Luminance and chrominance data is then transferred to the DVTR through multiplexers 460 and 461 to fill 510 lines of the 517 (518) lines per field of the DVTR. First, lines 1-480 from field stores 456 and 457 are each transferred simultaneously to the first 480 lines of the DVTR field. After the first 480 lines are transferred to the DVTR, the 30 lines stored in each of special stores 458 and 459 are selected by multiplexers 460 and 461 to fill lines 481-510 of the DVTR. That is, the 30 lines of luminance LSBs stored in special store 458 are transferred to the DVTR as "luminance" samples, while the 30 lines of luminance LSBs stored in special store 459 are transferred to the DVTR as "chrominance" samples. The remaining 7 unused DVTR lines (or 8 unused lines, depending on the field) can be used to store synchronization or other signalling data, as previously described.

Data is read from the DVTR and converted to the original source format in the reverse order as the order just described. FIGS. 19-21 show one embodiment of the present invention for converting DVTR data to a source format while utilizing a ninth luminance bit for improving luminance S/N ratio. FIG. 19 shows the various functional units of this embodiment using the same reference designators as those shown in FIG. 5 where applicable.

Chrominance and luminance data is input to demultiplexers 540 and 541, respectively, and is directed to odd field processor 550 or even field processor 570, depending on the source format field being processed. When the odd field of source format is being processed, chrominance and luminance data is directed to odd field processor 550. Similarly, data is directed to even field processor 570 when the even field of the source format is being processed. Both field processors operate similarly, so only odd field processor will be described.

The first 480 lines of the 517 (518) line field of the DVTR of chrominance and luminance data is directed to chrominance field store 553 and luminance field store 554, respectively. Demultiplexers 551 and 552 are controlled to direct the next 30 lines of DVTR data to special stores 555 and 556. That is, the 30 lines of luminance LSBs stored as "chrominance" samples are directed to special store 556 and the 30 lines of luminance LSBs stored as "luminance" samples are directed to special store 555.

When chrominance store 553, luminance store 554 and specials stores 555 and 556 are filled, field multiplexer circuit 590 is controlled to pass data from odd field processor 550 to digital-to-analog converters 534-526. Simultaneously, demultiplexers 540 and 541 are controlled to direct chrominance and luminance data from the DVTR to even field processor 570. Even field processor 570 is filled in the same manner as odd field processor 550, described above.

Data is read out of odd field processor by clocking data from chrominance and luminance field stores 553 and 554 into 8-stage shift registers 557 and 558, respectively, at frequency fs. At the eighth clock cycle of fs, data read from special store 555 is latched into 8-bit latch 560 by an output strobe from divide-by-eight counter 561. The output of latch 560 is input to parallel-in/serial-out register 559. As fs continues, chrominance and luminance data is output from shift registers 557 and 558, respectively. Chrominance data is output as a stream of 8-bit bytes, while luminance data is output as a stream of 9-bit bytes.

Field multiplexer circuit 590, shown in greater detail in FIG. 21, includes multiplexers 591 and 593, and demultiplexer 592 for directing data to digital-to-analog converters 534-536. Multiplexer 591 selects chrominance data from either odd field processor 550 or even field processor 570, depending on whether the odd or the even field of the source format is being output from the present invention. Similarly, multiplexer 593 selects luminance data from either odd or even field processors 550, 570. Demultiplexer 592 demultiplexes the chrominance data stream so that the appropriate chrominance data is directed to the correct digital-to-analog converter for reproducing an image in the source format.

As should be understood, special memories 458, 459, 555 and 556 should be configured so that the order data y0 -y7 is stored in special stores 458 and 459 is the reverse order that data y0 -y7 is read out of special stores 555 and 556.

STRUCTURING THE CONVERTER PARAMETERS FOR THE FOURTH EXEMPLARY SOURCE FORMAT ACCORDING TO THE FIRST EMBODIMENT

Using the variables defined for the Sony HDD-1000 DVTR and for the fourth exemplary source format, the following parameters are derived.

Step #1, NDa field =NDa /iD =1035 active lines per frame/2 fields per frame=517.5 active lines per field, Na field =Na /i=1152/2=576 active lines per field.

Step #2, B=NDL NDa field=(1920 active luminance samples (bytes) per active line) (517.5 active lines per field)=993,600 active luminance samples (bytes) per field.

Step #3, Nc =B/Na field =(993,600 active luminance samples (bytes) per field)/(576 active lines per field)=1725 active luminance samples (bytes) per active line.

Step #4, Nc is an integer and ndec =0 dummy samples.

Step #5, N=, Nc /0.85=(1725 active luminance samples (bytes) per active line)/0.85=2029.41 total luminance samples per active DVTR line.

Step #6, N=2029 total luminance samples (bytes) per active DVTR line, rounded to the nearest integer.

Step #7, NDTL =2200=2225511

Step #8, N=2029. 2029 factors very poorly and has no factors in common with NDTL.

Step #9, increment and decrement N until the number of common factors of NDTL and N is greatest. Use the factors of 2200 to find numbers close to 2029 having many of the same factors of 2200. We shall use N=2200 so that 5 source fields are conveniently stored in 6 DVTR fields as described above.

Step #10, fs =N NT fv =(2200 total samples (bytes) per DVTR line)(1250 total lines per frame)(50 fields per second)(1 frame per 2 fields)=68.75 MHz.

Step #11, ##EQU8##

The product of the remaining factors of NDTL NDT /iD, 32, equals 9 and is designated r. The product of the uncommon factors of N and NT, 25, equals 10 and is designated q. N itself, 2200, is designated p. The control circuitry can be built for the fourth source format by conforming the circuitry of FIGS. 6(a) and 6(b) with the above calculated variables as illustrated in FIGS. 22(a) and 22(b).

STRUCTURING THE CONVERTER PARAMETERS FOR A FOURTH EXEMPLARY SOURCE FORMAT ACCORDING TO THE SECOND EMBODIMENT

Step #1, NDa field =NDa /iD =1035 active lines per frame/2 fields per frame=517.5 active lines per field, Na field =Na /i=1152 active lines per field/2 field per frame=576 active lines per field.

Step 1A, R=NDa field /Na field =517.5/576=0.9

Step #2, j/k=9/10. j=9 lines is a manageable number.

Step #3, Nc =(j/k) NDL =(9/10)(1920 active luminance samples (bytes) per line)=1728 active luminance samples (bytes) per line.

Step #4, Nc is an integer. Therefore, Nc =1728 active luminance samples (bytes) per line and d=0 dummy samples per line.

Step #5, N=Nc /0.85=(1728 active luminance samples (bytes) per line)/0.85=2032.94 total luminance samples (bytes) per DVTR line.

Step #6, N=2033 total luminance samples (bytes) per line, rounded to the nearest integer.

Step #7, NDTL =2200=2225511

Step #8, N=2033=19107. Again, 2033 factors very poorly and has no factors in common with NDTL.

Step #9, increment and decrement N until the number of common factors of NDTL and N is greatest. Use the factors of 2200 to find numbers close to 2033 having many of the same factors of 2200. We shall use N=2200 so that 5 source fields are conveniently stored in 6 DVTR fields as described above.

Step #10, fs =N NT fv =(2200 total samples (bytes) per DVTR line) (1250 total lines per frame)(50 fields per second)(1 frame per 2 fields)=68.75 MHz.

Step #11, ##EQU9##

The product of the remaining factors of NDTL NDT /iD, 32, equals 9. and is designated r. The product of the uncommon factors of N and NT, 25, equals 10 and is designated q. N itself, 2200, is designated p. The control circuitry can be built for the fourth source format by conforming the circuitry of FIGS. 6(a) and 6(b) with the above calculated variables as illustrated in FIGS. 22(a) and 22(b).

The above-described television source formats are examples for teaching the present invention. These exemplary source formats are in no way preferred over each other or any other source format. Using the principles taught by the above description, conversions for recording and playing back of any source format are possible.

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Classifications
U.S. Classification348/459, 348/E17.001, 386/E09.01, 348/E05.114, 386/E09.052, 348/E07.003, 348/E09.039, 348/E07.016, 348/441, 348/E09.038, 348/E09.047, 386/E09.012, 386/E09.001, 386/E09.009, 386/E05.012, 386/232
International ClassificationH04N9/804, H04N9/808, H04N5/253, H04N5/92, H04N9/80, H04N9/79, H04N17/00, H04N9/877, H04N5/926, H04N9/67, H04N9/64, H04N17/06, H04N5/46, H04N7/01
Cooperative ClassificationH04N9/804, H04N17/06, H04N9/7921, H04N9/642, H04N5/46, H04N17/00, H04N9/641, H04N9/79, H04N7/0125, H04N7/011, H04N7/01, H04N5/926, H04N5/253, H04N7/0112, H04N9/877, H04N9/7925, H04N9/67
European ClassificationH04N7/01F, H04N7/01B4, H04N5/253, H04N9/804, H04N9/79M2, H04N9/79, H04N7/01, H04N7/01H, H04N9/79M, H04N5/926, H04N9/877
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