|Publication number||US6834084 B2|
|Application number||US 10/139,560|
|Publication date||Dec 21, 2004|
|Filing date||May 6, 2002|
|Priority date||May 6, 2002|
|Also published as||US20030206056|
|Publication number||10139560, 139560, US 6834084 B2, US 6834084B2, US-B2-6834084, US6834084 B2, US6834084B2|
|Inventors||Alexander Wayne Hietala|
|Original Assignee||Rf Micro Devices Inc|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (19), Non-Patent Citations (1), Referenced by (77), Classifications (5), Legal Events (6)|
|External Links: USPTO, USPTO Assignment, Espacenet|
The present invention relates to a transmitter and particularly to a polar modulator in a transmitter.
Transmitters typically contain some sort of baseband processing, followed by a modulator, an amplifier, and an antenna that transmits signals to remote locations. With the proliferation of mobile terminals and wireless LANs, transmitters are becoming more and more common.
In transmitters using linear modulation schemes, the traditional method of realizing the transmit signal has been to use a quadrature modulator to create a signal containing both amplitude and phase components. This signal is then amplified by the amplifier to create the final output signal that passes to the antenna.
The problem with the traditional approach is that it requires a linear power amplifier, which is not as efficient as a non-linear power amplifier operating in saturation. Further, the quadrature modulator must draw significant current to make noise specifications without additional filtering. Still further, the transmit path is not compatible with newer, more efficient GSM transmit methodologies. For example, while a non-linear amplifier might work with a Gaussian minimum-shift keying (GMSK) mode, it would not work with an Enhanced Data Rates for GSM Evolution (EDGE) mode. This hinders the ability to use such approaches in multimode mobile terminals.
One alternative to the quadrature approach is the use of a polar modulator where phase information is passed through a non-linear power amplifier, and the amplitude signal is applied to the power amplifier by a second path. Such polar modulators have problems as well. Specifically, it is difficult to cause the amplitude and phase signals to arrive at the power amplifier at the same time. This is especially true in the analog systems used to date for polar modulated transmitters. Analog components not only have time delays that vary between the paths as a function of the number of components, but also vary as a result of manufacturing tolerances. Thus, no standard time alignment can be used for a transmitter. Instead, each transmitter must have a customized time alignment device, or the tolerances must be so precise that it becomes uneconomical for production. Most polar modulators also still have a quadrature modulator with its attendant current drain.
Thus, there remains a need for better modulators in transmitters.
The present invention uses a polar converter within a polar modulator to create an amplitude signal and a frequency signal, and digitally adjusts the signals so that the frequency and amplitude signals arrive at a power amplifier at the appropriate times. A digital predistortion filter is applied to the frequency signal. The frequency signal is then provided to a single port of a fractional N divider in a phase locked loop. The output of the phase locked loop drives an input of the power amplifier. Meanwhile, the amplitude signal is converted to an analog signal and controls the power supply input of the power amplifier.
In particular, the data representing the signal to be transmitted is received and mapped onto I and Q components. Each I and Q component is filtered and converted to frequency and amplitude signals in a polar coordinate system. The signals are adjusted in amplitude and time. The amplitude signal is converted to an analog signal and ramped up for use at the power amplifier. The frequency signal is digitally filtered and digitally predistorted before being introduced into a fractional N divider of a phase locked loop. The output of the phase locked loop drives the power amplifier.
Those skilled in the art will appreciate the scope of the present invention and realize additional aspects thereof after reading the following detailed description of the preferred embodiments in association with the accompanying drawing figures.
The accompanying drawing figures incorporated in and forming a part of this specification illustrate several aspects of the invention, and together with the description serve to explain the principles of the invention.
FIG. 1 illustrates a mobile terminal such as may use the present invention;
FIG. 2 illustrates a transmit chain according to an exemplary embodiment of the present invention; and
FIG. 3 illustrates an alternate dual-mode embodiment of the present invention.
The embodiments set forth below represent the necessary information to enable those skilled in the art to practice the invention and illustrate the best mode of practicing the invention. Upon reading the following description in light of the accompanying drawing figures, those skilled in the art will understand the concepts of the invention and will recognize applications of these concepts not particularly addressed herein. It should be understood that these concepts and applications fall within the scope of the disclosure and the accompanying claims.
The present invention is preferably incorporated in a mobile terminal 20, such as a mobile telephone, personal digital assistant, or the like. The basic architecture of a mobile terminal 20 is represented in FIG. 1, and may include a receiver front end 22, a radio frequency transmitter section 24, an antenna 26, a duplexer or switch 28, a baseband processor 30, a control system 32, a frequency synthesizer 34, and an interface 36. The receiver front end 22 receives information bearing radio frequency signals from one or more remote transmitters provided by a base station (not shown). A low noise amplifier 37 amplifies the signal. A filter circuit 38 minimizes broadband interference in the received signal, while a downconverter 40 downconverts the filtered, received signal to an intermediate or baseband frequency signal, which is then digitized into one or more digital streams. The receiver front end 22 typically uses one or more mixing frequencies generated by the frequency synthesizer 34.
The baseband processor 30 processes the digitized, received signal to extract the information or data bits conveyed in the received signal. This processing typically comprises demodulation, decoding, and error correction operations. As such, the baseband processor 30 is generally implemented in one or more digital signal processors (DSPs).
On the transmit side, the baseband processor 30 receives digitized data from the control system 32, which it encodes for transmission. The encoded data is output to the radio frequency transmitter section 24, where it is used by a modulator 42 to modulate a carrier signal that is at a desired transmit frequency. Power amplifier 44 amplifies the modulated carrier signal to a level appropriate for transmission from the antenna 26.
As described in further detail below, the power amplifier 44 provides gain for the signal to be transmitted under control of the power control circuitry 46, which is preferably controlled by the control system 32.
A user may interact with the mobile terminal 20 via the interface 36, which may include interface circuitry 48 associated with a microphone 50, a speaker 52, a keypad 54, and a display 56. The interface circuitry 48 typically includes analog-to-digital converters, digital-to-analog converters, amplifiers, and the like. Additionally, it may include a voice encoder/decoder, in which case it may communicate directly with the baseband processor 30.
The microphone 50 will typically convert audio input, such as the user's voice, into an electrical signal, which is then digitized and passed directly or indirectly to the baseband processor 30. Audio information encoded in the received signal is recovered by the baseband processor 30, and converted into an analog signal suitable for driving speaker 52 by the interface circuitry 48. The keypad 54 and display 56 enable the user to interact with the mobile terminal 20, input numbers to be dialed and address book information, or the like, as well as monitor call progress information.
While the present invention is well-suited for incorporation into a mobile terminal, such as the mobile terminal 20 just described, the present invention is also well-suited for use in wireless transmitters associated with wireless LANs and the like. As such, the present invention is not limited to a particular apparatus.
The present invention may be situated in the modulator 42 as illustrated in FIG. 2. Specifically, the modulator 42 may comprise several components, including, a serial interface 60, a mapping module 62, first and second filters 64, 66, a polar converter 68, magnitude adjusters 70, 72, and a time aligner 74. Other components of the modulator 42 will be discussed below.
The serial interface 60 receives Non-Return to Zero (NRZ) serial data from the baseband processor 30 at the bit rate of the system. NRZ data may be a 1B1B code with one line bit for each associated binary bit. In an exemplary embodiment, the modulation scheme for the modulator 42 is an Enhanced Data Rates for GSM Evolution (EDGE) modulation scheme and thus, the bit rate is 812.5 kbps. This data is passed to the mapping module 62, where the data is grouped into symbols of three consecutive data bits, Grey coded, and rotated by 3π/8 on each symbol as per European Telecommunications Standards Institute (ETSI) specifications. The resulting symbol is mapped to one of sixteen points in an I,Q constellation.
Both the I and the Q components for each point are then filtered by the first and second filters 64, 66 respectively. In an exemplary embodiment, the first and second filters 64, 66 are EDGE finite impulse response (FIR) filters. The filters, as dictated by the ETSI specifications, shape the response between symbol times.
After filtering, both the I and the Q components are sent to the polar converter 68 where they are converted into frequency (φ) and amplitude (r) equivalent signals by use of a classical CORDIC (coordinate rotation digital computer). The polar converter 68 also includes a conversion from a true phase signal to a frequency signal. This conversion is well understood in the art and for the purposes of the present invention, this conversion is treated as part of the CORDIC conversion. Further information about CORDIC algorithms may be found in Proceedings of the 1998 ACM/SIGDA Sixth International Symposium On Field Programmable Gate Arrays by Ray Andraka, February 22-24, pp.191-200 and “The CORDIC Trigonometric Computing Technique” by Jack E. Volder IRE Trans on Elect. Computers, p.330, 1959, both of which are hereby incorporated by reference in their entirety.
Magnitude adjusters 70, 72 then adjust the magnitude of the r and φ signals respectively to balance the paths such that they comply with the appropriate standard. Further, a relative time delay is applied to the signals for best Error Vector Magnitude (EVM) and spectrum by the time aligner 74.
At this point the r (amplitude) and φ (frequency) signals separate and proceed by different paths, an amplitude signal processing path and a frequency signal processing path, respectively, to the power amplifier 44. With respect to the amplitude signal processing path, a power ramping function is added by the PA ramp generator 76 by a multiplier 78. The combined signal is then converted to an analog signal by D/A converter 80. The output of the D/A converter 80 is used to set the collector voltage on the power amplifier 44 through a collector regulator 82. As the amplitude signal changes, the voltage at the power amplifier 44 collector changes and the output power will vary as V2/Rout (Rout is not shown, but is effectively the load on the power amplifier 44). This is sometimes known as “plate modulation”.
The φ signal, however, is initially digitally low pass filtered by digital filter 84 and then predistorted by digital predistortion filter 86 before being provided to a fractional N phase locked loop (PLL) 88. In this exemplary embodiment, the signal is applied to a single port on the fractional N divider 89. The digital predistortion filter 86 has approximately the inverse of the transfer function of the PLL 88. For more information about the digital predistortion filter 86, the interested reader is referred to U.S. Pat. No. 6,008,703, which is hereby incorporated by reference in its entirety.
The fractional N PLL 88 has a bandwidth associated therewith. The digital predistortion filter 86 is preferably formed so as to account for this bandwidth. Further, the bandwidth of the fractional N PLL 88 may be calibrated in front of each burst so that the predistortion lines up with the fractional N PLL 88.
In general, the fractional N PLL 88 comprises a reference source 90 that is fed to a phase comparator 92. The phase comparator 92 compares the edges of the reference source 90 to the output of the fractional N divider 89 and produces a correction signal. The correction signal is low pass filtered by filter 94 and input to a voltage controlled oscillator (VCO) 96. The output of the VCO 96 outputs a frequency modulated signal at the RF carrier, which in turn is applied as the signal input of the power amplifier 44 and is also fed back to the fractional N divider 89. The divisor of the fractional N divider 89 is modulated by the distorted φ signal from the digital predistortion filter 86. Further information on fractional N PLLs, how to modulate a signal by varying the fractional N divider 89, and the like may be found in U.S. Pat. Nos. 6,359,950; 6,236,703; 6,211,747; 5,079,522; 5,055,802; and 4,609,881, which are hereby incorporated by reference in their entireties.
It should be appreciated that the fractional N PLL 88 may be replaced with an integer PLL with a translational offset and a wideband digital modulator (neither shown). Antenna 26 then emits electromagnetic radiation corresponding to the output of the power amplifier 44.
By using digital components until just prior to the power amplifier 44, the concerns about the signals arriving at the appropriate times are minimized. This allows the time aligner 74 to provide the appropriate time shift without customization for each analog component.
In the alternate embodiment of FIG. 3, the modulator 42 may switch between EDGE and Gaussian minimum-shift keying (GMSK) modes. Switches 98, 100, and 102 operate in tandem to switch out the polar modulator components and switch in the GMSK processing components. As used herein, the switches 98, 100, and 102 may be any appropriate switching technology such as a transistor switching, a mapping function, or the like, as needed or desired. Specifically, switch 98 takes out the mapping module 62, the filters 64, 66, and the polar converter 68. Instead, the NRZ signal is passed to conventional GMSK processing circuitry 104 and a frequency signal is generated thereby. Exemplary GMSK processing circuitry is discussed in U.S. Pat. No. 5,825,257, which is hereby incorporated by reference in its entirety. It should be appreciated that other GMSK processing circuitry may also be used and the particular circuitry is not central to the present invention. This frequency signal is magnitude adjusted by magnitude adjuster 72 and aligned in time by time aligner 74. The frequency signal is then filtered and predistorted as previously described before being introduced to fractional divider 89 of the fractional N PLL 88. The amplitude signal is set at unity by the step function generator 106, and switch 102 introduces this signal to a multiplier 78A. The multiplier 78A multiplies the amplitude signal by the ramp function, and the output is converted by the D/A 80 for controlling the power supply of the power amplifier 44.
Those skilled in the art will recognize improvements and modifications to the preferred embodiments of the present invention. All such improvements and modifications are considered within the scope of the concepts disclosed herein and the claims that follow.
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|U.S. Classification||375/296, 332/100|
|May 6, 2002||AS||Assignment|
|May 20, 2008||CC||Certificate of correction|
|Jun 6, 2008||FPAY||Fee payment|
Year of fee payment: 4
|Jun 4, 2012||FPAY||Fee payment|
Year of fee payment: 8
|Mar 19, 2013||AS||Assignment|
Owner name: BANK OF AMERICA, N.A., AS ADMINISTRATIVE AGENT, TE
Free format text: NOTICE OF GRANT OF SECURITY INTEREST IN PATENTS;ASSIGNOR:RF MICRO DEVICES, INC.;REEL/FRAME:030045/0831
Effective date: 20130319
|Mar 30, 2015||AS||Assignment|
Owner name: RF MICRO DEVICES, INC., NORTH CAROLINA
Free format text: TERMINATION AND RELEASE OF SECURITY INTEREST IN PATENTS (RECORDED 3/19/13 AT REEL/FRAME 030045/0831);ASSIGNOR:BANK OF AMERICA, N.A., AS ADMINISTRATIVE AGENT;REEL/FRAME:035334/0363
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