|Publication number||US6850121 B1|
|Application number||US 10/018,796|
|Publication date||Feb 1, 2005|
|Filing date||May 30, 2000|
|Priority date||Jun 24, 1999|
|Also published as||CA2377790A1, CN1147991C, CN1370347A, DE19928998A1, DE19928998B4, EP1188228A1, EP1188228B1, WO2001001562A1|
|Publication number||018796, 10018796, PCT/2000/1759, PCT/DE/0/001759, PCT/DE/0/01759, PCT/DE/2000/001759, PCT/DE/2000/01759, PCT/DE0/001759, PCT/DE0/01759, PCT/DE0001759, PCT/DE001759, PCT/DE2000/001759, PCT/DE2000/01759, PCT/DE2000001759, PCT/DE200001759, US 6850121 B1, US 6850121B1, US-B1-6850121, US6850121 B1, US6850121B1|
|Inventors||Volker Detering, Stefan Heinen|
|Original Assignee||Siemens Aktiengesellschaft|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (6), Referenced by (7), Classifications (14), Legal Events (3)|
|External Links: USPTO, USPTO Assignment, Espacenet|
This application is based on and hereby claims priority to German Patent Application No. 19928998 filed on Jun. 24, 1999, the contents of which are hereby incorporated by reference.
1. Field of the Invention
The invention relates to an electronic circuit for generating a transmit frequency for a transceiver.
2. Description of the Related Art
The inventors are familiar with similar circuits from the prior art for generating corresponding transmit frequencies in a TDMA radio system (for example DECT, GSM, PHS). The abbreviation TDMA stands for “Time Division Multiple Access”. A typical circuit is composed of an oscillator for generating frequencies, a transmit amplifier, a receiver and a control device which determines the chronological sequence of alternating transmit and receive states. In general, the oscillator frequency for setting the transmission channel via the control device using a PLL (phase locked loop) is set before the switching on of the transmitter since, for technical reasons, a certain setting time is required for this process. The invention relates to the case of transmission in such a TDMA system as illustrated schematically in FIG. 1.
The problem of such a simple circuit is that the generation of frequencies is disrupted at the moment of the switching on of the transmit amplifier owing to the load change in the amplifier or due to feedback. As a result, an undesired frequency jump is generated. Such a load change occurs, for example, during the switching on of the transmit amplifier as a result of the change in its input impedance. An effect on the generation of frequencies can arise, for example, owing to irradiation by the antenna, or due to other coupling parts between the transmit output stage and the generation of frequencies, for example due to the supply voltage.
In particular in TDMA systems which, for costs reasons, operate with a slow PLL control loop, or open the control loop for the duration of the modulation, this effect is a large problem for the implementation because the frequency jump can no longer be corrected by the PLL circuit. An example of this is the open-loop modulation of a DECT system.
The abovementioned problem is tackled by various circuits known to the inventors. For example, there is a possibility of bringing about a reduction in the load change which is visible for the generation of frequencies by inserting damping elements and isolating stages between the frequency generating components and the transmit amplifier. In addition, additional shielding of the frequency generating components in the form of a Faraday cage can ensure that the irradiation is reduced. Furthermore, additional blocking against electromagnetic irradiation, for example by specially shaped plugs, can be provided on the lines which lead into the shield. An example of such a known circuit device is shown in FIG. 2.
It is also known that the insertion of frequency multiplication stages or divider stages in the frequency generating components prevents the feedback and thus the influence on the frequency generating components. Here, an oscillator oscillates at a harmonic or subharmonic of the desired frequency, as a result of which both a low load dependence and a lower sensitivity to the irradiation of undesired frequencies is produced in accordance with the degree of multiplication or division. This circuit is illustrated schematically in FIG. 3.
Finally, the relatively costly use of a transmission mixing concept, such as is illustrated schematically in
In this transmission mixing concept, the frequencies of two oscillators are mixed in a mixer stage and the desired frequency filtered out from the mixing products. Because the oscillators have a nonharmonic relationship with the desired frequency, there is a resulting high degree of immunity to the load changes and effects. As a result, the requirements made of the shielding, the blocking and the isolation stages are reduced considerably in comparison with the known solutions from
The greatest disadvantage of this transmission mixing concept is the large degree of technical expenditure which it requires because a transmission mixer stage, an oscillator including a PLL circuit for frequency stabilization and a band filter are additionally required. The additionally required electronic components alone result in a considerable cost disadvantage in comparison with the two preceding solutions.
A further disadvantage of this more costly transmission mixing concept is that the overall size of such a circuit is too large owing to the number of additional electronic components.
In this transmission mixing concept, it proves particularly difficult to achieve a high degree of integration because given the current state of the art the filters and oscillators or oscillator coils are very difficult to accommodate in integrated circuits, or require a very large chip area. In addition, it is frequently impossible to integrate to a sufficient degree the capacitors and resistors which are required for the PLL so that they have to be arranged as external components.
Because a total of two oscillators for frequency stabilization, two PLLs, including two external loop filters, are necessary in the known transmission mixing concept, and in particular oscillators with a low frequency require a particularly large chip area or have poor properties with respect to phase noise, this transmission mixing concept proves relatively unsuitable for a high integration density.
The object of the invention is therefore to disclose an electronic circuit for generating a transmission frequency which on the one hand offers the favorable technical requirements of the transmission mixing concept and on the other hand permits a high integration density of the circuit to be achieved, and thus makes cost-effective manufacture possible.
Accordingly, an electronic circuit is proposed for generating a transmit frequency fs for a transceiver, which circuit contains the following components: a controllable oscillator for generating an oscillator frequency fosz, a divider by a factor N and a mixer stage with a subsequent band filter, the components being connected to one another in such a way that the oscillator frequency fosz and an oscillator frequency fosz/N divided by the factor N are fed to the mixer as input signals and output by it as transmit frequency fs.
A significant advantage of this circuit is that a lower phase noise is produced with the circuit according to the invention than would be achievable with the two oscillators of the known transmission mixing concept because only a single oscillator can contribute to the phase noise.
A simplification of the structure of the circuit is achieved by virtue of the fact that, instead of the mixer stage with subsequent band filter, a single-sideband mixer or Image Reject Mixer is used. Single-sideband mixers are available as ready-made components and can be integrated into the circuit structure in a compact fashion.
A further advantageous refinement of the electronic circuit according to the invention can consist in using a PLL circuit for stabilization, to which PLL circuit a reference frequency, and either the oscillator frequency or the output frequency of the band filter or if appropriate of the single-sideband mixer, are fed as input signals.
Furthermore, it may be advantageous if the factor N of the divider supplies a multiple of the number 2 and/or is greater than 1 and supplies two output signals which are phase-shifted with respect to one another by 90°.
The desired phase shift by 90° can be achieved by phase shifting part of the signal by 90° and maintaining the original phase for the remaining part of the signal, or by phase shifting both parts of the signal by +45° and −45°, respectively. In both cases, a phase difference of 90° remains.
A further advantageous refinement of the electronic circuit according to the invention can consist in the fact that a control device is additionally provided which, at the time of the switching on of a transmit output stage connected to the output of the single-sideband mixer, superimposes on an oscillator control signal a data signal for generating a frequency modulation. Such a control device is used, for example, in what is referred to as TDMA systems.
In respect of optimal integration and simple implementation of the circuit it is also advantageous to implement the control device using an ASIC component.
Another advantageous refinement of the circuit provides for the control device to activate two switches alternately, which enables a connection of the oscillator control input either to a data modulator or, for the purpose of channel setting, to the PLL.
Furthermore, an alternative refinement to the electronic circuit according to the invention can consist in the fact that a superimposition receiver is provided which obtains a superimposition frequency directly from the oscillator frequency fosz, and that a changeover device is provided which in the case of transmission feeds the single-sideband mixer output frequency and in the case of reception feeds the oscillator frequency to the PLL.
The oscillator can advantageously operate in a voltage-controlled or current-controlled fashion, for example, and if appropriate a reference frequency can also be fed externally.
Of course, the abovementioned features of the invention which are to be explained can be used not only in the respective specified combination but also in other combinations or alone without departing from the scope of the invention.
Further features and advantages of the invention emerge from the following description of preferred exemplary embodiments with reference to the drawings.
The invention will be explained below in more detail with reference to the drawings, in which, in particular:
Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings, wherein like reference numerals refer to like elements throughout.
In this circuit, at the moment of the switching on of the transmitting amplifier 4, the generation of frequencies is disrupted owing to a load change and/or effects—indicated by the arrows 6 and 7—and an undesired frequency jump is produced. The load change occurs during the switching on of the transmitting amplifier 4 as a result of the change in its input impedance.
Effects on the frequency generating components are produced as a result of the irradiation by the antenna 5, or by other coupling paths (not illustrated here) between the transmit output stage and the frequency generating components. An example of this are the supply voltage feeder lines.
The best known circuit with the most effective suppression of feedback and frequency jumps during the switching on of the transmitting amplifier is illustrated in FIG. 4. This
If the frequencies of the oscillators 2 and 14 are selected such that they have a nonharmonic relationship with the desired frequency, there is a resulting high degree of immunity to load changes, that is to say during the switching on of the transmitting amplifier, and to its effects. As a result, the requirements made of the shielding, blocking and isolating stages are reduced considerably in comparison with the circuits illustrated in
The second arrangement is composed, at the input end, of a single oscillator 2 which is stabilized by a PLL circuit 1. A summing stage 18, by which an FM modulation signal 26 can be supplied, is arranged between the oscillator 2 and the PLL circuit 1. The frequency fosz of the oscillator 2 is fed to a frequency divider 19, and the frequency fosz/N is generated. Both frequencies fosz and fosz/N are then fed to a mixer 32 in order to form the transmit frequency fs. In the subsequent band filter 22, the undesired secondary frequencies which have also been produced are filtered out and the filtered frequency is conducted to the amplifier output stage 4. Either the oscillator frequency fosz can be fed back to the PLL circuit 1 via the line 34, or the transmit frequency fs can be fed back to the PLL circuit 1 from the output of the band filter 33.
The desired transmit frequency fs is thus obtained by:
f s =f osz±(f osz /N)=f osz*(1±1/N)
where fs=transmit frequency, fosz=oscillator frequency, N=divider factor
As is apparent from the mathematical relationship, a nonintegral relationship results between the transmit frequency fs and the oscillator frequency fosz, which promises a good degree of immunity to effects. The selection of the signs in the formula is determined by the connection of the single-sideband mixer. There is the freedom to allow the oscillator to oscillate either below or above the desired frequency. Basically, the oscillator frequency fosz can also be selected in such a way that the oscillator frequency fosz fulfils the criterion of the best phase noise (best quality of the coil) given the equipment.
In addition to the circuit according to the invention for generating the transmit frequency, a TDMA controller 31, known per se, for which the circuit for generating frequencies according to the invention is particularly suitable is also illustrated in FIG. 5.
In this further development, a single-sideband mixer or Image Reject Mixer 20 was used instead of the mixer 32 and the subsequent band filter 33. If the operating conditions require it, another filter element (not illustrated) for suppressing the harmonics of the divided signal can also be used downstream of the divider 19.
The single-sideband mixer 20 typically has a first phase shifter 21 for phase shifting and dividing the incoming oscillator frequency fosz and a second phase shifter 22 for phase shifting the incoming divided oscillator frequency fosz/N by 90° in each case. These frequencies which are each phase-shifted by 90° are mixed in the mixers 23 and 24, superimposed in the summing stage 25 and output as a desired transmit frequency fs.
It is to be noted that the purpose of the phase shifting of 0° and 90° illustrated here can also be achieved by a phase shift by −45 and +45°.
The desired transmit frequency fs is also obtained here and in all the further examples in accordance with the same formula to be described with respect to FIG. 5.
Since the frequency divider and single-sideband mixer can be integrated without difficulty with the contemporary technologies, this circuit leads to a considerable saving in chip area. Furthermore, there is a saving of a PLL with the external components of the loop filter connected thereto.
Another circuit according to the invention for generating a transmit frequency is illustrated in FIG. 7. The oscillator frequency fosz is fed on the one hand to a divider 19 and on the other hand to a phase shifter 36. By using a factor N which can be divided by two, the phase shift of 90° required for the principle of single-sideband mixing can advantageously be generated easily and precisely, as a result of which there is better suppression of the undesired sideband from the mixing process.
The output signals which are shifted by 90° are obtained in a generally known way in that the last divider stage of a divider chain is a double design, one of the two divider stages being fed the input signal in inverted form.
Such an embodiment has the advantage that any desired, even multivalued types of modulation can be generated with good frequency and/or phase stability.
The modulation signal 4 which is supplied can, for example, be the IQ baseband, generated by a digital signal processor, of a GMSK, N-PSK or quadrature amplitude modulation.
Another modification of the circuit according to the invention is illustrated in FIG. 9. This corresponds essentially to
In the mixer 32, the transmit frequency fs including secondary frequencies is in turn generated by mixing with the oscillator frequency fosz, the secondary frequencies are largely filtered out during passage through the subsequent band filter 33 and the remaining transmit frequency fs is conducted to the transmitting amplifier 4 and irradiated via the antenna 5. As in
A further possible way of transferring a modulation onto the transmit signal is illustrated in FIG. 10. The circuit also corresponds here to the simple design from
In the receiving mode, the oscillator 2 generates the superimposition signal, while in the case of transmission the same oscillator 2 is used to generate the transmit frequency. The intermediate frequency in the case of reception is selected in such a way that it lies in the vicinity of the oscillator offset frequency in the case of transmission. The tuning range of the receiver is somewhat smaller in accordance with the offset between the transmit frequency and oscillator frequency, which however has hardly any effect in practice with relatively large divider factors. The coupling with the PLL is carried out by the changeover switch 38, downstream of the single-sideband mixer 20 in the case of transmission and directly by the oscillator 2 in the case of reception, in order to permit a uniform tuning step size of the PLL with the same reference frequency. It is advantageous here that only a single oscillator 2 is necessary for the transmitting mode and the receiving mode and at the same time good stability of the transmit frequency is achieved in the TDMA mode.
This circuit design shown is particularly suitable for DECT systems.
A disadvantage of the circuit according to the invention in comparison with an oscillator which operates at the limit frequency, namely the additional undesired mixing products of a real single-sideband mixer, can be reduced by adding a high-frequency filter, necessary in any case in the receiver, upstream of the transmit/receive changeover switch. In this case, the filter is used both for the transmit branch and for the receive branch.
Such a solution is illustrated by way of example in
This is an arrangement such as is used, for example, in a DECT system with “open-loop modulation method”. When the switch 42 is closed, the oscillator 2 is set to the desired channel by means of the PLL circuit 1 during a time slot which is not required for the transmitting/receiving mode. Just before the start of transmission, the switch 42 opens and the control variable which is acquired up to that point is stored in a storage element, not illustrated separately in the FIGURE. A baseband signal for generating the DECT-GFSK (Gaussian frequency shift keying) modulation is superimposed by means of the switch 43 during the emission of the stored control variable. The necessary frequency stability is made possible during the emission by the arrangement according to the invention of the divider and mixer or single-sideband mixer. That is to say, high-frequency effects from the transmitter stage on the oscillator 2 do not bring about any frequency offset after the switching on of the transmitter.
In total, the circuit according to the invention therefore ensures that, on the one hand, the favorable technical requirements of the transmission mixing concept can be utilized and, on the other hand, a high integration density of the circuit, and thus cost-effective manufacture are made possible.
The invention has been described in detail with particular reference to preferred embodiments thereof and examples, but it will be understood that variations and modifications can be effected within the spirit and scope of the invention.
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|U.S. Classification||331/37, 455/323, 331/40, 327/255|
|International Classification||H03L7/185, H04B1/04, H03B21/02, H03L7/16|
|Cooperative Classification||H03B21/02, H03L7/185, H03L7/16|
|European Classification||H03L7/16, H03L7/185, H03B21/02|
|Jan 30, 2002||AS||Assignment|
Owner name: SIEMENS AKTIENGESELLSCHAFT, GERMANY
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:DETERING, VOLKER;HEINEN, STEFAN;REEL/FRAME:012523/0015;SIGNING DATES FROM 20011203 TO 20011213
|Jul 15, 2008||FPAY||Fee payment|
Year of fee payment: 4
|Jul 6, 2012||FPAY||Fee payment|
Year of fee payment: 8