US 7009428 B2
A system of the present invention uses small swing differential source synchronous voltage and timing reference (SSVTR and /SSVTR) signals to compare single-ended signals of the same slew rate generated at the same time from the same integrated circuit for high frequency signaling. The SSVTR and /SSVTR signals toggle every time the valid signals are driven by the transmitting integrated circuit. Each signal receiver includes two comparators, one for comparing the signal against SSVTR and the other for comparing the signal against /SSVTR. A present signal binary value determines which comparator is coupled to the receiver output, optionally by using exclusive-OR logic with SSVTR and /SSVTR. The coupled comparator in the receiver detects whether change in signal binary value occurred or not until SSVTR and /SSVTR have changed their binary value. The same comparator is coupled if the signal transitions. The comparator is de-coupled if no transition occurs.
1. A system for detecting a transition in an incoming signal, comprising:
first and second input terminals for receiving, respectively, an oscillating voltage reference and an incoming signal;
an output terminal providing an output signal at a current sampling time, the output signal logically equal to the previous logical state of the incoming signal, the previous logical state determined at a previous sampling time;
a first comparator coupled to the first and second input terminals for comparing the reference and the incoming signal at the current sampling time to generate a first result; and
a first controller coupled to the first comparator for coupling the first result to the output terminal based on the previous logical state of the incoming signal.
2. The system of
3. The system of
4. The system of
5. The method of
6. The system of
7. The system of
8. The system of
9. The system of
a third input terminal for receiving an oscillating voltage reference complement;
a second comparator coupled to the second and third input terminals for comparing the complement and the incoming signal at the current sampling time to generate a second result; and
a second controller coupled to the second comparator for coupling the second comparator to the output terminal based on the previous logical state.
10. The system of
11. The system of
12. The method of
13. The method of
14. The method of
15. The method of
16. A method, comprising the steps of:
obtaining an oscillating voltage reference and an oscillating voltage reference complement, the oscillating reference complement being a complement of the oscillating reference;
receiving the incoming signal;
comparing by a first comparator the oscillating reference against the incoming signal at a current sampling time to generate a first result;
comparing by a second comparator the oscillating reference complement against the incoming signal at the current sampling time to generate a second result;
using a control signal based on the previous logical state of the incoming signal to control whether the first result or the second result passes as an output signal, the output signal defining the current logical state of the incoming signal and acting as the previous logical state of the incoming signal at the next sampling time.
17. The method of
18. The method of
19. A receiver comprising:
a first comparator for comparing an oscillating voltage reference and a new signal at a current sampling time;
a second comparator for comparing a complement of the oscillating voltage reference and the new signal at the current sampling time;
an output terminal coupled to one of the first and second comparators;
circuitry for maintaining the comparator that is coupled to the output terminal when the new signal transitions relative to the previous logical state of the incoming signal as determined at the previous sampling time; and
circuitry for coupling to the output terminal one of the first and second comparators that is not coupled to the output terminal and de-coupling from the output terminal one of the first and second comparators that is coupled to the output terminal when the new signal does not transition relative to the previous logical state of the incoming signal as determined at the previous sampling time.
This application is a divisional of U.S. patent application Ser. No. 09/851,622 filed May 8, 2001, now U.S. Pat. No. 6,812,767, which is a continuation of U.S. patent application Ser. No. 09/475,087 filed Dec. 30, 1999, now U.S. Pat. No. 6,255,859, which is a divisional of Ser. No. 09/057,158 filed Apr. 7, 1998, now U.S. Pat. No. 6,160,423, issued on Dec. 12, 2000, which claims priority to provisional application Ser. No. 60/078,213, each of which is hereby incorporated by reference.
1. Field of the Invention
This invention relates generally to computer signal communication, and more particularly to an integrated circuit interface and method for high speed block transfer signaling of data, control and address signals between multiple integrated circuits on a bus or point-to-point with reduced power consumption.
2. Description of the Background Art
Semiconductor integrated circuits used in digital computing and other digital applications often use a plurality of Very Large Scale Integration (VLSI) interconnected circuits for implementing binary communication across single or multi-segmented transmission lines. Conventional transmission lines include traces, which are formed on a suitable substrate, such as a printed circuit board. Each transmission line may be designed, for example, using so-called micro-strip traces and strip line traces to form a transmission line having a characteristic impedance on the order of about 50–70 ohms. Alternatively, each transmission line may have its opposite ends terminated in their characteristic impedance. The output load on a driver for such a transmission line may be as low as 25–35 ohms.
To consume reasonable power, high frequency signaling requires small amplitude signals. For a receiver to detect voltage swings (e.g., 0.8v to 1.2v) easily in a noisy environment like GTL, HSTL, SSTL or RAMBUS, the current must also be very large (e.g., on the order of 50 to 60 milliamps per driver). A typical receiver uses a comparator with a voltage reference (VREF) signal configured midway between input high voltage (VIH) and input low voltage (VIL). The VREF signal is a high impedance DC voltage reference which tracks loosely with power supplies over time, but cannot respond to instantaneous noise. Conventionally, High Output Voltage (VOH) and Low Output Voltage (VOL) denote signals emerging from the transmitting source, and VIL and VIH denote signals arriving at the input of the receiving device, although they can be considered the same signal.
Accordingly, there is a need for low power drivers and reliable receivers for high frequency operation of a large number of single-ended signals in existing technology for low cost VLSI digital systems.
A system of the present invention uses small swing differential source synchronous voltage and timing reference signals (SSVTR and /SSVTR) to compare single-ended signals of the same swing generated from the same integrated circuit for high frequency signaling. It will be appreciated that “/” is being used to indicate a logical NOT. All signals are terminated with their characteristic impedance on both ends of the transmission lines. SSVTR and /SSVTR toggle every time the valid signals are driven by the transmitting integrated circuit. Each signal receiver includes two comparators, one for comparing the signal against SSVTR and the other for comparing the signal against /SSVTR. A present signal binary value determines which comparator is coupled, optionally by using exclusive-OR logic with SSVTR and /SSVTR. Until SSVTR and /SSVTR have changed their binary value, the coupled comparator in the receiver detects whether a change in signal binary value occurred. Again, it will be appreciated that SSVTR and /SSVTR change their binary value every time the signal can change its binary value. SSVTR and /SSVTR are preferably synchronized with the signal.
The method of the present invention includes the steps of obtaining an oscillating source synchronous voltage and timing reference and its complement (SSVTR and /SSVTR), and receiving an incoming single-ended signal. The method compares the oscillating reference against the incoming signal by a first comparator to generate a first result, and compares the complement against the incoming signal by a second comparator to generate a second result. The method then selects one of the first result or the second result as an output signal based on the previous signal. The step of selecting one of the results includes comparing the output signal to the reference (SSVTR) and to the complement (/SSVTR). The step of selecting further includes manipulating the output signal from the previous signal towards the first result or second result, based on the comparator which is currently coupled. If the incoming signal changes, the step of selecting includes maintaining the same comparator coupled. If the incoming signal stays the same, the step of selecting includes de-coupling the currently coupled comparator and coupling the other comparator. The method then allows the circuit to stabilize.
The system and method advantageously eliminate the need for a high impedance VREF signal for comparison of small swing single-ended signals. This reduces the need for three distinct voltage levels (the output high level, output low level and the VREF level) to two distinct voltage levels (the output high level and the output low level). Eliminating VREF reduces necessary voltage swing and accordingly reduces power consumption. Using a receiver with dual comparators allows coupling of the receiver to the same comparator when the signal changes every cycle. Only one comparator is coupled based on the current binary value of the signal and SSVTR. The system has an individually adjustable delay for each receiver to couple or de-couple the comparator, thereby reducing the effect of skew during transmission of source synchronous signals. The system may have multiple differential source synchronous voltage and timing reference signals to compare multiple single-ended signals in the same integrated circuit such as a microprocessor or system controller that has many signals. The system and method provide differential signaling benefits in a single-ended signaling system.
Using the same concept, the system may have bi-directional complementary source synchronous voltage and timing reference signals to compare bi-directional single-ended signals. The system may have a driver or transmitter for controlling the signal slew rate to be a substantial portion the total signal period, thereby reducing output current. The system may have internal impedance matching circuitry such as pull-up resistors or grounded gate p-channel for matching the characteristic impedance of the transmission line on both ends of a point-to-point connection between CPU and cache or CPU and system controller. The system has a dual comparator circuit to convert a single-ended bus with two complimentary signals to be transmitted and received with comparable noise immunity of differential bus for internal data bus of memory, processor or other wide data bus type integrated circuits. The system preferably has variable device size of the transmitter with slow turning-on and slow turning-off to have similar slew rates for all signals in each group of SSVTR and /SSVTR and plurality of signals which are transmitted together.
The present invention provides a signaling system and method for high-speed communication on multiplexed bus or point-to-point connections between multiple VLSI devices with lower power consumption relative to current methodology of interfacing single-ended signals. The signaling system can be used to connect multiple memory devices with a multiplexed bus to a memory controller for block transfer of data, addresses and control information. By using multiple buses, devices such as DRAMs, cross-point switches, processors, wide SRAMs and System controllers can be put together to achieve bandwidths above four gigabytes/sec. Virtually all of the signals needed for computer or other digital systems can be sent over this bus. Persons skilled in the art will recognize that all devices like CPUs in the computer system need include the methodologies and bus structures of this system.
As shown in
For chip-to-chip communication on a bus or point to point, all signals are transmitted preferably at substantially the same time from the same chip to another chip or plurality of chips connected on the bus and preferably have substantially the same loading, swing and slew rate (when the signals are transitioning). Also, for intra-chip communication, the signals are driven preferably at substantially the same time from the same area or block to other areas or other blocks in the same chip and preferably have substantially the same loading, swing and slew rate (when the signals are transitioning)
To facilitate extremely high data transmission rates over this external bus, the bus cycles are initiated when SSVTR is low (i.e., /SSVTR is high). All block transfer begins during the cycle when SSVTR is low and ends with SSVTR going low to ease presetting the receiver 405 for the last binary value of the signal. This allows burst transfers of even number of bits. When the signals need to change direction (due to the multiplex nature of signals), one or more dead cycles may be required for settling down the bus due to propagation delays or settling of SSVTR and /SSVTR, when they are bi-directional.
As stated above, SSVTR is initially set to VOL and /SSVTR and SNx are initially set to VOH. SN is initially set to a full rail high output voltage. Accordingly, the comparator 410 a amplifies high voltage SNx minus low voltage SSVTR, thereby providing a high output signal. The comparator 410 b amplifies high voltage SNx minus high voltage /SSVTR, providing a noise-amplified unknown output signal. Switch 415 selection is controlled by exclusive-OR (XOR) logic gates 425. More particularly, XOR gate 425 a compares a full rail SSVTR amplified signal (VT) against output signal SN, and generates a control signal for controlling switch 415 a. XOR gate 425 b compares full rail /SSVTR (/VT) against output signal SN, and generates a control signal for controlling switch 415 b. In this initial state, only SSVTR and accordingly VT are low, thereby causing XOR 425 a to drive switch 415 a closed. Accordingly, the comparator 410 a output (high) reaches output terminal 420. XOR 425 drives switch 415 b open, thereby preventing the entry of the unwanted output signal from comparator 410 b. Receiver 405 is stable.
Following the example illustrated in
Still following the example of
Each receiver 405 can easily detect and amplify very small signals on the order of 100–250 mV. If the transition has occurred in the single-ended signal SNx, the output signal SN has the new level opposite to its previous signal level. Since both SSVTR (or /SSVTR) and single-ended signals have transitioned, the same comparator 410 is still coupled to the signal output terminal. If the single-ended signals SNx have not transitioned, then the signal output SN does not change, the comparator 410 coupled at the start of the transition is de-coupled from the output after the SSVTR and /SSVTR receiver has amplified their new binary state (VT & /VT), and the other comparator 410 which has opposite /SSVTR (or SSVTR) is coupled to provide the signal output. The old output level is thereby restored.
It will be appreciated that a receiver 405 may be implemented without using XORs. This may be implemented by using the known polarity of SSVTR and /SSVTR in the initial cycle and all single-ended signals starting high. The SSVTR and /SSVTR transition in every cycle. Thus, their polarity in every cycle may be determined by examining the system clock in a synchronous system and defining cycle start in even clock cycles (i.e., SSVTR is low in the even clock cycle and /SSVTR is high). Then, only the output signal “SN” is monitored to couple and de-couple the comparators 410 based upon whether output signal SN changes state every cycle or not. If output signal SN changes state, the coupled comparator is left alone. If the output signal SN does not change, the coupled comparator is de-coupled and the other comparator is coupled and so on.
It will be further appreciated that a system embodying the invention enables all signals to be connected to low impedance sources, enables all signals to present voltage and noise conditions virtually similar to differential signaling in noise immunity, and enables reduction of voltage swing compared to other single-ended signaling technologies like RAMBUS, HSTL or GTL. The small swing of 0.5v implemented in this exemplary embodiment allows for very high signal rates with much lower power consumption as compared to other existing single-ended signaling technologies. Further, it will be appreciated that each receiver 405 amplifies the single-ended signals SNx during the transition of the signals without the need of a conventional clock or other timing signal except SSVTR, /SSVTR and their amplified versions VT and /VT.
The master 205 in step 520 simultaneously drives SSVTR and /SSVTR to their opposite states and all single-ended signals SNx to their desired levels. The receiver 405 in step 530 compares the single-ended signal SNx against SSVTR and /SSVTR in respective comparators 410. The receiver 405 in step 540 determines whether the single-ended signal transitioned. If so, then the receiver 405 in step 545 passes the result to the output terminal 420, and keeps the same comparator 410 coupled to the terminal 420. If not, then the receiver 405 in step 550 decouples the previous comparator 410, couples the other comparator 410 to the output terminal 420, and keeps the same output signal (SN). The transmitter 405 in step 555 determines whether the signal burst continues. If so, then method 500 returns to step 520. Otherwise, method 500 ends.
Maintaining the current at 20 ma and having parallel terminations of 50 ohms on both ends of the transmission line 610 (as controlled by R1 and R2) generates a 500 mv swing under all conditions. To have slow rise and fall times on the output and to minimize reflections, signal coupling and termination network switching noises, the skew control circuitry 665 controls the pull down transistor T1 to turn on and turn off slowly. The preferred slew rate is 1.6 ns/volt with transition times of 0.8 ns for 500 mv.
For a uniformly transitioning ramp-like signal, the preferred slew rate of signals is four times the sum of two inverter delays and an exclusive-OR delay in a given technology. In 0.25μ CMOS technology with an operating voltage of 2.5V, the inverter delay is 50 picoseconds and the exclusive-OR delay is approximately 120 picoseconds. Thus, the preferred slew rate is approximately 880 picoseconds. For signals transmitted above the rate of 600 MHz, the signal slew rate is preferably less than 110% of the signal rate. The preferred slew rate for exponential signals is slightly faster if the signal reaches 75% of its final value earlier than ¾ of the transition time. The differential signals preferably cross half way through the voltage transition. At around ¾ of the way through the voltage transition, the signals have a difference of about 250 mv which can be converted quickly to a large swing signal. To avoid noise amplification and to prevent signal coupling to the receiver output upon receipt non-transitioning single-ended signals, the transition time between 75% and the final signal value is preferably higher than the sum of two inverter delays and the exclusive-OR delay. It will be appreciated that the slew rate can go as fast as it takes amplified noise to reach the output of the comparator 410 whose output is coupled to the output terminal 420. That is, upon receiving a non-transitioning signal, the switches 415 switch state before the comparator output changes state based on noise amplification. The output of the currently coupled comparator 410 approaches an undetermined (noise amplified) state. The switches 415 must switch states before the undetermined output becomes available. It will be further appreciated that device mismatches, manufacturing tolerances and signal reflection will affect the speed at which the output of the comparator 410 reaches the undetermined state. As the technology improves, gate delays, faster slew rates and faster signal rates will be achievable.
It will be appreciated that the signal transition times may be slightly higher than the signal rate. In some heavily loaded buses, the swing can be increased to take care of transmission losses, still presenting 500 mv for the receiver 210 to sense easily. It will be further appreciated that various slew rates, exponential transition times and voltage swings are possible based on technology, loading, and receiver acquisition and resolution delays. Even transition times slightly higher than signal rate are possible with transitioning signals reaching 90 to 95% percent of their final value, while bursting. Also during testing the skew between single-ended signals and SSVTR and /SSVTR is adjusted using NMOS pull down size and resistors in the gate prior to it, using well known techniques like laser fuse blowing or setting the register code to achieve the signal waveform shape as shown in
The outputs of the differential amplifiers are amplified and inverted by an inverter 704, pass through CMOS transmission gates 706 and are tied together at node 708. The transmission gates 706 are selectively operated depending on the amplified state of previous signal (SN) exclusively-ORed with an amplified state of SSVTR or /SSVTR, i.e. VT or /VT respectively. The exclusive-OR is designed to be stable without glitches for small timing variations between SN, VT and /VT reaching their respective logic levels.
Various embodiments are shown.
By using a 1.2v termination and receiver 405 as shown in
It will be appreciated that the DC bias point of each differential amplifier in the receiver 405 is configured so that the receiver 405 output voltage is above half-VCC when both the small swing voltages (single-ended signal SNx and SSVTR or /SSVTR of the enabled differential amplifier) are close to VIH and below half-VCC when both the small swing voltages are close to VIL. This DC biasing allows for adequate margin and preservation of output signal SN when the single-ended signal SNx does not change state and the SSVTR or /SSVTR of the enabled differential amplifier is closing the differential signal before it is de-coupled.
Since the receiver 405 operates during the signal transition for a small swing single-ended signal, the concept of set-up and hold-time from a specified time after the signal level reaches VIH /VIL or VREF in previous signaling techniques no longer applies. Also, there is no VREF (reference voltage) for comparison with the signal voltage. By eliminating the timing necessary for set-up and hold and the timing needed to enable voltage margins for sensing around VREF, the operating frequency is considerably increased with lower power consumption. Further, all receivers 405 are self timed, without the need of a global clock, allowing the receivers 405 to be adjusted individually for elimination of board or package level transmission skew.
By using devices with very low power dissipation and close physical packing, the bus can be made as short as possible, which in turn allows for short propagation times and high data rates. As shown in
It will be appreciated that multiple buses are required for devices like SLDRAM, DDR SDRAM or DDR SRAMs, where signals are transmitted and received simultaneously.
For an SLDRAM, the 40-bit command and address is sent in a packet of four 10 bit words. SSVTR0 and /SSVTR0, which may be referred to as the system differential clock, operates at 500 Mhz. A Phase-Locked Loop (not shown) is used to lock the clock frequency and timing for various internal purposes and driving the data output with SSVTR1 and /SSVTR1 on both edges for a data rate of 1 Ghz. All the high frequency signals are terminated on both ends of the bus with their characteristic impedance. The termination on the memory controller end can include external resistances, internal resistances or internal grounded gate P-channel devices, since this memory controller is usually the master and is fixed. Since the number of components (SLDRAMs) 1310 (which operate like slaves) is variable, components 1310 are preferably terminated by external resistors at the end of the transmission lines. The -18 bit bi-directional data bus 1330 and 1335 operates at the same frequency as the system clock for synchronization and sends data in eight 18-bit words in four clock cycles (8 ns) or 2.25 gigabytes/sec from a single SLDRAM. Care is taken to balance the load on SSVTR0 and /SSVTR0 by adding dummy gates and lines to look comparable to SSVTR1 and /SSVTR1. This load balancing makes the slew rate due to loading be similar and allows similar margins for all signals.
When higher bandwidth is required, a system 1350 can use four buses as shown in
The foregoing description of the preferred embodiments of the present invention is by way of example only, and other variations and modifications of the above-described embodiments and methods are possible in light of the foregoing teaching. For example, although the system and method have been described as transmitting SSVTR and /SSVTR from a master 205 to a receiver 405, one skilled in the art will recognize that one reference may be sent and the complement generated on the receiver 405 side. Using the technique with other technologies, such as bipolar or gallium arsenide, which have similar switching devices and gates, can alternatively be used. Components of this invention may be implemented using a programmed general purpose digital computer, using application specific integrated circuits, or using a network of interconnected conventional components and circuits. The embodiments described herein are not intended to be exhaustive or limiting. The present invention is limited only by the following claims.