|Publication number||US7116173 B2|
|Application number||US 10/498,489|
|Publication date||Oct 3, 2006|
|Filing date||Feb 28, 2002|
|Priority date||Feb 28, 2002|
|Also published as||CN1623281A, US7336132, US20050218989, US20060279359, WO2003073627A1|
|Publication number||10498489, 498489, PCT/2002/1840, PCT/JP/2/001840, PCT/JP/2/01840, PCT/JP/2002/001840, PCT/JP/2002/01840, PCT/JP2/001840, PCT/JP2/01840, PCT/JP2001840, PCT/JP2002/001840, PCT/JP2002/01840, PCT/JP2002001840, PCT/JP200201840, PCT/JP201840, US 7116173 B2, US 7116173B2, US-B2-7116173, US7116173 B2, US7116173B2|
|Inventors||Takayuki Tsutsui, Masahiro Tsuchiya, Tetsuaki Adachi|
|Original Assignee||Renesas Technology Corp.|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (11), Referenced by (17), Classifications (27), Legal Events (5)|
|External Links: USPTO, USPTO Assignment, Espacenet|
The present invention relates to a high frequency power amplifier circuit for amplifying an input high frequency signal with power amplifying elements which are field effect transistors, and to a technique which can be applied effectively to the wireless communication equipment such as a portable telephone unit which incorporates this high frequency power amplifier circuit. The invention particularly relates to a technique for operating a high frequency power amplifier circuit of the wireless communication equipment at optimal bias conditions in both the operation mode of using the saturation region and the linear operation mode of using the non-saturation region of power amplifying FETs (field effect transistors), and for detecting the output level for feedback control accurately based on the current detection scheme.
Among the conventional schemes of wireless communication equipment (mobile communication equipment) such as portable telephone units, one is GSM (Global System for Mobile Communication) which is adopted in European countries. The GSM scheme performs the phase modulation called GMSK (Gaussian Minimum Shift Keying) which shifts the phase of carrier wave in accordance with transmission data.
Generally, the transmission output stage of the wireless communication equipment incorporates a high frequency power amplifier circuit. Some GSM-based wireless communication equipment is designed to establish the communication output power level in need by controlling the bias voltage of the high frequency power amplifier circuit in accordance with the control voltage which is produced by the APC (Automatic Power Control) circuit based on the demanded transmission level from the baseband LSI and the signal from the transmission output detector.
In the field of recent portable telephone units, there is a proposal of the EDGE (Enhanced Data Rate for GMS Evolution) scheme having a dual mode communication function, in which audio signal communication is performed based on GMSK modulation and data communication is performed based on 8-PSK (Phase Shift Keying) modulation.
The 8-PSK modulation is the phase shift of carrier wave derived from the GMSK modulation, with amplitude shift being added. It is capable of sending 3-bit information per symbol, in contrast to the GMSK modulation which sends 1-bit information per symbol. Accordingly, the EDGE scheme can perform communication at the higher transmission rate than the GSM scheme.
The high frequency power amplifier circuit of the GSM-based communication system can operate in the saturation region for amplifying the phase-modulated signal in accordance with the demanded output level, whereas the high frequency power amplifier circuit of the wireless communication system, which performs the EDGE-based transmission/reception, necessitates the amplitude control and therefore must have a linear operation in the non-saturation region.
For the high frequency power amplifier circuit of the communication system which is operative based on both the GSM scheme and EDGE scheme, a conceivable operational manner of the high frequency power amplifier circuit is to control the gate bias voltage of the output FET in accordance with the demanded output level, with the input signal amplitude being fixed, in the GSM mode which has GMSK modulation, and to control the output power by varying the input signal amplitude, with the gate bias voltage of the output FET being fixed, in the EDGE mode which has 8-PSK modulation.
However, this manner necessitates a variable-gain amplifier and its control circuit for varying the input signal amplitude in the EDGE mode, resulting in an increased circuit scale (refer to
In the case of the fixed gate bias voltage scheme, with satisfactory linear characteristics being intended in the EDGE mode in which the high frequency power amplifier circuit must have a linear operation, a large idle current is needed to flow by setting such an output FET bias voltage as to have a higher gain relatively to the GSM mode. However, the gain is too high for a small input signal level, and a resulting amplified noise component gives rise to an increased noise leakage to the reception frequency band which is separated by 20 MHz or more from the transmission frequency.
In the wireless communication system, the high frequency power amplifier circuit has its output power controlled in accordance with the demanded output level from the control circuit (baseband circuit, etc.) and based on the feedback of the output of high frequency power amplifier circuit or antenna detected with a coupler and detection circuit. In this respect, the inventors of the present invention studied a current sensing scheme for detecting the output level from the output current of the high frequency power amplifier circuit within the semiconductor chip with the intention of making the circuit scale smaller. The study revealed a problem of this scheme when applied to the control of output power by varying the input signal amplitude, with the gate bias voltage being fixed. Namely, the high frequency power amplifier circuit produces a too small output current variation relative to the DC bias component, particularly at a small output level, and a resulting poor sensitivity of output level detection disables accurate output control and invites the fluctuation of detection level in response to the temperature variation and power voltage variation.
It is an object of the present invention to provide, for a wireless communication system having both of phase modulation and amplitude modulation, a high frequency power amplifier circuit and an electronic component part (module) incorporating the circuit which are capable of reducing the circuit scale by eliminating the need of an amplifier circuit dedicated to vary the input signal amplitude to meet the demanded output level.
Another object of the present invention is to provide, for a wireless communication system having both of phase modulation and amplitude modulation, a high frequency power amplifier circuit and an electronic component part incorporating the circuit which are capable of alleviating the signal leakage to the reception frequency band by lowering the gain of amplifier circuit during a linear operation.
Still another object of the present invention is to provide, for a wireless communication system having both of phase modulation and amplitude modulation, a high frequency power amplifier circuit and an electronic component part incorporating the circuit which are capable of detecting the output level to be fed back based on the current sensing scheme and capable of having a sufficient detection sensitivity even at a small output level thereby to perform accurate output control.
These and other objects and novel features of the present invention will become apparent from the following description and attached drawings.
Among the affairs of the present invention disclosed in this specification, representatives are briefed as follows.
A first part of the present invention, which is intended for a wireless communication system having a first operation mode of amplifying a phase-modulated high frequency signal with a high frequency power amplifier circuit and a second operation mode of amplifying a phase and amplitude-modulated high frequency signal with the high frequency power amplifier circuit, is designed to put in a high frequency signal of a fixed amplitude and frequency to the high frequency power amplifier circuit and control the bias states of the amplifying stages of amplifier circuit in accordance with a control signal which is produced by a control circuit based on a demanded output level and detected output level, thereby performing the signal amplification to meet the demanded output level. In consequence, the output level can be controlled by the same control system for both the first and second operation modes without the need of a variable-gain amplifier for the preceding stage of the high frequency power amplifier circuit, whereby the control system can be simplified.
A second part of the present invention, which is intended for a wireless communication system which requires a linear operation of a high frequency power amplifier circuit, is designed to configure the first-stage amplifier of amplifier circuit with a dual-gate FET or two FETs in serial connection, with an input high frequency signal and a first bias voltage being put in to the first gate terminal of the FET and with a second bias voltage which is higher than the first bias voltage being put in to the second gate terminal, and to create by the first bias voltage such a bias state as to render the linear characteristics to the first-stage amplifier and suppress the gain of the first-stage amplifier by the second bias voltage. In consequence, the high frequency power amplifier circuit has its gain lowered during the linear operation, whereby the signal leakage to the reception frequency band can be alleviated.
A third part of the present invention, which is intended for a wireless communication system having its output level controlled based on the demanded output level and the output level detected by an output detection means, is designed to include an output level detection means made up of a transistor which receives the input voltage of the last-stage amplifier of the high frequency power amplifier circuit, a current mirror circuit which replicates the transistor current, and a resistor which converts the replicated current into a voltage. In consequence, the amplified output level to be fed back can be detected based on the current detection scheme and a sufficient detection sensitivity is ensured even at a small output level, whereby accurate output control can take place.
Embodiments of this invention will be explained in detail with reference to the drawings.
The output level command signal PLS is produced by the high frequency IC 100 under control of the baseband LSI 300. The command signal level is low when the communication distance to the base station is near or it is high when the distance is far. Specifically, the characteristics of the high frequency IC 100 and power module 200 are examined in advance to produce data indicative of the correspondence between the demanded output level and the output level command signal PLS and recorded in the internal nonvolatile memory or the like of the baseband LSI 300 so that the high frequency IC 100 releases an output level command signal PLS by looking up the table of recorded data in response to a demanded output level resulting from the communication with the base station. In case the high frequency IC 100 includes a circuit for correcting the disparity of characteristics, correction data may also be recorded in the internal nonvolatile memory of the baseband LSI 300.
The high frequency IC 100 releases a transmission start signal TXON under control of the baseband LSI 300 besides the output level command signal PLS. Alternatively, the transmission start signal TXON may be issued directly from the baseband LSI 300 to the power module 200.
The arrangement of
The high frequency IC 100 of this embodiment has functions of sending a signal which is GMSK-modulated based on the GSM scheme and sending a signal which is 8-PSK-modulated based on the EDGE scheme. Which of GSM-based transmission (GSM mode) or EDGE-based transmission (EDGE mode) is instructed by the base band LSI 300. For GSM mode transmission, the high frequency IC 100 implements the GMSK modulation for rendering phase modulation to the carrier wave in accordance with information to be sent, and a resulting phase-modulated transmission signal φ TX is fed as a high frequency signal Pin to the power module 200. For EDGE mode transmission, the high frequency IC 100 implements the 8-PSK modulation for rendering phase shift and amplitude shift to the carrier wave in accordance with information to be sent, and a resulting phase/amplitude-modulated signal is fed to the power module 200.
The following Table 1 shows the setting manners of the input signal Pin and output control voltage Vapc to the power amplifier 210 for the system of this embodiment shown in
In the GSM mode, both of the system of this embodiment and the studied system have their input signal Pin fixed in frequency and amplitude and have their output power controlled (varied) in accordance with the output control voltage Vapc.
In the EDGE mode, the studied system has its input signal Pin varied in amplitude by the variable-gain amplifier (AGC) 230 and has its output control voltage Vapc kept constant. For the operational control, there is provided a switch for turning the destination of the control voltage Vapc produced by the bias control circuit 500 from the power amplifier (PA) 210 to the variable-gain amplifier 230 so as to control the gain of the variable-gain amplifier 230, thereby controlling the amplitude of output signal (input signal Pin to the power amplifier 210).
The system of this embodiment, also in the EDGE mode, has its input signal Pin fixed in frequency and amplitude and has its output power controlled (varied) in accordance with the output control voltage Vapc; Accordingly, it can control the high frequency power amplifier circuit with the same control circuit in both the GSM mode and EDGE mode, and does not necessitate a variable-gain amplifier for amplitude control. Consequently, the system arrangement can be simplified and the number of component parts or the chip size in the case of semiconductor integration can be reduced. The power amplifier 210 is designed to allow the use of the same control circuit in both modes as will be explained in the following.
The high frequency power amplifier circuit 210 of this embodiment includes three power amplifying FETs 211,212 and 213, with the FETs 212 and 213 having their gate terminals connected to the drain terminals of the FETs 211 and 212, respectively, thereby constituting a 3-stage amplifier circuit, although this affair is not compulsory. The FETs 211,212 and 213 have their gate terminals connected to the gate terminals of MOSFETs 214,215 and 216, respectively, thereby forming current mirror circuits. The bias control circuit 230 feeds control currents Ic1, Ic2 and Ic3 to the current mirror MOSFETs 214, 215 and 216 so that idle currents I IIDLE1, I IIDLE2 and I IIDLE3, which are equal or proportional to the control currents Ic1, Ic2 and Ic3, flow through the FETs 211, 212 and 213. The bias control circuit 230 and the MOSFETs 214,215 and 216 of current mirror circuits in combination can be conceived to be a biasing circuit.
The FETs 211,212 and 213 have their drain terminals supplied with a power voltage Vdd through respective inductance elements L1, L2 and L3. The high frequency input signal Pin is put in through a capacitor C1 to the gate terminal of the first-stage FET 211. The last-stage FET 213 has its drain terminal connected through a capacitor C10 to the output terminal OUT. Accordingly, the amplifier circuit 210 amplifies only the a.c. component of the high frequency input signal Pin and puts out an amplified signal Pout. The output signal Pout has its power level controlled by the control currents Ic1, Ic2 and Ic3 provided by the bias control circuit 230.
In this embodiment, the first-stage FET 211 and the current mirror MOSFET 214 each consist of a so-called dual-gate MOSFET having two gate electrodes for a channel. The current mirror MOSFET 214 has a serial connection of a resistor Re or Rg, with the voltages on both ends of Re or Rg being applied to the first gate and the second gate of the first-stage power FET 211, causing the power FET 211 to have a flow of idle current I IIDLE1 which is equal or proportional to the control current Ic1. The high frequency input signal Pin is put in to the first gate terminal of the FET 211 having the idle current I IIDLE1.
Although a dual-gate MOSFET is used in this embodiment for the expedience of fabrication, it can be replaced with two MOSFETs 211 a and 211 b connected in series as shown in
The bias control circuit 230 starts operating in response to a start control signal TXON provided by the high frequency IC 100 (or baseband LSI 300). For biasing the 3-stage FETs 211,212 and 213, the circuit 230 produces control currents Ic1, Ic2 and Ic3 for the EDGE mode or produces control currents Ic1′, Ic2′ and Ic3′ for the GSM mode (Ic1′>Ic1, Ic2′>Ic2, Ic3′>Ic3) depending on the mode command signal MODE issued by the high frequency IC 100 (or baseband LSI 300).
The first-stage power FET 211 is connected to the bias control circuit 230 through the resistor Re or Rg and a switch 240 so that the control current Ic1 of the EDGE mode is fed through the resistor Re or the control current Ic1′ of the GSM mode is fed through the resistor Rg. The Rg has its resistance set larger than that of Re so that the first-stage FET 211 has better linearity for the input signal on the first gate in the EDGE mode. The first-stage FET 211 has its second gate supplied with a bias voltage Vcg which is always higher than the bias voltage Vg1 of the first gate by a voltage drop across the resistor Re or Rg.
Based on the input of high frequency signal Pin to the first gate of the dual-gate MOSFET which serves for the first-stage FET 211 and the application of bias voltages Vg1 and Vcg which comply with the output control voltage Vapc to the first gate and second gate, respectively, the gain of FET can be more suppressed according to this embodiment on condition that the input high frequency signal Pin is constant, as compared with the case of putting in the high frequency signal Pin and applying the bias voltage which comply with the output control voltage Vapc to the gate terminal of the usual single-gate power FET.
A conceivable reason is as follows. In case a high frequency signal Pin of certain magnitude from a transmission oscillator or the like is put in to the gate of power FET 211 and the demanded output level is low, it is necessary to attenuate the input signal. If attenuation is attempted by solely controlling the gate bias voltage Vg1 while fixing the FET drain voltage, the FET has its bias state varied significantly, causing the signal to be distorted. In contrast, by lowering the voltage of the first gate and, at the same time, lowering the voltage of the second gate of the dual-gate MOSFET which serves for the power FET 211, it is possible to lower the gain of FET. This situation is more easily understood in the case of FETs 211 a and 211 b connected tandem as shown in
This embodiment is designed to increase the idle current I IIDLE 1 based on the bias voltage Vg1 of the first gate of dual-gate MOSFET, while suppressing the gain with the second gate, whereby it is possible to prevent the emergence of increased noise in the reception band caused by excessive gain of the power amplifier in the linear operation of the first-stage FET 211 in the EDGE mode.
Specifically, in the case of a single-gate FET for the first-stage FET 211 in the circuit of
In the circuit of
In this embodiment, the voltages produced on the ends of resistor Re or Rg by the current Ic1 which is fed to the current mirror circuit of the first-stage amplifier are applied to the first gate and second gate, and therefore Vcg is always higher than Vg1. Alternatively, voltages produced independently to meet this condition may be applied to the first and second gates. In addition to the condition of Vcg>Vg1, it is desirable to control the value of Vcg−Vg1 within the range which meets Vcg−Vg1<1.2Vth, where Vth is the threshold voltage of FET.
The circuit shown in
In the circuit of
Consequently, the distortion of output signal decreases, improving the EVM. By operating the amplifier circuit to have the voltages Vg1 and Vcg of the first and second gates retaining the condition of Vcg>Vg1 so that the drain current of the last-stage FET 213 varies vigorously, the sensitivity of output current detection can be improved as will be explained in the following.
Next, the output detection circuit 220 in the power module 200 of this embodiment shown in
The MOSFET 222 has its gate and drain connected together and the MOSFET 223 has its gate connected to the gate of MOSFET 222, thereby forming a current mirror circuit. For minimizing the current flowing through the output detection circuit 220, an FET which is lower in rating than the last-stage power amplifying FET 213 is used for the output detecting MOSFET 221.
By application to the gate of MOSFET 221 of the same voltage as the gate voltage of the last-stage power amplifying FET 213, a current which is proportional to the drain current of FET 231 flows through the MOSFET 221, and this current is duplicated by the current mirror circuit to flow through the resistor Rs. Accordingly, the voltage VSNS on the connection node N1 of the resistor Rs and MOSFET 223 is proportional to the current of the last-stage power amplifying FET 213. The voltage VSNS which represents the output level detection signal is fed back to the output control circuit 500 of
In case an output detecting MOSFET such as the FET 221 of
Whereas, according to the embodiment of
Owing to the formation of a current mirror circuit, the output detection circuit 220 of
Furthermore, owing to the connection of the input impedance matching capacitor Ci between the gate terminal of MOSFET 211 and the ground, the output detection circuit 220 of the embodiment of
More specifically, the FET 213 has a low gain at a low output level and the drain current Id does not saturate as shown by the dashed line in
In the circuit arrangement of
In this embodiment, the first and second stage power amplifying FETs 211 and 212 and the corresponding current mirror MOSFETs 214 and 215 of the high frequency power amplifier circuit 210 (for each of GSM and EDGE), the bias control circuit 230, and the current mirror MOSFETs 222 and 223 of the output detection circuit 220 are fabricated together as a semiconductor integrated circuit IC1 on a semiconductor chip, although this affair is not compulsory. The MOSFETs 214 and 215 are of the same conductivity type (n-channel type) as the power amplifying FETs 211 and 212, thereby having the same structure and thus having the same thermal characteristics, so that the variation of characteristics of the circuit 210 caused by temperature variation can be minimized. The resistors Re and Rg which conduct a control current from the bias control circuit 230 are connected as externally attached parts to the module.
The last-stage FET 213 of the high frequency power amplifier circuit 210, the corresponding current mirror MOSFET 216, and the output detecting FET 221 are fabricated together as a semiconductor integrated circuit on another semiconductor chip. The gate input resistor Ri of the output detecting FET 221, the current-voltage converting resistor Rs connected in series to the current mirror MOSFET 223, and the input impedance matching capacitor Ci are externally attached parts.
For the dual-band design, the semiconductor integrated circuits including the last-stage FET 213, corresponding current mirror MOSFET 216, and the MOSFETs 221–223 of output detection circuit 220 are fabricated on a chip IC2 for GSM and a chip IC3 for PCS. These semiconductor chips IC1, IC2 and IC3 and the discrete parts including the resistors Re, Rg and Rs and capacitors Ci and C1–C11 are mounted on a common ceramic substrate to become a electronic component part for wireless communication. The above-mentioned micro-strip lines MS1–MS7 are formed of conductor patterns of copper or the like on the ceramic substrate so as to have intended inductance values. The assembly on the ceramic substrate including the power amplifying elements or their integrated circuit, resistors and capacitors are called here “a power module”.
The power module shown in
The first, second and third dielectric sheets 11 have rectangular openings for accommodating the semiconductor chips IC1, IC2 and IC3. Each chip is fixed to the bottom of opening (fourth sheet) by means of a fixing member 14. The fourth and lower dielectric sheets 11 have bear holes 15, which are filled with conductor for conducting the heat produced by the IC1, IC2 and IC3 to the bottom conductor layer so as to enhance the efficiency of heat dissipation.
The IC1, IC2 and IC3 have their top-surface electrodes connected electrically to certain conductor layers 12 through bonding wires 31. The first dielectric sheet 11 has on its top surface a conductor pattern 12 a for forming the micro-strip lines MS1–MS8, and externally attached electronic parts including the capacitors Ci and C1–C11 and resistors Ri and Rs used for the power amplifier circuit 210 and output detection circuit 220 are mounted on this surface. Among these parts, the capacitors can alternatively be formed within the substrate by utilization of the conductor layers on the top and rear surfaces of the dielectric sheet 11.
While the present invention has been described in connection with the specific embodiments, the invention is not confined to these embodiments, but various alterations are obviously possible without departing from the essence of the invention.
For example, in the high frequency power amplifier circuit of the foregoing embodiment, the first-stage power amplifier (power FET 211) is a dual-gate MOSFET and the biasing resistors Re and Rg are used so that the second gate always has a higher voltage than that of the first gate. An alternative scheme is providing an appropriate level shift circuit, instead of the current mirror circuit and resistors Re and Rg, and biasing the first-stage FET 211 by applying the output control voltage Vapc or a derivative thereof to the second gate and applying a shift-down version of the voltage to the first gate.
The power amplifying FETs, which are three stages in the high frequency power amplifier circuits of the foregoing embodiments, may be of two stages or four or more stages. The second and third-stage FETs 212 and 213 each may be two FETs connected in parallel.
In the foregoing embodiments, the first and second-stage power amplifying FETs 211 and 212, the corresponding current mirror MOSFETs 214 and 215, and the bias control circuit 230 which feeds the bias voltages to the gate terminals of these FETs are fabricated as a semiconductor integrated circuit, and the third-stage FET 213 and the output detecting FET 221 are fabricated as another semiconductor integrated circuit. Otherwise, all of these parts, i.e., the power amplifying FETs 211–213, the corresponding current mirror FETs 214–216, and the bias control circuit 230 may be fabricated as one semiconductor integrated circuit on a chip.
Although the present invention has been explained in connection with the embodiments of a high frequency power amplifier circuit used for the communication of EDGE scheme in which power amplifying FETs are operated in the linear region, the present invention is also applicable to the communication equipment of other schemes such as the cdmaOne system which implements multiplexing based on the CDMA (Code Division Multiple Access) scheme and IS95 system in which power amplifying FETs of the high frequency power amplifier circuit are operated in the linear region.
For communication systems of dual-band, triple-band or higher multiplex band scheme capable of performing communication in various communication schemes such as the CDMA of WCDMA (Wideband CDMA) scheme and the GSM scheme having EDGE mode and the DCS scheme, the present invention can be used for the biasing technique for high frequency power amplifier circuits which perform amplification for the transmission based on the CDMA scheme and the transmission of EDGE mode based on the GSM scheme and DCS scheme.
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|U.S. Classification||330/285, 330/296|
|International Classification||H03G3/30, H03G3/10, H04B1/40, H04B1/04, H03F3/193, H03F1/02, H03G3/20, H03F1/06|
|Cooperative Classification||H03F2200/393, H03G3/3047, H04B2001/0416, H04B1/0475, H04B1/406, H03F2200/372, H03F1/0205, H03F1/0272, H03F3/193, H03G3/3042|
|European Classification||H03G3/30D2, H04B1/04L, H03F3/193, H03F1/02T2S, H04B1/40C4, H03G3/30D2B, H03F1/02T|
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