|Publication number||US7242241 B2|
|Application number||US 10/514,243|
|Publication date||Jul 10, 2007|
|Filing date||May 19, 2003|
|Priority date||May 21, 2002|
|Also published as||DE60316314D1, DE60316314T2, EP1537463A1, EP1537463B1, US20060033557, WO2003098368A1|
|Publication number||10514243, 514243, PCT/2003/2156, PCT/GB/2003/002156, PCT/GB/2003/02156, PCT/GB/3/002156, PCT/GB/3/02156, PCT/GB2003/002156, PCT/GB2003/02156, PCT/GB2003002156, PCT/GB200302156, PCT/GB3/002156, PCT/GB3/02156, PCT/GB3002156, PCT/GB302156, US 7242241 B2, US 7242241B2, US-B2-7242241, US7242241 B2, US7242241B2|
|Inventors||Christofer Toumazou, Julius Georgiou|
|Original Assignee||Dna Electronics Limited|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (24), Non-Patent Citations (1), Referenced by (13), Classifications (6), Legal Events (5)|
|External Links: USPTO, USPTO Assignment, Espacenet|
This application is a National Stage entry of International Application Number PCTGB03/02156, filed May 19, 2003. The disclosure of the prior application is hereby incorporated herein in its entirety by reference.
The present invention relates to a reference circuit, and particularly though not exclusively to a reference circuit suitable for providing a current.
Current reference circuits are fundamental building blocks of integrated circuits, and biasing for most integrated circuits can be traced back to an on-chip current reference circuit.
A conventional prior art current reference circuit  is shown in
The prior art circuit shown in
Alternative reference circuits based on replacing the resistor with active devices have been proposed , but these circuits are much more complicated and occupy substantial chip areas.
It is an object of the present invention to provide a reference circuit that overcomes or mitigates one or more of the above disadvantages.
According to the invention there is provided a reference circuit comprising first and second field effect transistors connected to form a first current mirror, and a third and fourth field effect transistors connected to form a second current mirror, wherein a property of the first transistor is mismatched relative to the second transistor such that the threshold voltage of the first transistor is significantly higher than the threshold voltage of the second transistor, and the drain current versus gate-source voltage responses of the first and second transistors have substantially different gradients for current levels at which the reference circuit is operated.
Suitably, the property of the first transistor is selected such that, for a particular voltage applied to the common gate of the first transistor and the second transistor, the second transistor operates substantially in its strong inversion saturation region whilst the first transistor operates substantially in its weak inversion saturation region.
Suitably, the mismatch is obtained by providing the first transistor with an oxide layer having a thickness which is greater than the oxide layer of the second transistor.
Suitably, the thickness of the oxide layer provided on the first transistor is at least twice the thickness of the oxide layer provided on the second transistor.
Suitably, the thickness of the oxide layer provided on the first transistor is at least 5 nanometers greater than the thickness of the oxide layer provided on the second transistor.
Suitably, the thickness of the oxide layer provided on the first transistor is at least 10 nanometers greater than the thickness of the oxide layer provided on the second transistor.
Suitably, the mismatch is obtained by providing more doping to the substrate of the first transistor than the substrate of the second transistor.
Suitably, the first transistor comprises a modified twin tub configuration, in which a well layer separating an upper tub layer and a substrate layer is omitted during fabrication such that the upper tub layer is located directly on the substrate layer, the upper tub layer thereby providing a substrate layer having increased doping.
Suitably, the third and fourth transistors are matched such that either side of the second current mirror is constrained to draw substantially the same current, the circuit having a stable operating point where the drain current versus gate-source voltages of the first and second transistors intersect.
Suitably, the third and fourth transistors are not matched, so that one side of the second current mirror is constrained to draw more current than the other side.
Suitably, the third and fourth transistors are field effect transistors, and the width of the channel of one of the transistors is selected to be different to the width of the channel of the other transistor so that that side of the current mirror is constrained to draw a current which is a ratio of the current on the other side.
Suitably, the third and fourth transistors are field effect transistors, and the length of the channel of one of the transistors is selected to be different to the length of the channel of the other transistor so that that side of the current mirror is constrained to draw a current which is a ratio of the current on the other side.
Suitably, the third and fourth transistors are bipolar transistors.
Suitably, the length of the first transistor is selected to be different to the length of the second transistor.
Suitably, the width of the first transistor is selected to be different to the width of the second transistor.
Suitably, a reference voltage is obtained from the common gate of the third and fourth transistors.
Suitably, a copy of the reference current is obtained by connecting a FET to the common gate of the third and the fourth transistor.
Suitably, the first and second transistors are p-channel field effect transistors, and the third and fourth transistors are n-channel field effect transistors.
A specific embodiment of the invention will now be described by way of example only, with reference to the accompanying figures in which:
In a conventional arrangement the field effect transistors M1 and M2 would be of different width and have a resistor in series with M1 such that, for a given gate voltage, the current provided from the drain of each of the transistors is equal. However, in the circuit shown in
In operation, M2 is turned on first and enters the square law saturation region (i.e. the rate of increase of current with respect to gate source voltage is quadratic). M1 is weakly turned on at a higher voltage, and operates in the weak inversion region (i.e. the rate of increase of current with respect to gate source voltage is exponential). Since M1 is turned on at a higher voltage than M2, and provides current which increases with a steeper gradient, it follows that there is a value of gate voltage for which both M1 and M2 provide the same output current. This is the stable operating point of the circuit, given that the gain of the current mirror comprising of M3 and M4 is unity.
It will be understood that the output current of the reference circuit is stable, under stable ambient conditions. If the ambient conditions, e.g. the temperature varies, then the output current of the reference circuit may vary accordingly. This property may be used to provide a reference current which tracks changes in ambient conditions.
M1 enters the weak inversion region at a gate source voltage of around 1.6V, and remains in the weak inversion region over the range of currents represented in
The currents provided by M1 and M2 intersect at a value of approximately 3.21 μA for a gate source voltage of approximately 2.25V. This intersection provides a current which satisfies the operating requirements of the current mirror formed by n-channel transistors M3 and M4, i.e. that the current provided by each side of the circuit is equal. The intersection is a stable operating point for the circuit, and the circuit will consequently generate a fixed current of approximately 3.2 μA which is independent of the voltage Vdd at the bias rail.
The currents provided by M1 and M2 will not intersect at higher values, since the gradient of M1 will never be less than the gradient of M2 (M1 will eventually enter the square law saturation region). This means that the circuit has no stable operating points at higher currents. The currents provided by M1 and M2 will converge at zero gate-source voltage and zero current, therefore this could be considered to be a stable operating point of the circuit. The circuit will leave the zero current operating point given a sufficient voltage at Vdd and an initial startup charge at the gates of M3 and M4 and move to the stable intended operating point which generates the approximately 3.2 μA current, in this particular case. Leakage currents can sometime be sufficient to start-up the circuit.
Different current settings for the circuit may be achieved by scaling the response of M1 and M2 with respect to each other. For example, by providing M1 with a thicker oxide layer, the voltage at which M1 is weakly turned on will increase, and the current provided by the stable operating point will increase.
The quadratic behaviour of a field effect transistor operating in the square law saturation region is determined by the following:
where Id is the drain current, μh is the mobility of holes, Cox is the capacitance per unit area of the gate, W is the width of the channel, L is the length of the channel, Vgs is the gate/source voltage and VT is the threshold voltage.
The exponential behaviour of a field effect transistor operating in the weak inversion region is determined by the following:
where Id is the drain current, Ik is a constant, W is the width of the channel, L is the length of the channel, n is a constant, Vgs is the gate/source voltage and VT is the threshold voltage.
From the above it is clear that different current settings for the circuit may be achieved by modifying the channel width and/or the channel length of the transistors, and in particular by selecting the ratio of width to length. For example, referring to
The modification of the width or length need not be confined to the p-channel transistors M1 and M2, but may instead be used to adjust the properties of the n-channel transistors M3 and M4. For example, instead of matching M3 and M4, the channel width of M3 could be double of M4. This would constrain the circuit to provide twice as much current on the left hand side as on the right hand side. The stable operating point of the circuit would then be at approximately 2.13 volts as indicated by the vertical line A in
One suitable combination of channel widths and channel lengths is as follows:
Width = 2
Width = 40
Length = 10
Length = 5
Width = 2
Width = 2
Length = 20
Length = 20
Since M3 and M4 are matched, the stable operating point of the circuit is the point at which M1 and M2 provide the same current. The large channel width of M1 compared to M2 is necessary since the threshold voltage of M1 is much higher than that of M2.
The circuit may be used to generate a reference current via a copy of the drain current of M3 by connecting yet another matched device to the common gate of M1 and M2. Alternatively, the gate voltage of M1 and M2, can be used as a reference voltage.
In the described embodiment of the invention the transistors are field effect transistors. It will be appreciated that any suitable field effect transistors may be used. M1 and M2 could be bipolar transistors, for example where biCMOS is used.
The invention may be implemented as a single semiconductor chip, making it particularly suited to biomedical applications.
In the above mentioned semiconductor technology process, the chip has a standard feature size of 0.8 μm, and the low voltage field effect transistors provided on the chip has a typical standard gate oxide layer of around 17 nm. Where the invention is used, the mid-gate oxide layer of transistor M1 is approximately 50 nm. Some semiconductor manufacturers already provide transistors with oxide layers of similar this thickness, for use in high voltage processors (high voltage typically means around 20–30V rather than a normal voltage of around 5.5V). It would be possible therefore to manufacture a chip which incorporates the invention using existing techniques.
For a semiconductor chip having a standard feature size of 0.25 μm, the field effect transistors provided on the chip will have an oxide layer of around 5 or 6 nm. Where the invention is used, the oxide layer of transistor M1 could be approximately 13 nm. Again, a chip that incorporates the invention could be manufactured using existing manufacturing processes that support two different voltages e.g. 3V and 5V devices.
An alternative, or additional, means of modifying the threshold voltage of transistor M1 is by modifying the doping of the substrate. An increase of the doping of the substrate will cause a corresponding increase of the threshold voltage required to invert the channel of the transistor.
One manner in which the substrate doping may be increased in a silicon chip is by modifying a conventional twin tub field effect transistor configuration. A prior art twin tub FET is shown in
The invention may be implemented by omitting the n-well 15 during fabrication of the FET, so that the p-well layer lies directly over the p-substrate layer. The effect of doing this will be to provide a conventionally configured FET having a substrate layer which is more strongly doped than the substrate layers of other FET's provided on the chip. The threshold of the FET is increased by the higher doping of the p-substrate.
Use of technologies with two different gate oxide thickness is preferred over modification of the doping because the oxide thickness is better controlled and supplied device models are more accurate.
Where different gate oxide thickness devices are used to implement the invention, the thicknesses should be carefully controlled in order to ensure that the invention functions correctly. The thickness of the oxide layer provides separation of the current versus gate source voltage curves as shown in
It will be appreciated by those skilled in the art that the circuit shown in
|Cited Patent||Filing date||Publication date||Applicant||Title|
|US4300091||Jul 11, 1980||Nov 10, 1981||Rca Corporation||Current regulating circuitry|
|US4777019||Apr 11, 1986||Oct 11, 1988||Thomas Dandekar||Biosensor|
|US5103159 *||Oct 19, 1990||Apr 7, 1992||Sgs-Thomson Microelectronics S.A.||Current source with low temperature coefficient|
|US5466348||Mar 12, 1993||Nov 14, 1995||Holm-Kennedy; James W.||Methods and devices for enhanced biochemical sensing|
|US5632957||Sep 9, 1994||May 27, 1997||Nanogen||Molecular biological diagnostic systems including electrodes|
|US5635869 *||Sep 29, 1995||Jun 3, 1997||International Business Machines Corporation||Current reference circuit|
|US5644216 *||May 31, 1995||Jul 1, 1997||Sgs-Thomson Microelectronics, S.A.||Temperature-stable current source|
|US5793248 *||Jul 31, 1996||Aug 11, 1998||Exel Microelectronics, Inc.||Voltage controlled variable current reference|
|US5827482||Aug 20, 1996||Oct 27, 1998||Motorola Corporation||Transistor-based apparatus and method for molecular detection and field enhancement|
|US5939933 *||Feb 13, 1998||Aug 17, 1999||Adaptec, Inc.||Intentionally mismatched mirror process inverse current source|
|US6015714||Jun 16, 1998||Jan 18, 2000||The United States Of America As Represented By The Secretary Of Commerce||Characterization of individual polymer molecules based on monomer-interface interactions|
|US6060327||May 14, 1997||May 9, 2000||Keensense, Inc.||Molecular wire injection sensors|
|US6096610||Jan 8, 1997||Aug 1, 2000||Intel Corporation||Transistor suitable for high voltage circuit|
|US6333662 *||Dec 22, 1999||Dec 25, 2001||Kabushiki Kaisha Toshiba||Latch type level shift circuit|
|US6413792||Aug 31, 2000||Jul 2, 2002||Eagle Research Development, Llc||Ultra-fast nucleic acid sequencing device and a method for making and using the same|
|US6953958||Mar 19, 2003||Oct 11, 2005||Cornell Research Foundation, Inc.||Electronic gain cell based charge sensor|
|US20010020844||Dec 27, 2000||Sep 13, 2001||Shunsuke Andoh||Voltage generating circuit and reference voltage source circuit employing field effect transistors|
|US20030186262||Mar 1, 2001||Oct 2, 2003||Fabrice Cailloux||Novel dna chips|
|US20040262636||Apr 8, 2004||Dec 30, 2004||The Regents Of The University Of California||Fluidic nanotubes and devices|
|US20050032075||Oct 1, 2003||Feb 10, 2005||Hidenobu Yaku||Method of detecting primer extension reaction, method of discriminating base type, device for discriminating base type, device for detecting pyrophosphate, method of detecting nucleic acid and tip for introducing sample solution|
|US20050062093||Nov 11, 2002||Mar 24, 2005||Kazuaki Sawada||Fet type sensor, ion density detecting method comprising this sensor, and base sequence detecting method|
|EP0483537A2||Oct 4, 1991||May 6, 1992||TEMIC TELEFUNKEN microelectronic GmbH||Current source circuit|
|EP0992871A2||Oct 1, 1999||Apr 12, 2000||CSELT Centro Studi e Laboratori Telecomunicazioni S.p.A.||CMOS circuit for generating a current reference|
|FR2732129A1||Title not available|
|1||Sakurai and Husimi, "Real-Time Monitoring of DNA Polymerase Reactions by a Micro ISFET pH Sensor," Anal. Chem, 64, pp. 1996-1997 (1992).|
|Citing Patent||Filing date||Publication date||Applicant||Title|
|US7521993 *||May 13, 2005||Apr 21, 2009||Sun Microsystems, Inc.||Substrate stress signal amplifier|
|US7859328||Mar 10, 2009||Dec 28, 2010||Oracle America, Inc.||Substrate stress measuring technique|
|US8564275 *||Jun 25, 2010||Oct 22, 2013||The Regents Of The University Of Michigan||Reference voltage generator having a two transistor design|
|US8969002||Oct 1, 2012||Mar 3, 2015||Genapsys, Inc.||Methods and systems for electronic sequencing|
|US9150915||Jan 13, 2015||Oct 6, 2015||Genapsys, Inc.||Systems and methods for automated reusable parallel biological reactions|
|US9187783||Oct 4, 2011||Nov 17, 2015||Genapsys, Inc.||Systems and methods for automated reusable parallel biological reactions|
|US9274077||Mar 15, 2013||Mar 1, 2016||Genapsys, Inc.||Systems and methods for genetic and biological analysis|
|US9399217||May 27, 2012||Jul 26, 2016||Genapsys, Inc.||Chamber free nanoreactor system|
|US9434983||Nov 15, 2013||Sep 6, 2016||The Board Of Trustees Of The Leland Stanford Junior University||Nano-sensor array|
|US9533305||Aug 25, 2015||Jan 3, 2017||Genapsys, Inc.||Systems and methods for automated reusable parallel biological reactions|
|US20090273393 *||Mar 10, 2009||Nov 5, 2009||Sun Microsystems, Inc.||Substrate stress measuring technique|
|US20100327842 *||Jun 25, 2010||Dec 30, 2010||The Regents Of The University Of Michigan||Reference voltage generator having a two transistor design|
|US20150277470 *||Mar 26, 2015||Oct 1, 2015||Megachips Corporation||Current mirror circuit and receiver using the same|
|U.S. Classification||327/543, 323/315|
|International Classification||G05F3/24, G05F3/26|
|Jul 14, 2005||AS||Assignment|
Owner name: FREEWILL, SINGAPORE
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:TOUMAZOU, CHRISTOFER;GEORGIOU, JULIUS;REEL/FRAME:016929/0219;SIGNING DATES FROM 20050131 TO 20050403
|Oct 31, 2006||AS||Assignment|
Owner name: DNA ELECTRONICS LIMITED, UNITED KINGDOM
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:FREEWILL;REEL/FRAME:018457/0524
Effective date: 20060502
|Jan 6, 2011||FPAY||Fee payment|
Year of fee payment: 4
|Dec 31, 2014||FPAY||Fee payment|
Year of fee payment: 8
|Jun 17, 2016||AS||Assignment|
Owner name: DNAE GROUP HOLDINGS LIMITED, UNITED KINGDOM
Free format text: CHANGE OF NAME;ASSIGNOR:DNA ELECTRONICS LIMITED;REEL/FRAME:038946/0895
Effective date: 20150902