|Publication number||US7272021 B2|
|Application number||US 11/407,699|
|Publication date||Sep 18, 2007|
|Filing date||Apr 20, 2006|
|Priority date||Jan 24, 1997|
|Also published as||US20060262575, US20080175024, US20110176333|
|Publication number||11407699, 407699, US 7272021 B2, US 7272021B2, US-B2-7272021, US7272021 B2, US7272021B2|
|Inventors||Martin F. Schlecht, Richard W. Farrington|
|Original Assignee||Synqor, Inc.|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (49), Non-Patent Citations (21), Referenced by (25), Classifications (8), Legal Events (11)|
|External Links: USPTO, USPTO Assignment, Espacenet|
This application is a Continuation-in-Part of U.S. application Ser. No. 10/729,430, filed on Dec. 5, 2003 now U.S. Pat. No. 7,050,309, which claims the benefit of U.S. Provisional Application No. 60/431,673, filed Dec. 6, 2002 and a Continuation-in-Part to U.S. application Ser. No. 10/812,314, filed Mar. 29, 2004 now U.S. Pat. No. 7,072,190, which is a continuation of application Ser. No. 10/359,457, filed Feb. 5, 2003 now U.S. Pat. No. 6,731,520, which is a continuation of application Ser. No. 09/821,655, filed Mar. 29, 2001, now U.S. Pat. No. 6,594,159, which is a divisional of application Ser. No. 09/417,867, filed Oct. 13, 1999, now U.S. Pat. No. 6,222,742, which is a divisional of Ser. No. 09/012,475, filed Jan. 23, 1998, now U.S. Pat. No. 5,999,417, which claims the benefit of U.S. Provisional Application 60/036,245 filed Jan. 24, 1997. The entire teachings of the above applications are incorporated herein by reference.
This invention pertains to switching power converters. A specific example of a power converter is a DC-DC power supply that draws 100 watts of power from a 48 volt DC source and converts it to a 5 volt DC output to drive logic circuitry. The nominal values and ranges of the input and output voltages, as well as the maximum power handling capability of the converter, depend on the application.
It is common today for switching power supplies to have a switching frequency of 100 kHz or higher. Such a high switching frequency permits the capacitors, inductors, and transformers in the converter to be physically small. The reduction in the overall volume of the converter that results is desirable to the users of such supplies.
Another important attribute of a power supply is its efficiency. The higher the efficiency, the less heat that is dissipated within the supply, and the less design effort, volume, weight, and cost that must be devoted to remove this heat. A higher efficiency is therefore also desirable to the users of these supplies.
A significant fraction of the energy dissipated in a power supply is due to the on-state (or conduction) loss of the diodes used, particularly if the load and/or source voltages are low (e.g. 3.3, 5, or 12 volts). In order to reduce this conduction loss, the diodes are sometimes replaced with transistors whose on-state voltages are much smaller. These transistors, called synchronous rectifiers, are typically power MOSFETs for converters switching in the 100 kHz and higher range.
The use of transistors as synchronous rectifiers in high switching frequency converters presents several technical challenges. One is the need to provide properly timed drives to the control terminals of these transistors. This task is made more complicated when the converter provides electrical isolation between its input and output because the synchronous rectifier drives are then isolated from the drives of the main, primary side transistors. Another challenge is the need to minimize losses during the switch transitions of the synchronous rectifiers. An important portion of these switching losses is due to the need to charge and discharge the parasitic capacitances of the transistors, the parasitic inductances of interconnections, and the leakage inductance of transformer windings.
In certain embodiments of the invention, a power converter system comprises a normally non-regulating isolation stage and a plurality of non-isolating regulation stages, each receiving the output of the isolation stage and regulating a regulation stage output. The non-regulating isolation stage may comprise a primary winding circuit and a secondary winding circuit coupled to the primary winding circuit. The secondary winding circuit comprises a secondary transformer winding in series with a controlled rectifier having a parallel uncontrolled rectifier. A control circuit controls duty cycle of the primary winding circuit, the duty cycle causing substantially uninterrupted control of power through the primary and secondary winding circuits during normal operation.
The duty cycle of the primary winding circuit may be reduced to cause freewheeling periods in other than normal operation. Duty cycle might be reduced during the startup or to limit current and may be a function of sensed current.
The primary winding circuit may include a single primary winding, and the secondary winding circuit may include plural secondary windings coupled to the single primary winding. The primary winding may be in a full bridge circuit having a capacitor in series with the primary winding. In one implementation of the full bridge circuit, during freewheeling, only two top FETs or two bottom FETs are turned off.
A control signal of the controlled rectifier may be derived from a waveform of the secondary winding circuit. The secondary winding circuit may include a filter inductor and have a capacitor coupled across its output.
The isolation stage may be a step down stage. For example, it may provide an output of about 12 volts from a DC power source that provides a voltage varying over the range of 36–75 volts. The regulation stages may be down converters to provide outputs of voltage levels to drive logic circuitry. A regulation stage output may, for example, be 5 volts or less, such as 3.3 volts.
The foregoing and other objects, features and advantages of the invention will be apparent from the following more particular description of preferred embodiments of the invention, as illustrated in the accompanying drawings in which like reference characters refer to the same parts throughout the different views. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention.
A description of preferred embodiments of the invention follows.
During the second half of the cycle, MOSFETs 102 and 104 are turned on while MOSFETs 101 and 103 are left off, and the voltage VB is applied negatively across the transformer's primary winding. This negative polarity causes MOSFET 106 to be turned on, MOSFET 105 to be turned off, and power to flow into the primary winding and out of the second secondary winding 109 to the output across capacitor 110.
The secondary windings are not tightly coupled to each other, as indicated with the parasitic inductances 113 and 114, to achieve the advantages discussed in the '417 patent. A similar setup was shown in the topology of
Care must be taken in this isolation stage topology to insure that the magnetizing inductance of the transformer does not saturate. One way to do this is to place a large capacitor 215 in series with the primary winding, as shown in
The filters at the output of the isolation stages in the '417 patent are composed of one or more capacitive and inductive elements. When the isolation stage is voltage-fed, it may be desirable to have the output filter begin with an inductor 316, as shown in
Under variable duty cycle control, the percentage of the overall cycle (the duty cycle) that MOSFETs 101 and 103 (or MOSFETs 102 and 104) conduct is reduced from the 50% value described above. For the remaining, freewheeling fraction of the half-cycle, either all of the primary-side MOSFETs are turned off, or at least the two top MOSFETs 101 and 104 or the two bottom MOSFETs 102 and 103 are turned off. During the freewheeling part of the cycle, both diodes 111 and 112 conduct the current flowing through inductor 316, and the voltage across the transformer windings is approximately zero. As is well know, this additional portion of the cycle permits the output voltage to be less than VB divided by the transformer's turns-ratio. How much less depends on the duty cycle. Since during normal operation the isolation stage is operated at a fixed duty cycle in which power is always flowing from input to output (except during the brief switch transitions), the value of inductor 316 can be relatively small to achieve an acceptable output ripple. This reduces the size, cost, and power dissipation of this inductor compared to what it might have been. During those times when the isolation stage is operated under a variable duty cycle, the ripple in the inductor current may then become large, but the larger output voltage ripple that results can usually be tolerated for start-up and short-circuit conditions.
As mentioned above, during the freewheeling part of the cycle the diodes are carrying the inductor current. This is because the gate drive scheme shown in
If the output voltage is high, then it may be desirable to use a capacitive divider technique described in the '417 patent to reduce the voltages applied to the gates of the MOSFET synchronous rectifiers below that of the voltages appearing at Nodes A and B.
In addition, since the converter of
Note in this schematic that the IC labeled U100 is a pulse width modulator (PWM) control chip that is normally operated such that the gate drive signals that pass through gate drivers U101 and U105 give the fixed duty cycle operation of the full-bridge described above. If the current sensing amplifier U104-A senses that the current flowing on the primary side of the circuit exceeds a threshold value, it commands the PWM control chip to reduce its duty cycle by an amount determined by how large the current gets above the threshold value. This provides a current limiting scheme for the product that protects against a short-circuit condition.
Note also that comparator U106-A senses the duty cycle output of the PWM control chip, and compares it to a threshold. If the duty cycle falls below this threshold value, the output of the comparator causes the PWM control IC to shut down. The circuitry around this comparator, including transistors Q111 and Q114, provides a latching mechanism such that the PWM control IC remains off once this condition is observed.
As described in the '417 patent and illustrated in
The DC power source to the full bridge primary circuit may provide a voltage that varies over the range of 36–75 volts. The output of the isolation stage may be 12 volts, and the regulation stage output may be 5 volts or less. In particular, the regulation stage output may be 3.3 volts. Typically, the regulation stage output is of a voltage level to drive logic circuitry.
Because the isolation stage uses synchronous rectifiers, it is possible for the current to flow from the output back to the input if, for a given input voltage and duty cycle, the output voltage is too high. This condition might, for example, occur during start-up where the duty cycle is slowly raised from its minimum value to its maximum value, but the output capacitor is already pre-charged to a high voltage, perhaps because it had not fully discharged from a previous on-state condition. It might also occur when the input voltage suddenly decreases while the output voltage remains high due to the capacitors connected to this node.
The negative current that results could cause destructive behavior in the converter or in the system if it is not kept small enough.
One way to avoid this condition is to turn off either just the top two primary-side MOSFETs 101 and 104, or just the bottom two primary-side MOSFETs 102 and 103, during the freewheeling period, as described above. By leaving the other two primary-side MOSFETs on, the voltage across the primary and secondary windings of the transformer is guaranteed to be essentially zero during the freewheeling period. Given the gate drive scheme shown in
With the controlled rectifiers off, negative current cannot flow during the freewheeling period. Negative current can flow during the non-freewheeling part of the cycle, but since it must always start at zero, its value is limited to the ripple that the inductor permits, which is typically small enough to not cause a problem. This negative current will be reset to zero at the start of each freewheeling period, either by providing a clamp circuit, as shown in
To limit the negative current, the isolation stage could operate in a reduced duty-cycle mode. While the control circuit is typically designed to achieve this mode during start-up and shutdown of the isolation stage, it is not the normal mode of operation. If, during normal operation, the input voltage drops suddenly, a large negative current can flow because there are no freewheeling periods.
To avoid this condition, the current flowing through the converter can be sensed, either by sensing the load current directly, or by sensing a signal indicative of the load current. When the load current falls below some threshold, the duty cycle of the isolation stage can be reduced from its maximum value to provide freewheeling periods. Given the drive scheme for the primary-side MOSFETs outlined above, the negative current will then be kept small since the controlled rectifiers will be turned off for a portion of the cycle.
While this invention has been particularly shown and described with references to preferred embodiments thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the scope of the invention encompassed by the appended claims. For example, whereas the Figures show the secondary side rectification circuit arranged in a center tapped configuration with two secondary windings and two synchronous rectifiers, as is well known it could be a full wave rectification configuration. One could use a full-bridge rectification circuit in which there is only one secondary winding and four synchronous rectifiers. Such a circuit reduces voltage stress on the synchronous rectifiers when they are off by a factor of two during normal operation of the converter.
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|U.S. Classification||363/17, 363/97|
|Cooperative Classification||Y02B70/1475, H02M3/33592, H02M3/33561|
|European Classification||H02M3/335M, H02M3/335S2S|
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