|Publication number||US7548046 B1|
|Application number||US 11/945,197|
|Publication date||Jun 16, 2009|
|Filing date||Nov 26, 2007|
|Priority date||May 18, 2005|
|Also published as||US7317305, US7859238|
|Publication number||11945197, 945197, US 7548046 B1, US 7548046B1, US-B1-7548046, US7548046 B1, US7548046B1|
|Inventors||Anthony Stratakos, Jieli Li, Biljana Beronja, David Lidsky, Michael McJimsey, Aaron Schultz, Charles R. Sullivan, Charles Nickel|
|Original Assignee||Volterra Semiconductor Corporation|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (13), Non-Patent Citations (10), Referenced by (27), Classifications (6), Legal Events (1)|
|External Links: USPTO, USPTO Assignment, Espacenet|
This application is a continuation of U.S. patent application Ser. No. 11/131,761 filed 18 May 2005, now U.S. Pat. No. 7,317,305 the disclosure of which is incorporated herein by reference.
The invention relates generally to switching power converters, and particularly to multiphase DC-to-DC converters having coupled inductors.
Switching DC-to-DC power converters having a multi-phase coupled inductor topology like that described in U.S. Pat. No. 6,362,986 to Schultz, et al., the disclosure of which is incorporated herein by reference, are known in the art. These converters have advantages, including reduced ripple current in the inductors and switches allowing reduced per-phase inductance or reduced switching frequency, over converters having conventional multi-phase dc-dc converter topologies. Switching DC-to-DC converters as described in U.S. Pat. No. 6,362,986 typically operate in continuous conduction mode (CCM) for high efficiency while driving heavy loads.
DC-DC power converters are often used in applications where the load may vary considerably as a system operates.
For example, the processor of a modern notebook computer may demand tens to more than one hundred amps of current when performing processor-intensive computation at maximum clock rate, while it needs much less current, possibly only a few milliamps, when the system is idle. When a DC-DC converter is designed to power such a processor, the inductors, capacitors, and switching transistors of the converter are typically designed to handle the maximum sustained current required by the processor without overheating.
There are many other applications for power converters where converter load current levels may vary over time. Variation between maximum and minimum load current of factors of hundreds to thousands are not unusual.
Continuous Conduction and Discontinuous Conduction Modes of Operation
Most DC-DC converters that deliver high current operate in Continuous Conduction Mode (CCM). CCM is an operating mode wherein the high and low side switches keep switching on and off alternatively and the current in the output inductor keeps ramping up and down continuously. In CCM in a synchronous DC-DC converter, inductor current never stops flowing, although it may cross through zero. At high current outputs, CCM enables the converter to deliver high current with high efficiency.
In CCM, the inductor carries significant AC current even at low loads. Therefore, there are power losses, such as those due to resistive loss in converter switches and inductor windings, and those due to charging and discharging the parasitic capacitors of the switches, that are present even when the converter operates at low load.
Therefore, with CCM operation, the switching loss and the AC current related loss do not scale down with decreasing load current and they may become a significant part of the total power absorbed by the converter when the load current is small. Since many systems spend considerable portions of their operating lifetime operating at low power levels, they may waste considerable energy over their lifetimes. It is especially important in battery powered systems that DC-DC converters operate at high efficiency over the entire range of possible output power demand to optimize battery life.
Discontinuous conduction mode (DCM) is an operating mode of a DC-DC converter where energy is delivered to the output only when needed. When energy delivery is not required, the switches stop switching and remain off until energy delivery is required. When the switches are off, the current in the inductor remains zero and an output capacitor, the key component of an output filter, supports the output current during the time both switches are off. In this way, switching loss and AC current related loss scale down with decreasing load current and the DC-DC converter maintains high efficiency even at light load.
A multi-phase, coupled-inductor, DC-DC voltage converter operates in discontinuous conduction mode (DCM) when the system is operated at low output power levels. The converter achieves high efficiency in the described DCM operation at low output power levels.
A two-phase buck-type voltage converter as described in U.S. Pat. No. 6,362,986 to Schultz (
The prior-art voltage converter of U.S. Pat. No. 6,362,986 operates in continuous conduction mode (CCM). During CCM operation controller 114 monitors an output voltage at capacitor 116. In periodic steady-state, controller 114 produces pulses by turning on alternately one of the first 106 or second 108 high-side switches, which connect to the high side supply voltage 130, thereby building a current through the associated winding 102 or 104; as this current builds, magnetic coupling also produces a current through the undriven, coupled, of windings 104 or 102, and the associated low-side switch 112 or 110 is turned on such that currents through both windings 102, 104 can charge the filter 116. At the conclusion of this part of the cycle, the associated high-side switch 106 or 108 is turned off and low-side switch 110 or 112 is turned on, such that both low-side switches 110, 112 are on. Winding currents will then decrease, and may reverse. As current decreases, the controller 114 may turn on a different of the high-side switches 108 or 106, while turning off the corresponding low-side switch 112 or 110, thereby building a current through the associated winding 104 or 102; as this current builds, magnetic coupling also produces a current through the now un-driven winding of 102 or 104, and the associated low-side switch 110 or 112 is turned on such that currents through both windings 104, 102 can charge the filter 116. This cycle repeats indefinitely and without pause as the prior converter operates in continuous conduction mode.
There is generally a low-side overshoot diode 122, which is often a parasitic diode component of each low-side switch 110, 112; in non-synchronous converters the low side overshoot diode 122 replaces the low-side switch 110, 112. Since high efficiency at low operating voltages requires that dissipation of power through the forward current voltage drop in diodes be avoided, it is preferable that low-side current be carried through active low-side switches 110, 112; N-channel FET transistors are typically used for this purpose.
Output voltage control in a typical converter is performed by varying the length of time each high-side switch 108, 106 is held on during each energy delivery pulse, as needed to maintain a suitable voltage at filter 116. This can be done using voltage mode control, current mode control, or any other control method known in the art.
A two-phase embodiment of the present voltage converter, as illustrated in
Current sensors or estimators 214 may be provided for monitoring or estimating current in the switches or windings to permit precise control of switching at zero current points.
The major difference between present two-phase converter in
After high-side switch 206 has been on for a pulse-width Tpw delay 573, the high side switch 206 is turned OFF 574, and the corresponding low side switch 210 is turned ON. After waiting 576 for winding 202, 204 currents to decay and eventually reach zero, low side switches 210, 212 are turned 578 OFF. In an embodiment, the cycle repeats as controller 214 monitors 570 voltage Vout at filter 216, testing for that voltage to decay to a voltage below the threshold Vth.
In an embodiment, pulse-width Tpw is determined by monitoring or estimating a current, such as a current in high side switch 206, low-side switch 212, winding 202, or winding 204, and turning off 574 the high side switch 206, upon the monitored current reaching a predetermined current. In an alternative embodiment, Tpw may also be determined as a preprogrammed time duration. Or it may be controlled by monitoring other signals, or through hysteresis such as by comparing Vout against a second threshold.
In a bit more detail, during each energy delivery pulse, if there were perfect magnetic coupling among windings 202, 204 of both phases, then the current in windings 202, 204 will be identical and will ramp up from zero at a rate of:
At the end of this energy delivery pulse, Tpw, the current in each winding peaks. The high-side switch 206 then turns OFF and the low-side switch 210 turns ON to pick up the inductive winding current. At that time, the low-side switch 212 in the other phase remains ON. During this interval, the winding currents in both phases have a slope equal to:
The low-side switches of both phases are then turned off when their winding current eventually falls to zero.
During the full period in single-sided DCM, the alternate-phase high side switch 208 remains OFF. In practice, the low-side switch is often an NMOS transistor. This scheme has the advantage that only one of the high-side switches is switched per energy delivery pulse, conserving switching energy, while both phases deliver output current.
For practical DCM implementation with non-ideal coupling between the windings, especially during operation at medium output currents, it may be desirable to operate the system in a phase-alternating DCM mode, as illustrated in
Phase-alternating mode is a two-phase example of a phase-rotating DCM mode as hereinafter described. The controller 214 monitors 602 voltage Vout at filter 216, testing for that voltage to decay to a voltage below a threshold Vth. When Vout drops below Vth, a first energy delivery pulse begins 608. During a first energy delivery pulse, a high-side switch 206 is turned ON, and low-side switch 212 for a magnetically coupled opposite phase is turned ON. As current builds in winding 202, a current is induced in winding 204.
After high-side switch 206 has been on for a pulse-width Tpw delay 610, the high side switch 206 is turned OFF 612, and the corresponding low side switch 210 is turned ON. After waiting 614 for winding 202, 204 currents to decay and eventually reach zero, low side switches 210, 212 are turned OFF 616.
The controller 214 continues to monitor 618 voltage Vout at filter 216. When Vout drops below Vth, a second energy delivery pulse begins 620. During a second energy delivery pulse, the alternate phase high-side switch 208 is turned ON, and low-side switch 210 for a magnetically coupled opposite phase is turned ON. As current builds in winding 204, a current is induced in winding 202.
After high-side switch 208 has been on for a pulse-width Tpw delay 622, the high side switch 208 is turned OFF 624, and the corresponding low side switch 212 is turned ON. After waiting 626 for winding 202, 204 currents to decay and eventually reach zero, low side switches 210, 212 are turned OFF 628.
In an embodiment, pulse-width Tpw is determined by monitoring or estimating a current, such as a current in high side switch 206, low side switch 212, high side switch 208, low side switch 210, winding 202, or winding 204, and turning off 612 the high side switch 206, 208 upon the monitored current reaching a predetermined peak current during DCM mode. In an alternative embodiment, Tpw may also be determined as a preprogrammed time duration. Or it may be controlled by monitoring other signals, or through hysteresis such as by comparing Vout against a second threshold.
Once current reaches zero, the cycle repeats and operation continues with the controller 214 monitoring 602 voltage Vout at filter 216, testing for that voltage to decay to a voltage below threshold Vth. When Vout drops below Vth, a next energy delivery pulse begins 608.
In an embodiment, during phase-alternating DCM mode the duration of OFF time between energy delivery pulses is compared against a lower limit L1. If the OFF time is below lower limit L1, indicating operation at high output current, the converter switches to operation in CCM. In an alternative embodiment, output current is directly monitored or estimated and compared to a limit.
Similarly, during CCM operation, output current is monitored or estimated. When the monitored or estimated current is less than a limit L2, the converter switches to operation in DCM.
In an alternative embodiment, when the converter operates at very low output currents, operation changes to the single-sided discontinuous conduction mode previously discussed with reference to
There may be more than two phases in a converter, in context of this document a converter having two or more phases is a multiphase converter.
In a system having more than two phases, in phase-alternating DCM operation, high-side switches are activated in sequence as required for the number of phases in operation. One or more high-side switches can be turned on simultaneously depending on the desired winding current ramp rates. Turnoff of low side switches occurs at or near respective zero-current points and need not be exactly simultaneous.
The system waits with zero current in the inductive windings when the output voltage Vout is greater than threshold, and therefore operates in discontinuous conduction mode.
In an embodiment of the converter, pulse width Tpw is dynamically adjusted based on operating conditions of the converter. In another embodiment, four phases are used, each phase having associated high and low side switches and inductor.
While two phases are illustrated in
The four-phase converter of
A more general embodiment of a multiphase converter system having coupled inductors is illustrated in
During rotating-phase DCM operation, energy pulses are provided in sequence by high-side switches 810, 812, 814, and 816 to their respective windings 802, 804, 806, and 808 together with any additional high-side switches not illustrated. The sequence pauses whenever Vout is greater than threshold Vth, but continues and repeats whenever Vout is less than threshold Vth. The converter remains in a low-current idle state when Vout is greater than or equal to the threshold. Multiple high-side switches may turn on simultaneously to increase the slew rate of the current in the windings. Each time a high-side switch, such as high side switch 810, turns ON to provide an energy delivery pulse through a driven winding 802, low-side switches 822, 824, and 826 coupled to those other windings 804, 806 that are magnetically coupled to the driven winding are turned ON. Each time an ON high-side switch turns OFF, the associated low-side switch 820 turns ON. The low side switches 820, 822, 824 and 826 then turn OFF when current decays to zero.
In an alternative embodiment of the multiphase converter of
For purposes of this document, the term “magnetically coupled” with reference to inductors shall mean magnetically coupled windings having a coupling coefficient σ greater than or equal to 3, where σ=Lm/L1 as defined in column 8 of U.S. Pat. No. 6,362,986 to Schultz, et al.
It is possible to reference characteristics of the illustrated embodiments with variables. In doing so, let P represent the number of phases in the converter system, where each phase incorporates at least a low-side switch and an output inductor, and typically incorporates a high side switch. Let N represent the number of phases having high side switches. Let C represent the number of phases that are magnetically coupled together. Also, let D represent the number of phases that are driven through high-side switches in a given operating mode of the converter, and of those S phases, let S represent the number of phases driven to the converter input voltage synchronously, with the remaining D-S phases being driven in rotation alternately with the first S phases.
The embodiment of
It is anticipated that coupled-inductor converters capable of operating in discontinuous conduction mode according to the present document may be built in many configurations such that P>=2, P>=N, P>=C>=2, P>=D, and S<D system in DCM operation.
In particular it is anticipated that D may be less than or equal to N in some low power modes. It is also anticipated that P may be an odd number, as an example a three phase converter embodying these concepts operating with three phases in rotation and having P=3, C=3, N=3, D=3, and S=1 should be practical.
It is also anticipated that an individual converter may have more than one discontinuous conduction operating mode, such as a mode where D=N and a mode where D is less than N, with S<=D, and that the converter may automatically change from a mode with a low D to a mode with a higher D as converter output load or slew rate requirement is increased.
While the invention has been particularly shown and described with reference to particular embodiments thereof, it will be understood by those skilled in the art that various other changes in the form and details may be made without departing from the spirit and scope of the invention. It is to be understood that various changes may be made in adapting the invention to different embodiments without departing from the broader inventive concepts disclosed herein and comprehended by the claims that follow.
|Cited Patent||Filing date||Publication date||Applicant||Title|
|US5570276||Nov 15, 1993||Oct 29, 1996||Optimun Power Conversion, Inc.||Switching converter with open-loop input voltage regulation on primary side and closed-loop load regulation on secondary side|
|US5912552||Feb 12, 1997||Jun 15, 1999||Kabushiki Kaisha Toyoda Jidoshokki Seisakusho||DC to DC converter with high efficiency for light loads|
|US5959443||Nov 14, 1997||Sep 28, 1999||Toko, Inc.||Controller circuit for controlling a step down switching regulator operating in discontinuous conduction mode|
|US5999433||Jan 12, 1998||Dec 7, 1999||Vpt, Inc.||Half-bridge DC to DC converter with low output current ripple|
|US6094035||Aug 20, 1999||Jul 25, 2000||Gain Technology Corporation||Amplifying power converter circuits|
|US6307356||Jun 18, 1998||Oct 23, 2001||Linear Technology Corporation||Voltage mode feedback burst mode circuit|
|US6362986||Mar 22, 2001||Mar 26, 2002||Volterra, Inc.||Voltage converter with coupled inductive windings, and associated methods|
|US6366066||Jun 16, 2000||Apr 2, 2002||Milton E. Wilcox||Circuit and method for reducing quiescent current in a switching regulator|
|US6693411||May 24, 2002||Feb 17, 2004||Cleansun Pty. Ltd.||Switch mode power stage|
|US6707281||Jun 28, 2002||Mar 16, 2004||Intel Corporation||Method and apparatus for operating a voltage regulator based on inductor current detection|
|US6853562||Jun 26, 2003||Feb 8, 2005||Optimum Power Conversion, Inc.||Voltage sense method and circuit which alleviate reverse current flow of current bi-directional converters|
|US7239530||Feb 17, 2005||Jul 3, 2007||Volterra Semiconductor Corporation||Apparatus for isolated switching power supply with coupled output inductors|
|US7317305 *||May 18, 2005||Jan 8, 2008||Volterra Semiconductor Corporation||Method and apparatus for multi-phase DC-DC converters using coupled inductors in discontinuous conduction mode|
|1||Analog Devices, "Multiphase IMVP-IV Core Controller for Mobile CPIs, ADP3205", copyright 2003, 22 pages.|
|2||Arbetter et al, "Control Method for Low-Voltage DC Power Supply in Battery-Powered Systems with Power Management", IEEE Power Electronics Specialist Conference, St. Louis, Missouri, Jun. 22-27, 1997, 7 pages.|
|3||U.S. Appl. No. 11/131,761; Advisory Action mailed Jul. 9, 2007.|
|4||U.S. Appl. No. 11/131,761; Amendment and Response filed Jul. 26, 2007.|
|5||U.S. Appl. No. 11/131,761; Issue Fee Payment; Nov. 13, 2007.|
|6||U.S. Appl. No. 11/131,761; Notice of Allowance mailed Aug. 13, 2007.|
|7||U.S. Appl. No. 11/131,761; Office Action mailed Apr. 26, 2007.|
|8||U.S. Appl. No. 11/131,761; Office Action mailed Dec. 7, 2006.|
|9||U.S. Appl. No. 11/131,761; Response to Office Action of Apr. 26, 2007.|
|10||U.S. Appl. No. 11/131,761; Response to Office Action of Dec. 7, 2006.|
|Citing Patent||Filing date||Publication date||Applicant||Title|
|US7859238 *||May 18, 2009||Dec 28, 2010||Volterra Semiconductor Corporation||Method and apparatus for multi-phase DC-DC converters using coupled inductors in discontinuous conduction mode|
|US7898236 *||Aug 15, 2008||Mar 1, 2011||Intersil Americas Inc.||Varying operation of a voltage regulator, and components thereof, based upon load conditions|
|US7994888||Dec 21, 2009||Aug 9, 2011||Volterra Semiconductor Corporation||Multi-turn inductors|
|US8125207||Nov 23, 2010||Feb 28, 2012||Intersil Americas Inc.||Varying operation of a voltage regulator, and components thereof, based upon load conditions|
|US8174348||May 24, 2010||May 8, 2012||Volterra Semiconductor Corporation||Two-phase coupled inductors which promote improved printed circuit board layout|
|US8299885||May 13, 2011||Oct 30, 2012||Volterra Semiconductor Corporation||Method for making magnetic components with M-phase coupling, and related inductor structures|
|US8330567||Jan 14, 2010||Dec 11, 2012||Volterra Semiconductor Corporation||Asymmetrical coupled inductors and associated methods|
|US8362867||Jan 29, 2013||Volterra Semicanductor Corporation||Multi-turn inductors|
|US8547076||Mar 10, 2011||Oct 1, 2013||Volterra Semiconductor Corporation||Multiphase control systems and associated methods|
|US8674802||Oct 7, 2011||Mar 18, 2014||Volterra Semiconductor Corporation||Multi-turn inductors|
|US8779885||Mar 10, 2013||Jul 15, 2014||Volterra Semiconductor Corporation||Method for making magnetic components with M-phase coupling, and related inductor structures|
|US8786395||Mar 10, 2013||Jul 22, 2014||Volterra Semiconductor Corporation||Method for making magnetic components with M-phase coupling, and related inductor structures|
|US8836461||Mar 10, 2013||Sep 16, 2014||Volterra Semiconductor Corporation||Method for making magnetic components with M-phase coupling, and related inductor structures|
|US8847722||Dec 21, 2012||Sep 30, 2014||Volterra Semiconductor Corporation||Method for making magnetic components with N-phase coupling, and related inductor structures|
|US8860390||Jun 4, 2010||Oct 14, 2014||Apple Inc.||Switching power supply opposite polarity inductor arrangement|
|US8890644||Apr 30, 2012||Nov 18, 2014||Volterra Semiconductor LLC||Two-phase coupled inductors which promote improved printed circuit board layout|
|US8975995||Aug 29, 2012||Mar 10, 2015||Volterra Semiconductor Corporation||Coupled inductors with leakage plates, and associated systems and methods|
|US9019064||Oct 29, 2012||Apr 28, 2015||Volterra Semiconductor Corporation||Method for making magnetic components with M-phase coupling, and related inductor structures|
|US9147515||May 13, 2014||Sep 29, 2015||Volterra Semiconductor LLC||Method for making magnetic components with M-phase coupling, and related inductor structures|
|US9281115||Feb 3, 2014||Mar 8, 2016||Volterra Semiconductor LLC||Multi-turn inductors|
|US20090256535 *||Aug 15, 2008||Oct 15, 2009||Intersil Americas Inc.||Varying operation of a voltage regulator, and components thereof, based upon load conditions|
|US20110062930 *||Nov 23, 2010||Mar 17, 2011||Intersil Americas Inc.||Varying operation of a voltage regulator, and components thereof, based upon load conditions|
|US20110148560 *||Jun 23, 2011||Alexandr Ikriannikov||Two-Phase Coupled Inductors Which Promote Improved Printed Circuit Board Layout|
|US20110169476 *||Jan 14, 2010||Jul 14, 2011||Alexandr Ikriannikov||Asymmetrical Coupled Inductors And Associated Methods|
|US20110216560 *||Mar 4, 2010||Sep 8, 2011||Sheng Ye||Two stage isolated switch-mode ac/dc converter|
|USRE45773||Jan 31, 2014||Oct 20, 2015||Intersil Americas Inc.||Varying operation of a voltage regulator, and components thereof, based upon load conditions|
|WO2011088048A2||Jan 11, 2011||Jul 21, 2011||Volterra Semiconductor Corporation||Asymmetrical coupled inductors and associated methods|
|Cooperative Classification||H02M3/157, H02M3/1584|
|European Classification||H02M3/158P, H02M3/157|