|Publication number||US7595648 B2|
|Application number||US 11/565,454|
|Publication date||Sep 29, 2009|
|Filing date||Nov 30, 2006|
|Priority date||Dec 2, 2005|
|Also published as||CN101033984A, CN101033984B, DE602005027713D1, EP1793497A1, EP1793497B1, US20070152682|
|Publication number||11565454, 565454, US 7595648 B2, US 7595648B2, US-B2-7595648, US7595648 B2, US7595648B2|
|Inventors||Tommaso Ungaretti, Ernesto Lasalandra|
|Original Assignee||Stmicroelectronics S.R.L.|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (17), Referenced by (43), Classifications (7), Legal Events (3)|
|External Links: USPTO, USPTO Assignment, Espacenet|
1 Field of the Invention
The present invention relates to a device and to a method for reading a capacitive sensor, in particular of a micro-electromechanical type.
2 Description of the Related Art
As is known, the use of capacitive sensors is continuously spreading to numerous applications, in which the reduction of consumption is a fundamental target. For example, capacitive inertial micro-electromechanical-system (MEMS) sensors of a differential type are increasingly frequently used in a wide range of portable electronic devices, such as cell phones, palm-top computers, digital camcorders and cameras, and the like, which are supplied autonomously by batteries. Clearly, in cases of this sort the reduction of the consumption is indispensable for increasing the autonomy of the device.
In order to minimize the power absorption, very frequently traditional continuous-time read circuits for capacitive sensors have been replaced by switched-capacitor (SC) read circuits, which are much more suitable for operating with low supply voltages and an extremely low current consumption. In a parallel manner, reading techniques have been developed for optimizing the reading precision and sensitivity. For example, the so-called “correlated-double-sampling” (CDS) technique enables effective elimination of the disturbance caused by possible offsets and low-frequency noise (1/f noise, or flicker noise) of the electronics used (typically, a charge-voltage converter including a charge amplifier).
By way of example,
A read circuit 3 is associated to the inertial sensor 1 and comprises a signal source 4, a charge-voltage converter 5, and a canceling stage 7.
The signal source 4 is connected to the driving terminal 1 c of the inertial sensor 1 and supplies a step read voltage VRD.
The charge-voltage converter 5 includes a fully differential switched-capacitor charge amplifier 10, having a first integration capacitor 11 a connected between a first input and a first output and a second integration capacitor 11 b connected between a second input and a second output. Furthermore, the first input and the second input of the charge amplifier 10 are connected to the first sense terminal 1 a and to the second sense terminal 1 b of the inertial sensor 1, respectively.
The canceling stage 7 comprises a first hold capacitor 12 a and a second hold capacitor 12 b, respectively connected in series to the first output and to the second output of the charge amplifier 10. Furthermore, terminals of the first hold capacitor 12 a and of the second hold capacitor 12 b form a first output 3 a and, respectively, a second output 3 b of the read circuit 3.
In a first step, or reset step, the signal source 4 (herein illustrated with a dashed line) sends the driving terminal 1 c of the inertial sensor 1 to a ground value. The first input and the second input of the charge amplifier 10 are instead brought to a reference line 15, which supplies a constant reference voltage VREF, whereas the first output and the second output are short-circuited. For this purpose, first reset switches 16 a, 16 b, connected between the reference line 15 and a respective input of the charge amplifier 10, and a second reset switch 16 c, arranged between the outputs of the charge amplifier 10 itself, go into a closed condition.
In a second step, or offset-canceling step, the first reset switches 16 a, 16 b and the second reset switch 16 c are opened condition, while the driving terminal 1 c of the inertial sensor 1 is still kept at the ground voltage. Furthermore, a first canceling switch 18 a, connected between the reference line 15 and the first output 3 a, and a second canceling switch 18 b, connected between the reference line 15 and the second output 3 b, are closed. In this way, any possible disturbance, such as 1/f noise, and a possible offset introduced by the charge amplifier 10 cause a canceling voltage VC between the outputs by the charge amplifier 10 itself. The canceling voltage VC is in practice stored in the first hold capacitor 12 a and in the second hold capacitor 12 b.
Finally, a third step or sensing step is executed, in which the first reset switches 16 a, 16 b, the second reset switch 16 c, and the canceling switches 18 a, 18 b are opened, and the signal source (illustrated with a solid line) supplies the step read voltage VRD to the driving terminal 1 c of the inertial sensor 1. Through the first sense capacitor 2 a and the second sense capacitor 2 b, a first sense charge QA and, respectively, a second sense charge QB, correlated to the capacitive unbalancing of the inertial sensor 1, are supplied to the inputs of the charge amplifier 10 and converted into a rough output voltage VOR, which includes the contributions of noise and of offset due to the charge amplifier 10. The canceling stage 7 subtracts the canceling voltage VC, stored by the first hold capacitor 12 a and the second hold capacitor 12 b. In response to the step of the read voltage VRD, then, the read circuit 3 supplies a corrected output voltage VOC, which substantially is not affected by the contribution of the low-frequency noise and of the offsets introduced by the electronics.
Albeit effective for eliminating the disturbance described, the CDS technique does not, however, enable suppression of the low-frequency disturbance generated upstream of the charge amplifier 10. Said disturbance can have different origins, but, basically, takes the form of a differential disturbance current ID supplied in parallel to the currents due to the injection of charge (QA, QB) performed by the inertial sensor 1 (the disturbance is represented schematically by a current generator 20 in
One embodiment of the present invention provides a device and a method for reading a capacitive sensor, which will enable the drawbacks described above to be overcome.
For a better understanding of the invention, there is now described an embodiment thereof, provided purely by way of non-limiting example and with reference to the attached plate of drawings, wherein:
The inertial sensor 101, in itself known, comprises a movable body (not illustrated), which is movable with respect to a fixed body (which is not illustrated either) and is capacitively coupled thereto so as to form in practice a first sense capacitor 102 a, and a second sense capacitor 102 b (see for this item
The read device 103 is of the discrete-time type and comprises a signal source 104, a modulator stage 105, a sense circuit 107, a demodulator stage 110, and a low-pass filter 112. Furthermore, a phase-generator stage 113 generates a plurality of timing signals S1-S6, necessary for coordinating operation of the different components of the read device 103. In greater detail, the phase-generator stage 113 comprises an oscillator 113 a and a plurality of timing modules C1-C6, controlled by the oscillator 113 a, each of which generates a respective timing signal S1-S6.
The signal source 104 supplies a step read voltage VRD, preferably of an amplitude equal to the maximum dynamics available, which, in the embodiment described, is equal to the supply voltage VDD (for example, 3 V).
The modulator stage 105 is arranged between the signal source 104 and the inertial sensor 101 and modulates the read voltage VRD, multiplying it at each read cycle alternately by +1 and −1 (for example, the reading frequency is comprised between 10 and 100 kHz). In practice, a square-wave modulated read voltage VRDM is present on the output of the modulator stage 105 and is supplied to the inertial sensor 101. Reading of the inertial sensor 101 is performed on each edge, both the leading edge and the trailing edge, of the modulated read voltage VRDM.
The sense circuit 107 comprises a charge-voltage converter 108 and a canceling stage 109.
The charge-voltage converter 108, of the switched-capacitor type, is set downstream of the inertial sensor 101. In particular, the charge-voltage converter 108 receives a first sense charge QA and a second sense charge QB, which are supplied by the inertial sensor 101 in response to the modulated read voltage VRDM and are correlated to the differential variations of capacitance ΔCS of the first sense capacitor 102 a and of the second sense capacitor 102 b.
The canceling stage 109 is connected in series to outputs of the charge-voltage converter 108 and is used for canceling the offset and the low-frequency noise introduced by the charge-voltage converter 108 itself. A modulated output voltage VOM is present between the outputs of the canceling stage 109.
The demodulator 110 is cascaded to the canceling stage 109, for receiving the modulated output voltage VOM, which is again multiplied alternately by +1 and −1 at each read cycle and converted into a demodulated output voltage VOD.
Finally, the low-pass filter 112 receives the demodulated output voltage VOD, eliminates the high-frequency components, and supplies a corrected output voltage VOC.
The read device 103 enables suppression also of the disturbance due to the inertial sensor 101, in addition to the low-voltage noise and to the offset introduced by the charge-voltage converter 108. The modulation, in fact, enables separation of the harmonic content of the read voltage VRD and of the variations of capacitance ΔCS from that of the disturbance caused by the inertial sensor 101. In particular, the modulation translates at high frequency the spectrum of the variations of capacitance ΔCS that are linked to the quantities detected by the inertial sensor 101 (i.e., the useful signal,
The inertial sensor 101 is represented schematically by the first sense capacitor 102 a and the second sense capacitor 102 b, which have a common terminal forming the driving input 101 c and respective second terminals forming a first sense terminal 101 a and a second sense terminal 101 b.
The signal source 104 and the modulator stage 105 are formed by a single circuit, hereinafter referred to as modulated source 106, comprising a ground line 115, set at a ground voltage VGND (0 V), a supply line 116, supplying a supply voltage VDD, and a first read switch 118 a and a second read switch 118 b, respectively controlled by a first timing signal S1 and a second timing signal S2 (the first timing signal S1 and the second timing signal S2 are generated by a first timer module C1 and by a second timer module C2, respectively). More precisely, the first read switch 118 a and the second read switch 118 b have a terminal in common connected to the driving terminal 101 c of the inertial sensor 101; moreover, the first read switch 118 a is connected to the ground line 115, whereas the second read switch 118 b is connected to the supply line 116.
The charge-voltage converter 108 comprises a fully differential switched-capacitor charge amplifier 120, having a first integration capacitor 121 a connected between a first input and a first (non-inverting) output and a second integration capacitor 121 b connected between a second input and a second (inverting) output. The first input and the second input of the charge amplifier 120 are connected to the first sense terminal 101 a and, respectively, to the second sense terminal 101 b of the inertial sensor 101. Furthermore, the first input and the second input of the charge amplifier 120 can be selectively connected to a reference line 123, supplying a constant reference voltage VREF by a first reset switch 125 a and, respectively, a second reset switch 125 b. A third reset switch is connected between the outputs of the charge amplifier 120. All the reset switches 125 a, 125 b, 125 c are controlled by one and the same third timing signal S3, generated by a third timer module C3 of the phase-generator stage 113.
The canceling stage 109 comprises a first hold capacitor 122 a and a second hold capacitor 122 b, respectively connected in series to the first output and to the second output of the charge amplifier 109.
The demodulator stage 110 has a first input 110 a and a second input 110 b and a first output 110 c and a second output 110 d. A first direct-connection switch 127 a and a second direct-connection switch 127 b are connected between the first input 110 a and the first output 110 c and between the second input 110 b and the second output 110 d, respectively, and are controlled by a same fourth timing signal S4 generated by a fourth timer module C4. A first crossed-connection switch 127 c and a second crossed-connection switch 127 d are connected between the first input 110 a and the second output 110 d and between the second input 110 b and the first output 110 c, respectively, and are controlled by a same fifth timing signal S5 generated by a fifth timer module C5. In particular, the direct-connection switches 127 a, 127 b and the crossed-connection switches 127 c, 127 d are controlled so that the connection between the inputs 110 a, 110 b and the outputs 110 c, 110 d of the demodulator stage 110 is reversed in consecutive read cycles. In other words, if in a read cycle RDK the first input 110 a is connected to the first output 110 c and the second input 110 b is connected to the second output 110 d, in the immediately ensuing read cycle RDK+1 the first input 110 a is connected to the second output 110 d, and the second input 110 b is connected to the first output 110 c. In consecutive read cycles RDK, RDK+1, then, the sign of the demodulated output voltage VOD is once equal (multiplication by +1) and once opposite (multiplication by −1) to the sign of the modulated output voltage VOM.
The first output 110 c and the second output 110 d of the demodulator stage 110 are moreover selectively connectable to the reference line 123 by a first canceling switch 128 a and, respectively, a second canceling switch 128 b, which are controlled by a same sixth timing signal S6, generated by a sixth timer module C6 of the phase-generator stage 113.
Finally, in the embodiment described herein, the low-pass filter 112 is of a discrete-time analog type and is substantially configured so as to calculate the arithmetic mean of pairs (or in any case an even number) of successive samples of the demodulated output voltage VOD. The corrected output voltage VOC is hence proportional to said arithmetic mean.
Operation of the acceleration-detection apparatus 100 will be explained hereinafter with reference also to
Throughout the read cycle RDK, the direct-connection switches 127 a, 127 b of the demodulator stage 110 are in closed, and the crossed-connection switches 127 c, 127 d (not illustrated) are open.
In the reset step of the read cycle RDK, initial levels of d.c. voltage are fixed for the sense circuit 107. In detail, the switched source 106 keeps the driving terminal 101 c of the inertial sensor 101 at ground (
In the subsequent canceling step (
A sensing step is then executed (
The read cycle RDK comes to an end, and the next read cycle RDK+1 is executed. In this case, the modulated source 106 keeps the driving terminal 101 c of the inertial sensor 101 at the supply voltage VDD during the steps of reset and canceling and supplies a negative-voltage step in the sensing step.
More in detail, in the reset step (
In the subsequent canceling step (
In the sensing step (
In practice, owing to the action of the modulated source 106 and of the demodulator 110, the effect of the disturbance current ID on the demodulated output voltage VOD has opposite sign in any two consecutive read cycles. Since the low-pass filter 112 executes an arithmetic mean of an even number of samples of the demodulated output voltage VOD, the effect of the disturbance current ID and hence of all the possible dispersions upstream of the charge amplifier 120 is substantially eliminated. The modulated source 106 and the demodulator stage are controlled by timing signals (first, second, fourth, and fifth timing signal S1, S2, S4, S5), which have a period twice that of the ones used for reset and canceling (third and sixth timing signal S3, S6). In this way, it is possible to invert in sign the correlation between the modulated read voltage VRDM and the demodulated output voltage VOD in consecutive read cycles, which have the same logic sequence of steps.
The device described herein exploits in a particularly advantageous way the characteristics of discrete-time analog circuits, in particular switched-capacitor ones, for carrying out modulation of the read voltage and demodulation of the output voltage. The modulated source 106 and the demodulator stage 110 are provided by extremely simple and efficient circuit solutions, since, in practice, switches appropriately controlled by the phase-generator stage 113 are used. The solution described affords considerable advantages also from the standpoints of the speed and of power consumption.
In accordance with a third embodiment of the invention (illustrated in
The output of the sigma-delta converter 312 generates a demodulated numeric acceleration signal A starting from the demodulated output voltage VOD, supplied by the demodulator stage 310. In the demodulated numeric acceleration signal A, there are still present the effects of the disturbance current ID due to the dispersions upstream of the charge-voltage converter 308.
The low-pass filter 312 receives the demodulated numeric acceleration signal A and calculates a mean value thereof on an even number of samples, for generating a corrected numeric acceleration signal AC.
As illustrated schematically in
Finally, it is evident that modifications and variations may be made to the device and to the read method described, without departing from the scope of the present invention, as defined in the annexed claims. In particular, the signal source and the modulator stage can be provided by separate circuits.
|Cited Patent||Filing date||Publication date||Applicant||Title|
|US3879660 *||Oct 24, 1973||Apr 22, 1975||John S Piso||Capacitive measuring system|
|US4208625 *||Apr 20, 1978||Jun 17, 1980||Micro Sensors, Inc.||Capacitive measuring system with automatic calibration|
|US5343156||Apr 15, 1992||Aug 30, 1994||Nat Semiconductor Corp||IC for producing an output voltage related to fuel composition in a capacitive fuel sensor|
|US5637798||Nov 3, 1995||Jun 10, 1997||Robert Bosch Gmbh||Circuit arrangement for evaluating an acceleration sensor signal|
|US5801307 *||Jan 10, 1997||Sep 1, 1998||Netzer; Yishay||Differential windshield capacitive moisture sensors|
|US5808198||May 19, 1997||Sep 15, 1998||The Charles Stark Draper Laboratory, Inc.||RF balanced capacitive vibration sensor system|
|US6388453 *||Jan 25, 1999||May 14, 2002||Bryan D. Greer||Swept-frequency dielectric moisture and density sensor|
|US6470748 *||Oct 4, 2000||Oct 29, 2002||Analog Devices, Inc.||Feedback mechanism for rate gyroscopes|
|US6621334 *||Dec 30, 2002||Sep 16, 2003||Infineon Technologies Ag||Frequency-compensated, multistage amplifier configuration and method for operating a frequency-compensated amplifier configuration|
|US6714025 *||Feb 19, 2002||Mar 30, 2004||Robert Bosch Gmbh||Method and apparatus for synchronous demodulation of multiply modulated signals|
|US6856144||Feb 20, 2002||Feb 15, 2005||Stmicroelectronics S.R.L.||Method and circuit for detecting movements through micro-electric-mechanical sensors, compensating parasitic capacitances and spurious movements|
|US7134336 *||Mar 10, 2005||Nov 14, 2006||Denso Corporation||Vibration type angular velocity sensor|
|US20030048036 *||Aug 31, 2001||Mar 13, 2003||Lemkin Mark Alan||MEMS comb-finger actuator|
|US20030098699 *||Nov 16, 2001||May 29, 2003||Lemkin Mark A.||Sense interface system with velocity feed-through rejection|
|US20050218911 *||Apr 6, 2004||Oct 6, 2005||Denison Timothy J||Linearity enhancement for capacitive sensors|
|JPH10111207A||Title not available|
|WO2005068959A2 *||Jan 5, 2005||Jul 28, 2005||Case Western Reserve University||High performance integrated electronic for mems capacitive strain sensors|
|Citing Patent||Filing date||Publication date||Applicant||Title|
|US7969167 *||Jan 28, 2009||Jun 28, 2011||Freescale Semiconductor, Inc.||Capacitance-to-voltage interface circuit with shared capacitor bank for offsetting and analog-to-digital conversion|
|US8219331 *||May 13, 2009||Jul 10, 2012||Texas Instruments Deutschland Gmbh||Electronic device and method for evaluating a variable capacitance|
|US8265769||Jan 31, 2007||Sep 11, 2012||Medtronic, Inc.||Chopper-stabilized instrumentation amplifier for wireless telemetry|
|US8347715 *||Dec 10, 2009||Jan 8, 2013||Stmicroelectronics S.R.L.||Variable capacitance electronic device and microelectromechanical device incorporating such electronic device|
|US8354881||Dec 2, 2010||Jan 15, 2013||Medtronic, Inc.||Chopper-stabilized instrumentation amplifier|
|US8478402||Oct 31, 2008||Jul 2, 2013||Medtronic, Inc.||Determining intercardiac impedance|
|US8667842||Dec 6, 2012||Mar 11, 2014||Stmicroelectronics S.R.L.||Variable capacitance electronic device and microelectromechanical device incorporating such electronic device|
|US8714012 *||Feb 15, 2011||May 6, 2014||Stmicroelectronics S.R.L.||Microelectromechanical gyroscope with inversion of actuation forces, and method for actuating a microelectromechanical gyroscope|
|US8742964 *||Jan 16, 2013||Jun 3, 2014||Fairchild Semiconductor Corporation||Noise reduction method with chopping for a merged MEMS accelerometer sensor|
|US8781595||Apr 30, 2007||Jul 15, 2014||Medtronic, Inc.||Chopper mixer telemetry circuit|
|US8813564||Jan 31, 2013||Aug 26, 2014||Fairchild Semiconductor Corporation||MEMS multi-axis gyroscope with central suspension and gimbal structure|
|US8978475||Feb 1, 2012||Mar 17, 2015||Fairchild Semiconductor Corporation||MEMS proof mass with split z-axis portions|
|US8981834 *||Dec 19, 2013||Mar 17, 2015||Stmicroelectronics S.R.L.||Circuit and method for dynamic offset compensation in a MEMS sensor device|
|US9006846||Sep 20, 2011||Apr 14, 2015||Fairchild Semiconductor Corporation||Through silicon via with reduced shunt capacitance|
|US9062972||Jan 31, 2012||Jun 23, 2015||Fairchild Semiconductor Corporation||MEMS multi-axis accelerometer electrode structure|
|US9069006||Feb 12, 2013||Jun 30, 2015||Fairchild Semiconductor Corporation||Self test of MEMS gyroscope with ASICs integrated capacitors|
|US9091713 *||Nov 6, 2012||Jul 28, 2015||Stmicroelectronics (Rousset) Sas||Method and device for characterizing or measuring a capacitance|
|US9094027||Apr 11, 2013||Jul 28, 2015||Fairchild Semiconductor Corporation||Micro-electro-mechanical-system (MEMS) driver|
|US9095072||Sep 16, 2011||Jul 28, 2015||Fairchild Semiconductor Corporation||Multi-die MEMS package|
|US9156673||Sep 18, 2011||Oct 13, 2015||Fairchild Semiconductor Corporation||Packaging to reduce stress on microelectromechanical systems|
|US9197173 *||Oct 14, 2009||Nov 24, 2015||Medtronic, Inc.||Chopper-stabilized instrumentation amplifier for impedance measurement|
|US9217805||Oct 1, 2010||Dec 22, 2015||Westerngeco L.L.C.||Monitoring the quality of particle motion data during a seismic acquisition|
|US9246018||Sep 18, 2011||Jan 26, 2016||Fairchild Semiconductor Corporation||Micromachined monolithic 3-axis gyroscope with single drive|
|US9248288||Jan 14, 2013||Feb 2, 2016||Medtronic, Inc.||Patient directed therapy control|
|US9278845||Jan 31, 2013||Mar 8, 2016||Fairchild Semiconductor Corporation||MEMS multi-axis gyroscope Z-axis electrode structure|
|US9278846||Sep 18, 2011||Mar 8, 2016||Fairchild Semiconductor Corporation||Micromachined monolithic 6-axis inertial sensor|
|US9352961||Sep 16, 2011||May 31, 2016||Fairchild Semiconductor Corporation||Flexure bearing to reduce quadrature for resonating micromachined devices|
|US9425328||Sep 11, 2013||Aug 23, 2016||Fairchild Semiconductor Corporation||Through silicon via including multi-material fill|
|US9444404||Apr 5, 2013||Sep 13, 2016||Fairchild Semiconductor Corporation||MEMS device front-end charge amplifier|
|US9449501||Jun 4, 2014||Sep 20, 2016||Medtronics, Inc.||Chopper mixer telemetry circuit|
|US9455354||Sep 18, 2011||Sep 27, 2016||Fairchild Semiconductor Corporation||Micromachined 3-axis accelerometer with a single proof-mass|
|US9488693||Jan 16, 2013||Nov 8, 2016||Fairchild Semiconductor Corporation||Self test of MEMS accelerometer with ASICS integrated capacitors|
|US9506964||Nov 6, 2012||Nov 29, 2016||Stmicroelectronics Rousset Sas||Method and device for characterizing or measuring a floating capacitance|
|US20090295460 *||May 13, 2009||Dec 3, 2009||Texas Instruments Deutschland Gmbh||Electronic device and method for evaluating a variable capacitance|
|US20100033240 *||Oct 14, 2009||Feb 11, 2010||Medtronic, Inc.||Chopper-stabilized instrumentation amplifier for impedance measurement|
|US20100114223 *||Oct 31, 2008||May 6, 2010||Wahlstrand John D||Determining intercardiac impedance|
|US20100149721 *||Dec 10, 2009||Jun 17, 2010||Stmicroelectronics S.R.L.||Variable capacitance electronic device and microelectromechanical device incorporating such electronic device|
|US20100188107 *||Jan 28, 2009||Jul 29, 2010||Freescale Semiconductor, Inc.||Capacitance-to-voltage interface circuit with shared capacitor bank for offsetting and analog-to-digital conversion|
|US20100327887 *||Aug 31, 2010||Dec 30, 2010||Medtronic, Inc.||Chopper-stabilized instrumentation amplifier for impedance measurement|
|US20110197675 *||Feb 15, 2011||Aug 18, 2011||Stmicroelectronics S.R.L.||Microelectromechanical gyroscope with inversion of actuation forces, and method for actuating a microelectromechanical gyroscope|
|US20130057298 *||Nov 6, 2012||Mar 7, 2013||Stmicroelectronics (Rousset) Sas||Method and device for characterizing or measuring a capacitance|
|US20130265183 *||Jan 16, 2013||Oct 10, 2013||Fairchild Semiconductor Corporation||Noise reduction method with chopping for a merged mems accelerometer sensor|
|US20140176231 *||Dec 19, 2013||Jun 26, 2014||Stmicroelectronics S.R.L.||Circuit and method for dynamic offset compensation in a mems sensor device|
|U.S. Classification||324/678, 330/9, 324/676|
|International Classification||G01R27/26, H03F1/02|
|Mar 12, 2007||AS||Assignment|
Owner name: STMICROELECTRONICS S.R.L., ITALY
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:UNGARETTI, TOMMASO;LASALANDRA, ERNESTO;REEL/FRAME:018996/0880
Effective date: 20070212
|Feb 26, 2013||FPAY||Fee payment|
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