|Publication number||US7602835 B1|
|Application number||US 10/915,777|
|Publication date||Oct 13, 2009|
|Filing date||Aug 10, 2004|
|Priority date||Aug 10, 2004|
|Publication number||10915777, 915777, US 7602835 B1, US 7602835B1, US-B1-7602835, US7602835 B1, US7602835B1|
|Inventors||Samuel C. Kingston, Johnny M. Harris, Thomas R. Giallorenzi, Dan M. Griffin, David W. Thorson|
|Original Assignee||L-3 Communications Corporation|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (7), Non-Patent Citations (1), Referenced by (5), Classifications (25), Legal Events (2)|
|External Links: USPTO, USPTO Assignment, Espacenet|
The present invention relates to spread spectrum communication systems, particularly, a long code sequence and a method for building and deriving such a code that is less computationally expensive than known CDMA long codes.
In digital spread spectrum (DSS) communication, a wide band carrier signal is modulated by a narrow band message signal. The wide-band carrier is typically generated by modulating a single frequency carrier using a pseudo-random noise (P/N) code sequence. The data rate at which a message is communicated is usually much lower than the P/N code symbol or “chip” rate. The ability of DSS to suppress interference is proportional to a ratio of the chip rate to data rate. In many applications, there are thousands of code chips per data bit.
At the receiver, a carrier replica is generated by reducing the DSS signal to baseband and multiplying it with a locally generated replica of the original narrow-band carrier using a local oscillator. If the frequency and phase of the carrier replica is the same as that of the received original narrow-band carrier, then the multiplier output signal will be the product of the bipolar P/N code and intended message. The P/N code is removed by multiplying the wide-band data stream with the locally generated replica of the P/N code that is time aligned with the received P/N code. This is the de-spreading process.
Generating the carrier replica with proper carrier frequency and phase and generating the P/N code replica at the proper rate and time offset is a complex problem. In many DSS communication systems, the necessary carrier frequency, carrier phase, and P/N code offset are not known a priori at the receiver, which tries different values until a large signal is observed at the data-filter output. This is termed the search or acquisition process, and a DSS signal is said to be acquired when the proper frequency, phase, and code offset have been determined. A receiver selects and detects a particular transmitted signal by choosing the appropriate P/N code and performing the acquisition search. In some cases the acquisition search must include examination of different P/N codes from a known list when the transmitting node is not known, as is the likely scenario in
The above constraints are more pronounced in a secure environment such as that depicted in
The air interface should consist of a flexible and symmetric full-duplex or half-duplex link. The transmitting node or hailing node is that node that sends a discovery burst, essentially a message inquiring as to the presence of receiving nodes. Receiving nodes are the nodes that listen for that discovery burst. The receiving nodes are therefore target nodes, which may already have formed a network. These receiving nodes may become transmitting nodes when they send an acknowledgement back to the initiating new node. In this way, a new node that flies into range of an established network will transmit burst discovery messages on that transmitting node's transmit link. When a receiving node in the established network hears the discovery message on its receive link, it will respond via its transmit link which is the hailing node's receiving link. Subsequent handshaking can then be performed via the two node's transmit and receive links to bring the initiating new node into the network. The transmitting and receiving links may occupy separate time slots in a time division duplex (TDD) system, or may be separate frequency bands in a frequency division duplex (FDD) system.
An exemplary but non-limiting environment in which node discovery may be important is illustrated in perspective view at
The command node controls access to the network, identifying nodes and answering discovery bursts with a particularly long P/N code for data exchange to ensure networked members may communicate securely. In
Considering the issues apparent in light of
As to item (b), discovery and verification should preferably be done when the hailing node is beyond the range of direct communications with the network. In secure communications without satellite links where range is limited, hailing nodes are unaware of the location of nodes within the network and vice versa, so omni-directional antennas are generally used for discovery. This is because the reduced range of communications using omni-directional antennas is more than offset by having to scan various quadrants with a directional antenna for entering nodes that may come from any direction. Once within a network, nodes use directional antennas for increased range and reduced LPI. If all discovery handshaking can occur before the hailing node is in directional antenna range, then all synchronization can be done ahead of time. This synchronization will include the synchronization of chip clocks and carrier local oscillators in the presence of potentially large Doppler and reference clock offsets, antenna pointing, and the acquisition of very long PN codes.
To facilitate LPI/LPD of item (c), the waveform must support very fast acquisition; the receiving node must be capable of reliably extracting data from the transmitting node within a small fraction of a second from when the transmission begins, so that bursts are short and covert. The initial bursts should be at the lowest possible power level and spread over a band that is wider than the signal bandwidth required. This fast acquisition should occur regardless of whether a node is stationary or mobile at rates of up to several thousand km/hr in any direction, and when relative velocity between nodes is neither known nor estimable before discovery.
Item (d) is that any node is capable of discovering any other node. This implies, for example, that if an FDD approach is adopted, wherein nodes that transmit in frequency band A and receive in frequency band B, can only be heard by nodes that receive in band A, then the nodes must be capable of switching their FDD polarity. In other words, if a hailing node sends discovery bursts in band A and receives no response, then it must be capable of switching to band B to search for nodes listening in that band. Correspondingly, a TDD system imposes the trivial constraint that nodes be capable of switching which time slot in which they transmit
Item (e) is desirable for discovery to avoid a need for a complicated and cumbersome clock availability requirement across the system. This complicates discovery in that discovery bursts may be sent at any time rather than in designated slots (thus enhancing LPD, LPI, and discovery range for mobile nodes closing on one another), and eliminates the need for hailing nodes to be synchronized with precision.
What is needed in the art is a receiver that can quickly acquire a node discovery signal or a reply to one (e.g., determine the PN code, phase, offset, and frequency). One particularly flexible code is described in a paper by Yingwei Yao and H. Vincent Poor, entitled A Two-Layer Spreading Code Scheme for Dual Rate CDMA Systems, IEEE T
In accordance with one aspect, the present invention is a correlator to despread a composite long code defining a length of at least N*M, where N and M are integers each greater than one. The correlator may be considered as having a first and second stage. The first stage includes a first plurality of at least N−1 delay elements arranged in series with one another, a first plurality of at least N filter taps, and a first despread adder. Each of the first stage filter taps has an input that is coupled to either an input or an output of one of the N first stage delay elements. The first despread adder has an input coupled to an output of each of the at least N filter taps.
The second stage includes a second plurality of at least M−1 delay elements arranged in series with one another, a second plurality of at least M filter taps, and a second despread adder. Each of the second stage filter taps has an input that is coupled to either an input or an output of one of the M−1 second stage delay elements. The second despread adder has an input that is coupled to an output of each of the at least M filter taps. The first and second stages are related in that an output of the second despread adder is coupled to an input of the series that is the first plurality of at least N delay elements.
As will be described, preferably the delay elements impose a common time delay within a stage but different time delays between circuit stages so they run at different rates, while each delay element within a stage imposes a common time delay. Where the composite code is constructed from different constituent sub-codes, each stage decorrelates one of the constituent sub-codes. Additional stages may be imposed similar to the second stage, and arranged so that, for example, a third despread adder has an output that is coupled to an input of the second plurality of delay elements. This extension may be continued to fourth, fifth, etc. circuit stages.
The present invention is in another aspect a method of decorrelating a P/N code, where the code does not repeat for at least N*M elements, N and M again being integers greater than one. In accordance with the method, each of a set of M first constituent code elements are filtered at a first rate from a received input signal. Each of a set of N second constituent code elements are filtered at a second rate from a summation of the filtered set of first constituent code elements. Each of the second set of N constituent code elements are summed, and the summed set of N second constituent code elements is input into a register and stored. The above filtering at a first and second rate, summing, inputting and storing are repeating in sequence for N*M time intervals until N*M separate summed sets of the N second constituent code elements are stored. The P/N code is decorrelated when the N*M separate summed sets exhibit an energy peak. As with the correlator apparatus, the inventive method may be extended for more than two layers similar to the correlator being extended to more than two circuit stages.
In yet another aspect, the present invention is a mobile platform for transmitting and receiving a discovery burst. The mobile platform includes at least one antenna that may be an omni-directional antenna used for the discovery burst. It also includes a transmitter for transmitting a discovery burst, where the discovery burst has a preamble and a payload. The preamble is spread using a composite P/N code, which has a plurality of serially arranged code segments. Each code segment includes an nth element of a first constituent code operated using an exclusive OR operator with each of M elements of a second constituent code. That the code segments are serially arranged does not necessitate that they follow one another with no intervening elements.
The mobile platform further includes a receiver for receiving the discovery burst, when the discovery burst is transmitted by another platform. The receiver has a correlator which itself includes a first and second circuit stage. The first circuit stage has at least N−1 serially disposed delay elements, at least N filter taps, and a first despread adder. Each of the N filter taps has an input that is coupled to one of an input and an output of one of the N−1 delay elements. The first despread adder sums the output of the N filter taps. The second circuit stage includes at least M−1 serially disposed delay elements, at least M filter taps, and a second despread adder. Each of the M filter taps has an input that is coupled to one of an input and an output of one of the M−1 delay elements. The second despread adder sums the output of the M filter taps, and further has an output that is coupled to an input of the serially arranged N delay elements. Each of N and M are integers at least equal to two.
These and other features, aspects, and advantages of embodiments of the present invention will become apparent with reference to the following description in conjunction with the accompanying drawings. It is to be understood, however, that the drawings are designed solely for the purposes of illustration and not as a definition of the limits of the invention.
Where a discovery burst announcing the presence of the hailing node to a network (whose existence and location may be unknown to the hailing node at the time it transmits the discovery burst), it is anticipated that the composite codes described herein will be used at least in a preamble of that discovery burst. Preferably, a subsequent payload of the discovery burst will be spread using a much longer P/N code than the preamble. In this instance, the preamble may be used to resolve phase and/or timing for the payload P/N code, as well as the timing point in any encryption sequence. In such a scenario, it is incumbent that the phase be resolved very quickly, on the order of about 600 msec for a preamble of 500 symbols within a discovery burst of about 1200 symbols and a chip rate of about 10 Mcps. Prior art parallel correlators are not seen as capable of achieving this metric within the size constraints deemed necessary for implementation in the smaller nodes of
The present invention uses a plurality of constituent codes or sub-codes to construct a composite long code such as the pseudorandom number (PN) codes used in a CDMA communication system. Even if there were no Doppler shifts of the carrier and chip frequencies and perfect clocks were used between hailing 34 and command 24 nodes of
The multi-rate composite codes according to the present invention may be constructed from two to any number of constituent codes. For purposes of this description and the claims, a constituent code is represented by a capital letter A, B, C, etc., and elements of the codes are represented by lower case letters with subscripts, such as elements a1, a2, a3, . . . aN for code A that has a total of N elements. A lower case letter in the subscript indicates a variable number, such as an represents an nth element of the code A where n varies from 1 to N. An upper case letter in the subscript indicates the final element of the code, such as aN represents the Nth element of the code A that has N elements.
By way of example, consider two constituent codes A and B, wherein A is a first constituent code having N=ten elements and B is a second constituent code having M=ten elements. The composite code of the present invention need not be constructed from equal length constituent codes. The elements of the constituent codes may be any real or complex valued quantity, though in practice the elements are typically either +1 or −1. The resulting composite code will include at least 100 elements, and will exhibit ten code segments each bearing ten composite elements. Additional elements may be disposed between the code segments. The first code segment is obtained by operating the first element b1 of the second constituent code B with each element a1, a2, . . . aN of the first constituent code A using an exclusive OR operation. Label this code segment as Ab1. The second code segment is obtained by similarly operating the second element b2 of the second constituent code B with each element a1, a2, . . . aN of the first constituent code A to yield Ab2. This continues for each of the M elements of the second constituent code, yielding ten code segments each of length ten elements. The resulting composite code is then written as AB. The code segments Abm are arranged serially, and may have additional code elements disposed between the segments as is described in U.S. patent application Ser. No. 10/915,776 filed Aug. 10, 2004. The various codes and nomenclatures are depicted below.
1st constituent code A:
A = a1, a2, a3, . . . aN;
2nd constituent code B:
B = b1, b2, b3, . . . bM;
1st code segment:
Ab1 = a1 ⊕ b1, a2 ⊕ b1, a3 ⊕ b1, . . . aN ⊕ b1;
2nd code segment:
Ab2 = a1 ⊕ b2, a2 ⊕ b2, a3 ⊕ b2, . . . aN ⊕ b2;
AB = Ab1, Ab2, Ab3, . . . AbM
It is clear from the above that each code segment has the same length N, and the composite code has M code segments arranged seriatim for a total of N*M elements in the composite code (barring the addition of further elements between code segments). Because the above example constructs the composite code from two constituent codes, it will be termed a two-layer composite code. It is noted that the above code segments are constructed by operating code elements by an exclusive- or operation. That is valid for real-valued code elements, but a multiplication of elements may be required for imaginary code elements. For simplicity, this description generally presumes real code elements combined with an exclusive- or operation. Extension of these teachings to imaginary code elements follows logically from the above distinction.
Composite codes may be in two, three, four or more layers, constructed from two, three, four, and any number of constituent codes, respectively. For example, consider a third constituent code C having L elements c1, c2, c3, . . . cL combined with the constituent codes A and B above. The first code segment of the composite code would be each element of AB operated with an exclusive OR with the first element c1 and would be abbreviated ABc1, the second code segment would be each element of AB similarly operated with the second element c2 abbreviated ABc2, and so forth to yield L code segments each of length N*M, and the resultant three-layer composite code ABC would exhibit a length N*M*L, barring added code elements between the segments as noted above.
It is unnecessary that the constituent codes be orthogonal to one another. So long as the constituent codes A, B, C, etc. are non-repetitive in their lengths, the code segments will be non-repetitive in their lengths. This aspect of the composite codes is explored further in co-owned U.S. patent application Ser. No. 10/915,776.
Longer code sequences in a direct sequence CDMA system are advantageous in that they enable more users in a system. Longer codes additionally also offer lower probability of detection and intercept in the potentially hostile environment of
Additional inputs 35 to the correlator 23 may include tap weights, code offsets, frequencies and local-oscillator frequency error assumed by control 27. To insure detection of the signal, the RF/IF section 21 must generate both an in-phase (I) and a quadrature (Q) output signal (not shown in
A typical prior art correlator 40 is depicted in block diagram at
One advantage to the composite codes described herein is that the inventive correlator described herein is greatly simplified in hardware and computational burden.
For the example of constituent codes A and B being lengths N=100 and M=100 respectively, the hardware requirement is 200 filter taps, 200 memory locations for tap weights, 198 delay elements, a 200 length shift register, and two despread adders, as opposed to 10,000 filter taps, 10,000 tap weight memory locations, 9,999 unit delay elements, a 10,000-length register, and one despread adder for the prior art correlator 40 of
It is noted that the prior art correlator 40 employs all unit delay elements, whereas the correlator 52 of
Hardware can be further reduced in other embodiments of the inventive correlator, those other embodiments being based on how the composite code is constructed. Assume the received signal 57 uses a composite long code ABCD of length 10,000 constructed from four constituent codes, identified as A of length N=10, B of length M=10, C of length L=10, and D of length K=10. Assume also that the chip rate is 50 million chips per second, and the correlator runs at double the chip rate or 100 Mcps so that the 10,000 length composite long code can be resolved within 0.1 milliseconds. This is an adequate time for establishing contact between a hailing node 34 and a command node 24 for the environment of
An embodiment of a four-stage correlator 70 for the parameters in the above paragraph is presented in block diagram at
The received signal 71 of
The embodiment of
The constituent codes of the correlator of
It is noted that the drawings and description presented herein are illustrative of the invention and not exhaustive. For example, while only one correlator is shown in each of
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|U.S. Classification||375/142, 375/253, 375/350, 375/143, 375/135, 375/150, 375/145, 375/149, 375/316, 375/146, 375/147, 375/136, 375/137, 375/306, 375/326, 375/140, 375/152, 375/307, 375/134, 375/367|
|Cooperative Classification||H04B1/7095, H04B2201/70707, H04B2201/70703|
|Aug 10, 2004||AS||Assignment|
Owner name: L-3 COMMUNICATIONS CORPORATION, NEW YORK
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:KINGSTON, SAMUEL C.;HARRIS, JOHNNY M.;GIALLORENZI, THOMAS R.;AND OTHERS;REEL/FRAME:015682/0102;SIGNING DATES FROM 20040708 TO 20040721
|Mar 13, 2013||FPAY||Fee payment|
Year of fee payment: 4