Publication number | US7623572 B2 |

Publication type | Grant |

Application number | US 11/639,782 |

Publication date | Nov 24, 2009 |

Filing date | Dec 15, 2006 |

Priority date | Sep 21, 2006 |

Fee status | Paid |

Also published as | EP1903732A2, US20080075158, US20080075189 |

Publication number | 11639782, 639782, US 7623572 B2, US 7623572B2, US-B2-7623572, US7623572 B2, US7623572B2 |

Inventors | Junqiang Li |

Original Assignee | Broadcom Corporation |

Export Citation | BiBTeX, EndNote, RefMan |

Patent Citations (3), Non-Patent Citations (2), Referenced by (2), Classifications (6), Legal Events (3) | |

External Links: USPTO, USPTO Assignment, Espacenet | |

US 7623572 B2

Abstract

A Radio Frequency (RF) receiver includes a RF front end and a baseband processing module coupled to the RF front end that is operable to determine equalizer coefficients for a composite signal that includes a first information signal and a second information signal. First and second information signal channel estimates and channel powers based upon training symbols are determined. A composite signal power is determined. A noise variance of the composite signal is determined based upon the first information signal channel power, the second information signal channel power, and the composite signal power. A plurality of equalizer coefficients are determined based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal.

Claims(34)

1. A method for determining equalizer coefficients by a Radio Frequency (RF) receiver for a composite signal that includes a first information signal and a second information signal, the method comprising:

determining a first information signal channel estimate based upon training symbols of the first information signal;

determining a first information signal channel power based upon the first information signal channel estimate;

determining a second information signal channel estimate based upon training symbols of the second information signal;

determining a second information signal channel power based upon the second information signal channel estimate;

determining a composite signal power of the composite signal;

determining a noise variance of the composite signal based upon the first information signal channel power, the second information signal channel power, and the composite signal power; and

determining a plurality of equalizer coefficients based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal.

2. The method of claim 1 , wherein determining a noise variance of the composite signal based upon the first information signal channel power, the second information signal channel power, and the composite signal power comprises:

scaling the first information signal channel power and the second information signal channel power;

combining the scaled first information signal channel power and the second information signal channel power to produce a combined signal channel power;

subtracting the combined signal channel power from the composite signal power to produce a subtracted result;

averaging the subtracted result over at least one slot time to produce a slot time averaged subtracted result; and

producing the noise variance from the slot time averaged subtracted result and a pilot signal to interference ratio.

3. The method of claim 2 , further comprising gain scaling at least one of the combined signal channel power and the slot time averaged subtracted result.

4. The method of claim 2 , further comprising filtering the slot time averaged subtracted result.

5. The method of claim 1 , wherein determining a plurality of equalizer coefficients based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal comprises:

converting the first information signal channel estimate from the time domain to the frequency domain to produce a first information signal frequency domain channel estimate;

converting the second information signal time domain channel estimate from the time domain to the frequency domain to produce a second information signal frequency domain channel estimate;

converting the noise variance from the time domain to the frequency domain to produce a frequency domain noise variance;

producing first frequency domain equalizer coefficients and second frequency domain equalizer coefficients based upon the first information signal frequency domain channel estimate, the second information signal frequency domain channel estimate, and the frequency domain noise variance;

converting the first frequency domain equalizer coefficients to the time domain to produce first time domain equalizer coefficients; and

converting the second frequency domain equalizer coefficients to the time domain to produce second time domain equalizer coefficients.

6. The method of claim 1 , wherein:

the first information signal is a desired signal; and

the second information signal is an interfering signal.

7. The method of claim 1 , wherein the first information signal and the second information signal are Multiple Input Multiple Output (MIMO) information signals intended for the RF receiver.

8. The method of claim 1 , wherein determining a plurality of equalizer coefficients based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal comprises performing a Minimum Mean Squared Error (MMSE) algorithm.

9. The method of claim 1 , wherein the RF receiver supports wireless operations selected from the group consisting of cellular wireless communications, wireless metropolitan area network communications, wireless local area network communications, and wireless personal area network communications.

10. A method for determining equalizer coefficients by a Radio Frequency (RF) receiver for a composite signal that includes a first information signal, a second information signal, and a pilot signal, the method comprising:

equalizing the composite signal by a first equalizer to produce a first equalized information signal;

equalizing the composite signal by a second equalizer to produce a second equalized information signal;

determining a signal power of the composite signal;

determining pilot signal interference power;

determining pilot signal power;

determining composite signal interference power based upon the composite signal power;

determining a ratio of the composite signal interference power to the pilot signal interference power;

determining a noise variance of the composite signal based upon the ratio of the composite signal interference power to the pilot signal interference power; and

determining a plurality of equalizer coefficients based upon a first information signal channel estimate, a second information signal channel estimate, and the noise variance of the composite signal.

11. The method of claim 10 , wherein determining a noise variance of the composite signal based upon the ratio of the composite signal interference power to the pilot signal interference power comprises performing a table lookup based upon the ratio of the composite signal interference power to the pilot signal interference power.

12. The method of claim 10 , wherein the noise variance comprises:

a first noise variance corresponding to the first information signal; and

a second noise variance corresponding to the second information signal.

13. The method of claim 12 , further comprising separately gain scaling the first noise variance and the second noise variance.

14. The method of claim 10 , wherein determining a plurality of equalizer coefficients based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal comprises:

converting the first information signal channel estimate from the time domain to the frequency domain to produce a first information signal frequency domain channel estimate;

converting the second information signal time domain channel estimate from the time domain to the frequency domain to produce a second information signal frequency domain channel estimate;

converting the noise variance from the time domain to the frequency domain to produce a frequency domain noise variance;

producing first frequency domain equalizer coefficients and second frequency domain equalizer coefficients based upon the first information signal frequency domain channel estimate, the second information signal frequency domain channel estimate, and the frequency domain noise variance;

converting the first frequency domain equalizer coefficients to the time domain to produce first time domain equalizer coefficients; and

converting the second frequency domain equalizer coefficients to the time domain to produce second time domain equalizer coefficients.

15. The method of claim 10 , wherein:

the first information signal is a desired signal; and

the second information signal is an interfering signal.

16. The method of claim 10 , wherein the first information signal and the second information signal are Multiple Input Multiple Output (MIMO) information signals intended for the RF receiver.

17. The method of claim 10 , wherein determining a plurality of equalizer coefficients based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal comprises performing a Minimum Mean Squared Error (MMSE) algorithm.

18. The method of claim 10 , wherein the RF receiver supports wireless operations selected from the group consisting of cellular wireless communications, wireless metropolitan area network communications, wireless local area network communications, and wireless personal area network communications.

19. A Radio Frequency (RF) receiver operable to determine equalizer coefficients by a Radio Frequency (RF) receiver for a composite signal that includes a first information signal and a second information signal, the RF receiver comprising:

a RF front end; and

a baseband processing module coupled to the RF front end and operable to:

determine a first information signal channel estimate based upon training symbols of the first information signal;

determine a first information signal channel power based upon the first information signal channel estimate;

determine a second information signal channel estimate based upon training symbols of the second information signal;

determine a second information signal channel power based upon the second information signal channel estimate;

determine a composite signal power of the composite signal;

determine a noise variance of the composite signal based upon the first information signal channel power, the second information signal channel power, and the composite signal power; and

determine a plurality of equalizer coefficients based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal.

20. The RF receiver of claim 19 , wherein in determining a noise variance of the composite signal based upon the first information signal channel power, the second information signal channel power, and the composite signal power the baseband processing module is operable to:

scale the first information signal channel power and the second information signal channel power;

combine the scaled first information signal channel power and the second information signal channel power to produce a combined signal channel power;

subtract the combined signal channel power from the composite signal power to produce a subtracted result;

average the subtracted result over at least one slot time to produce a slot time averaged subtracted result; and

produce the noise variance from the slot time averaged subtracted result and a pilot signal to interference ratio.

21. The RF receiver of claim 20 , wherein the baseband processing module is further operable to gain scale at least one of the combined signal channel power and the slot time averaged subtracted result.

22. The RF receiver of claim 21 , wherein the baseband processing module is further operable to filter the slot time averaged subtracted result.

23. The RF receiver of claim 19 , wherein:

the first information signal is a desired signal; and

the second information signal is an interfering signal.

24. The RF receiver of claim 19 , wherein the first information signal and the second information signal are Multiple Input Multiple Output (MIMO) information signals intended for the RF receiver.

25. The RF receiver of claim 19 , wherein in determining a plurality of equalizer coefficients based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal, the baseband processing module is operable to perform a Minimum Mean Squared Error (MMSE) algorithm.

26. The RF receiver of claim 19 , wherein the RF receiver supports wireless operations selected from the group consisting of cellular wireless communications, wireless metropolitan area network communications, wireless local area network communications, and wireless personal area network communications.

27. A Radio Frequency (RF) receiver operable to determine equalizer coefficients by a Radio Frequency (RF) receiver for a composite signal that includes a first information signal and a second information signal, the RF receiver comprising:

a RF front end; and

a baseband processing module coupled to the RF front end and operable to:

equalize the composite signal by a first equalizer to produce an first equalized information signal;

equalize the composite signal by a second equalizer to produce an second equalized information signal;

determine a signal power of the composite signal;

determine pilot signal interference power;

determine pilot signal power;

determine composite signal interference power based upon the composite signal power;

determine a ratio of the composite signal interference power to the pilot signal interference power;

determine a noise variance of the composite signal based upon the ratio of the composite signal interference power to the pilot signal interference power; and

determine a plurality of equalizer coefficients based upon a first information signal channel estimate, a second information signal channel estimate, and the noise variance of the composite signal.

28. The RF receiver of claim 27 , wherein in determining a noise variance of the composite signal based upon the ratio of the composite signal interference power to the pilot signal interference power, the baseband processing module is operable to perform a table lookup based upon the ratio of the composite signal interference power to the pilot signal interference power.

29. The RF receiver of claim 27 , wherein the noise variance comprises:

a first noise variance corresponding to the first information signal; and

a second noise variance corresponding to the second information signal.

30. The RF receiver of claim 29 , wherein the baseband processing module is further operable separately gain scaling the first noise variance and the second noise variance.

31. The RF receiver of claim 29 , wherein:

the first information signal is a desired signal; and

the second information signal is an interfering signal.

32. The RF receiver of claim 29 , wherein the first information signal and the second information signal are Multiple Input Multiple Output (MIMO) information signals intended for the RF receiver.

33. The RF receiver of claim 29 , wherein in determining a plurality of equalizer coefficients based upon the first information signal channel estimate, the second information signal channel estimate, and the noise variance of the composite signal, the baseband processing module is operable to perform a Minimum Mean Squared Error (MMSE) algorithm.

34. The RF receiver of claim 29 , wherein the RF receiver supports wireless operations selected from the group consisting of cellular wireless communications, wireless metropolitan area network communications, wireless local area network communications, and wireless personal area network communications.

Description

The present application is a continuation-in-part of:

1. Utility Application Ser. No. 11/524,584 filed on Sep. 21, 2006, and entitled “FREQUENCY DOMAIN EQUALIZER FOR DUAL ANTENNA RADIO,” and

2. Utility Application Ser. No. 11/524,580 filed on Sep. 21, 2006, and entitled “FREQUENCY DOMAIN EQUALIZER WITH ONE DOMINATE INTERFERENCE CANCELLATION FOR DUAL ANTENNA RADIO”; and

3. Utility Application Ser. No. 11/593,911 filed on Nov. 7, 2006, and entitled “EQUALIZER COEFICIENT DETERMINATION IN THE FREQUENCY DOMAIN FOR MIMO/MISO RADIO,” all of which are incorporated herein in their entirety by reference for all purposes.

1. Technical Field

The present invention relates generally to wireless communication systems; and more particularly to the equalization of data communications by a wireless radio in a wireless communication system.

2. Related Art

Cellular wireless communication systems support wireless communication services in many populated areas of the world. Cellular wireless communication systems include a “network infrastructure” that wirelessly communicates with wireless terminals within a respective service coverage area. The network infrastructure typically includes a plurality of base stations dispersed throughout the service coverage area, each of which supports wireless communications within a respective cell (or set of sectors). The base stations couple to base station controllers (BSCs), with each BSC serving a plurality of base stations. Each BSC couples to a mobile switching center (MSC). Each BSC also typically directly or indirectly couples to the Internet.

In operation, each base station communicates with a plurality of wireless terminals operating in its serviced cell/sectors. A BSC coupled to the base station routes voice communications between the MSC and the serving base station. The MSC routes the voice communication to another MSC or to the PSTN. BSCs route data communications between a servicing base station and a packet data network that may include or couple to the Internet. Transmissions from base stations to wireless terminals are referred to as “forward link” transmissions while transmissions from wireless terminals to base stations are referred to as “reverse link” transmissions. The volume of data transmitted on the forward link typically exceeds the volume of data transmitted on the reverse link. Such is the case because data users typically issue commands to request data from data sources, e.g., web servers, and the web servers provide the data to the wireless terminals.

Wireless links between base stations and their serviced wireless terminals typically operate according to one (or more) of a plurality of operating standards. These operating standards define the manner in which the wireless link may be allocated, setup, serviced, and torn down. Popular currently employed cellular standards include the Global System for Mobile telecommunications (GSM) standards, the North American Code Division Multiple Access (CDMA) standards, and the North American Time Division Multiple Access (TDMA) standards, among others. These operating standards support both voice communications and data communications. More recently introduced operating standards include the Universal Mobile Telecommunications Services (UMTS)/Wideband CDMA (WCDMA) standards. The UMTS/WCDMA standards employ CDMA principles and support high throughput, both voice and data.

The wireless link between a base station and a serviced wireless terminal is referred to as a “channel.” The channel distorts and adds noise to wireless transmissions serviced by the channel. “Channel equalization” is a process employed by a wireless receiver, e.g., wireless terminal, in an attempt to obviate the effects of the channel. While channel equalization is certainly helpful in obviating the effects of the channel, the characteristics of the channel are constantly changing. Coefficients of a channel equalizer must be continually updated. However, generating coefficients of the channel equalizer is a difficult and time consuming process. Thus, a need exists for an improved methodology for determining equalizer coefficients.

The present invention is directed to apparatus and methods of operation that are further described in the following Brief Description of the Drawings, the Detailed Description of the Invention, and the claims. Other features and advantages of the present invention will become apparent from the following detailed description of the invention made with reference to the accompanying drawings.

**100** that supports wireless terminals operating according to the present invention. The cellular wireless communication system **100** includes a Public Switched Telephone Network (PSTN) Interface **101**, e.g., Mobile Switching Center, a wireless network packet data network **102** that includes GPRS Support Nodes, EDGE Support Nodes, WCDMA Support Nodes, and other components, Radio Network Controllers/Base Station Controllers (RNC/BSCs) **152** and **154**, and base stations/node Bs **103**, **104**, **105**, and **106**. The wireless network packet data network **102** couples to additional private and public packet data networks **114**, e.g., the Internet, WANs, LANs, etc. A conventional voice terminal **121** couples to the PSTN **110**. A Voice over Internet Protocol (VoIP) terminal **123** and a personal computer **125** couple to the Internet/WAN **114**. The PSTN Interface **101** couples to the PSTN **110**. Of course, this particular structure may vary from system to system.

Each of the base stations/node Bs **103**-**106** services a cell/set of sectors within which it supports wireless communications. Wireless links that include both forward link components and reverse link components support wireless communications between the base stations and their serviced wireless terminals. These wireless links support digital data communications, VoIP communications, and digital multimedia communications. The cellular wireless communication system **100** may also be backward compatible in supporting analog operations as well. The cellular wireless communication system **100** supports one or more of the UMTS/WCDMA standards, the Global System for Mobile telecommunications (GSM) standards, the GSM General Packet Radio Service (GPRS) extension to GSM, the Enhanced Data rates for GSM (or Global) Evolution (EDGE) standards, one or more Wideband Code Division Multiple Access (WCDMA) standards, and/or various other CDMA standards, TDMA standards and/or FDMA standards, etc.

Wireless terminals **116**, **118**, **120**, **122**, **124**, **126**, **128**, and **130** couple to the cellular wireless communication system **100** via wireless links with the base stations/node Bs **103**-**106**. As illustrated, wireless terminals may include cellular telephones **116** and **118**, laptop computers **120** and **122**, desktop computers **124** and **126**, and data terminals **128** and **130**. However, the cellular wireless communication system **100** supports communications with other types of wireless terminals as well. As is generally known, devices such as laptop computers **120** and **122**, desktop computers **124** and **126**, data terminals **128** and **130**, and cellular telephones **116** and **118**, are enabled to “surf” the Internet (packet data network) **114**, transmit and receive data communications such as email, transmit and receive files, and to perform other data operations. Many of these data operations have significant download data-rate requirements while the upload data-rate requirements are not as severe. Some or all of the wireless terminals **116**-**130** are therefore enabled to support the EDGE operating standard, the GPRS standard, the UMTS/WCDMA standards, the HSDPA standards, the WCDMA standards, and/or the GSM standards. Further, some or all of the wireless terminals **116**-**130** are enabled to perform equalization operations of the present invention to support these high speed data operating standards.

**202** and an associated radio **204**. For cellular telephones, the host processing components and the radio **204** are contained within a single housing. In some cellular telephones, the host processing components **202** and some or all of the components of the radio **204** are formed on a single Integrated Circuit (IC). For personal digital assistants hosts, laptop hosts, and/or personal computer hosts, the radio **204** may reside within an expansion card or upon a mother board and, therefore, be housed separately from the host processing components **202**. The host processing components **202** include at least a processing module **206**, memory **208**, radio interface **210**, an input interface **212**, and an output interface **214**. The processing module **206** and memory **208** execute instructions to support host terminal functions. For example, for a cellular telephone host device, the processing module **206** performs user interface operations and executes host software programs among other operations.

The radio interface **210** allows data to be received from and sent to the radio **204**. For data received from the radio **204** (e.g., inbound data), the radio interface **210** provides the data to the processing module **206** for further processing and/or routing to the output interface **214**. The output interface **214** provides connectivity to an output display device such as a display, monitor, speakers, et cetera such that the received data may be displayed. The radio interface **210** also provides data from the processing module **206** to the radio **204**. The processing module **206** may receive the outbound data from an input device such as a keyboard, keypad, microphone, et cetera via the input interface **212** or generate the data itself. For data received via the input interface **212**, the processing module **206** may perform a corresponding host function on the data and/or route it to the radio **204** via the radio interface **210**.

Radio **204** includes a host interface **220**, baseband processing module **222** (baseband processor) **222**, analog-to-digital converter **224**, filtering/gain module **226**, down conversion module **228**, low noise amplifier **230**, local oscillation module **232**, memory **234**, digital-to-analog. converter **236**, filtering/gain module **238**, up-conversion module **240**, power amplifier **242**, RX filter module **264**, TX filter module **258**, TX/RX switch module **260**, and antenna **248**. Antenna **248** may be a single antenna that is shared by transmit and receive paths (half-duplex) or may include separate antennas for the transmit path and receive path (full-duplex). The antenna implementation will depend on the particular standard to which the wireless communication device is compliant.

The baseband processing module **222** in combination with operational instructions stored in memory **234**, execute digital receiver functions and digital transmitter functions. The digital receiver functions include, but are not limited to, digital intermediate frequency to baseband conversion, demodulation, constellation demapping, descrambling, and/or decoding,. The digital transmitter functions include, but are not limited to, encoding, scrambling, constellation mapping, modulation, and/or digital baseband to IF conversion. The transmit and receive functions provided by the baseband processing module **222** may be implemented using shared processing devices and/or individual processing devices. Processing devices may include microprocessors, micro-controllers, digital signal processors, microcomputers, central processing units, field programmable gate arrays, programmable logic devices, state machines, logic circuitry, analog circuitry, digital circuitry, and/or any device that manipulates signals (analog and/or digital) based on operational instructions. The memory **234** may be a single memory device or a plurality of memory devices. Such a memory device may be a read-only memory, random access memory, volatile memory, non-volatile memory, static memory, dynamic memory, flash memory, and/or any device that stores digital information. Note that when the baseband processing module **222** implements one or more of its functions via a state machine, analog circuitry, digital circuitry, and/or logic circuitry, the memory storing the corresponding operational instructions is embedded with the circuitry comprising the state machine, analog circuitry, digital circuitry, and/or logic circuitry.

In operation, the radio **204** receives outbound data **250** from the host processing components via the host interface **220**. The host interface **220** routes the outbound data **250** to the baseband processing module **222**, which processes the outbound data **250** in accordance with a particular wireless communication standard (e.g., UMTS/WCDMA, GSM, GPRS, EDGE, et cetera) to produce digital transmission formatted data **252**. The digital transmission formatted data **252** is a digital base-band signal or a digital low IF signal, where the low IF will be in the frequency range of zero to a few kilohertz/megahertz.

The digital-to-analog converter **236** converts the digital transmission formatted data **252** from the digital domain to the analog domain. The filtering/gain module **238** filters and/or adjusts the gain of the analog signal prior to providing it to the up-conversion module **240**. The up-conversion module **240** directly converts the analog baseband or low IF signal into an RF signal based on a transmitter local oscillation **254** provided by local oscillation module **232**. The power amplifier **242** amplifies the RF signal to produce outbound RF signal **256**, which is filtered by the TX filter module **258**. The TX/RX switch module **260** receives the amplified and filtered RF signal from the TX filter module **258** and provides the output RF signal **256** signal to the antenna **248**, which transmits the outbound RF signal **256** to a targeted device such as a base station **103**-**106**.

The radio **204** also receives, an inbound RF signal **262**, which was transmitted by a base station via the antenna **248**, the TX/RX switch module **260**, and the RX filter module **264**. The low noise amplifier **230** receives inbound RF signal **262** and amplifies the inbound RF signal **262** to produce an amplified inbound RF signal. The low noise amplifier **230** provides the amplified inbound RF signal to the down conversion module **228**, which converts the amplified inbound RF signal into an inbound low IF signal or baseband signal based on a receiver local oscillation **266** provided by local oscillation module **232**. The down conversion module **228** provides the inbound low IF signal (or baseband signal) to the filtering/gain module **226**, which filters and/or adjusts the gain of the signal before providing it to the analog to digital converter **224**. The analog-to-digital converter **224** converts the filtered inbound low IF signal (or baseband signal) from the analog domain to the digital domain to produce digital reception formatted data **268**. The baseband processing module **222** demodulates, demaps, descrambles, and/or decodes the digital reception formatted data **268** to recapture inbound data **270** in accordance with the particular wireless communication standard being implemented by radio **204**. The host interface **220** provides the recaptured inbound data **270** to the host processing components **202** via the radio interface **210**.

As the reader will appreciate, all components of the radio **204**, including the baseband processing module **222** and the RF front end components may be formed on a single integrated circuit. In another construct, the baseband processing module **222** and the RF front end components of the radio **204** may be formed on separate integrated circuits. The radio **204** may be formed on a single integrated circuit along with the host processing components **202**. In still other embodiments, the baseband processing module **222** and the host processing components **202** may be formed on separate integrated circuits. Thus, all components of **222** equalizes the digital transmission formatted data (baseband TX signal) **252** in a novel manner. Various techniques for performing these equalization operations will be described further herein with reference to

**300** constructed according to an embodiment of the present invention. The radio **300** includes a baseband processing module **222** and a plurality of RF front ends, including RF front end **1** **302**, RF front end **2** **304**, RF front end **3** **306**, and RF front end N **308**. These RF front ends **302**, **304**, **306**, and **308** are serviced by antennas **310**, **312**, **318**, and **316**, respectively. The radio **300** may service a plurality of diversity paths of a single transmitted signal. Thus, in one simple embodiment of a diversity path implementation, the radio **300** includes a first RF front end **302**, a second RF front end **304**, and the baseband processing module **222**. This embodiment will be described further with reference to **302**-**308** may service Multiple Input Multiple Output (MIMO) communications, each RF front end **302**-**308** assigned a respective MIMO data path. MIMO communications are currently implemented in WLAN implementations such as IEEE 802.11n. In either case, the principles of the present invention may be applied to a radio **300** having two or more RF front ends.

**222** according to an embodiment of the present invention. The baseband processing module (baseband processor) **222** includes a processor **402**, a memory interface **404**, onboard memory **406**, a downlink/uplink interface **408**, TX processing components **410**, and a TX interface **412**. The baseband processing module **222** further includes an RX interface **414**, a cell searcher module **416**, a multi-path scanner module **418**, a rake receiver combiner **420**, and a turbo decoding module **422**. The baseband processing module **222** couples in some embodiments to external memory **234**. However, in other embodiments, memory **406** fulfills the memory requirements of the baseband processing module **402**.

As was previously described with reference to **250** from coupled host processing components **202** and provides inbound data **270** to the coupled host processing components **202**. Further, the baseband processing module **222** provides digital formatted transmission data (baseband TX signal) **252** to a coupled RF front end. The baseband processing module **222** receives digital reception formatted data (baseband RX signal) **268** from the coupled RF front end. As was previously described with reference to **222** produces the digital reception formatted data (baseband RX data) **268** while the DAC **236** of the RF front end receives the digital transmission formatted data (baseband TX signal) **252** from the baseband processing module **222**.

According to the particular embodiment of the present invention illustrated in **408** is operable to receive the outbound data **250** from coupled host processing components, e.g., the host processing component **202** via host interface **220**. Further, the downlink/uplink interface **408** is operable to provide inbound data **270** to the coupled host processing components **202** via the host interface **220**. TX processing component **410** and TX interface **412** communicatively couple to the RF front end as illustrated in **408**. The TX processing components **410** and TX interface **412** are operable to receive the outbound data from the downlink/uplink interface **404**, to process the outbound data to produce the baseband TX signal **252** and to output the baseband TX signal **252** to the RF front end as was described with reference to **414**, rake receiver combiner **420** and in some cases the processor **402** are operable to receive the RX baseband signal **268** from the RF front end.

Equalization processing operations implemented in an RF receiver according to the present invention may be implemented by one or more of the components of the baseband processing module **222**. In a first construct, the equalization operations are implemented as equalization operations **415** *a *by processor **402**. The equalization operations **415** *a *may be implemented in software, hardware, or a combination of software and hardware. When the equalization operations **415** *a *are implemented by software instructions, the processor **402** retrieves instructions via memory interface **404** and executes such software instructions to implement the equalization operations **415** *a. *

In another construct, a dedicated equalization block **415** *b *resides between the RX interface **414** and modules **416**, **418**, and **420** and performs the equalization operations of the present invention. With this construct, the equalization operations **415** *b *may be implemented via hardware, software, or a combination of hardware and software. In another construct of the equalization operations according to the present invention, the equalization operations **415** *c *are performed within rake receiver combiner module **420** by equalization operations **415** *c*. The equalization operations **415** *c *may be implemented via hardware, software, or a combination of these to execute the equalization operations of the present invention.

As is further shown in **268** may include a plurality of signal paths. Each one of these signal paths may be received from a respective RF front end such as was illustrated in **268** may be a different multi-path version of a single received signal or different RF signal such as in a MIMO system.

**222** perform equalization operations according to the present invention. Of course, a baseband processing module **222** would include additional components in addition to as those illustrated in

The components of the baseband processing module **222** of

The first diversity path component includes a cluster path processor/channel estimation block **504**, a Fast Fourier Transform (FFT) block **506**, multiplier **512**, Inverse Fast Fourier Transform (IFFT) block **514**, tap ordering block **516**, and time domain equalizer **518**. The second diversity path components include cluster path processing/channel estimation block **524**, FFT block **526**, multiplier **530**, IFFT block **532**, tap ordering block **534**, and time domain equalizer **536**. The shared processing blocks of the RF receiver of **510**, a noise variance estimation block **502**, and a combiner **538**.

In its operations, the first diversity path operates upon a first time domain signal **502**. The first time domain signal **502** includes first time domain training symbols and first time domain data symbols. As is generally known, frames of transmitted symbols in an RF system typically include a preamble that has training symbols and a payload portion that carries data symbols. The training symbols are used by channel estimation operations to produce equalizer coefficients that are then used for equalization of the data symbols. The CPP/channel estimation block **504** is operable to process the first time domain training symbols of the first time domain signal **502** to produce a first time domain channel estimate **508**. The FFT block **506** is operable to invert the first time domain channel estimate to the frequency domain to produce a first frequency domain channel estimate **508**.

Likewise, the second diversity path is operable to receive a second time domain signal **522** that includes second time domain training symbols and second time domain data symbols. The CPP/channel estimation block **524** is operable to process the second time domain training symbols to produce a second time domain channel estimate. The FFT block **526** is operable to convert the second time domain channel estimate to the frequency domain to produce a second frequency domain channel estimate **528**.

The MMSE/weight calculation block **510** is operable to receive noise variance estimation parameters from noise variance estimation block **502** and to produce first frequency domain equalizer coefficients **511** and second frequency domain equalizer coefficients **513** based upon the first frequency domain channel estimate **508** and the second frequency domain channel estimate **528**. The operations of the noise variation estimation block **502** will be described further herein with reference to

Referring again to the first diversity path, the multiplier **512** is operable to multiply an output of FFT block **506** with the first frequency domain equalizer coefficients **511**. However, in another embodiment, the multiplier **518** simply passes through the first frequency domain equalizer coefficients **511**. Then, the IFFT block **514** is operable to convert the first frequency domain equalizer coefficients **511**, as operated upon by multiplier **512**, to the time domain to produce first time domain equalizer coefficients. Next, the tap ordering block **516** is operable to order the first time domain equalizer coefficients to produce tap ordered time domain equalizer coefficients to the time domain equalizer **518**. Time domain equalizer **518** is operable to equalize the first time domain data symbols using the first time domain equalizer coefficients received from tap ordering block **516**.

Referring again to the second diversity path, the multiplier **530** is operable to multiply the second frequency domain equalizer coefficients **513** with an output received from FFT block **526**. In another embodiment, the multiplier block **530** is operable to simply pass through the second frequency domain equalizer coefficients **513**. The IFFT block **532** is operable to convert its input from the frequency domain to the time domain to produce second time domain equalizer coefficients. The tap ordering block **534** is operable to tap order the second time domain equalizer coefficients to produce an output of time domain equalizer. The time domain equalizer **536** is operable to equalize the second time data symbols using the second time domain equalizer coefficients. Finally, combiner **538** is operable to combine the equalized first time domain data symbols received from the first time domain equalizer **518** and the second equalized time domain data symbols received from time domain equalizer **536** to produce a composite time domain data symbols **540**.

According to another aspect of the baseband processing module **222** of **504** is operable to cluster path process the first time domain training signals of the first time domain signal **502**. Cluster path processing (CPP) is an operation that processes multi-path signal components that are relatively close to one another in time. A complete description of how cluster path processing is performed is described in co-pending patent application Ser. No. 11/173,854 filed Jun. 30, 2005 and entitled METHOD AND SYSTEM FOR MANAGING, CONTROLLING, AND COMBINING SIGNALS IN A FREQUENCY SELECTIVE MULTIPATH FADING CHANNEL, which is incorporated herein by reference in its entirety. With the cluster path processing operations completed, the CPP/channel estimation block **504** is operable to produce the first time domain channel estimate based upon cluster path processed first time domain training symbols. Further, with the second diversity path, the CPP/channel estimation block **522** may be operable to cluster path process the second time domain training symbols of the second time domain signal **522**. Then, the CPP/channel estimation block **524** is operable to produce the second time domain channel estimate based upon the cluster path process second time domain training symbols.

In its operations, the MMSE weight calculation block **510** is operable to perform a MMSE algorithm on the first frequency domain equalizer coefficients **508** and the second frequency domain equalizer coefficients **528** to produce the first frequency domain equalizer coefficients **511** and the second frequency domain equalizer coefficients **513**. One implementation of these operations is described below. Other operations may be used to generate equalizer coefficients according to the present invention that differ from those described below.

With the particular implementation described herein, in the time domain, a matrix signal model at each antenna servicing the dual diversity path structure of

*y* _{i} *=H* _{i} *x+n* _{1 } *i=*1,2 (Eq. 1)

The channel matrix H_{i }can be modeled as a circulant matrix which satisfies

H_{1}=F^{−1}Λ_{1}F; H_{2}=F^{−1}Λ_{2}F (Eq. 2)

where F is the orthogonal discrete Fourier transform matrix.

By multiplying by matrix F at both sides of the Eq. (1), a frequency domain channel model is represented as:

*Y* _{i} *=Fy* _{i}=Λ^{i} *X+N* _{i} (Eq. 3)

where X=Fx; N_{i}=Fn_{i }i=1,2

The channel model may be considered at the k-th subcarrier in the frequency domain as:

are 2×1 vectors.

The MMSE optimum weight at the k-th subcarrier is therefore represented by:

Thus, the estimated transmitted signal is given as

After simplifying Eq (7), the MMSE-FDE weight(s) for dual diversity path configuration of

The time domain signal after Equalization is given by:

**222** are operable to receive a time domain signal **602** from an RF front end such as was illustrated in **602** includes time domain training symbols and time domain data symbols. The components of **604**, an FFT block **606**, a weight calculator block **610**, an IFFT block **614**, a tap ordering block **616**, and a time domain equalizer **618**. The channel estimation block **604** is operable to process the time domain training symbols of the time domain signal **602** to produce a time domain channel estimate **603**. The FFT block **606** is operable to convert the time domain channel estimate **603** to the frequency domain to produce a frequency domain channel estimate **608**. The weight calculation block **610** is operable to produce frequency domain equalizer coefficients based upon the frequency domain channel estimate **608** and noise variation estimation received from noise variation estimation block **602**. The operations of the noise variation estimation block **602** will be described further herein with reference to **612** receives the frequency domain equalizer coefficient **611** and the receiving input from the FFT block **606**. The multiplier **612** produces an output to IFFT block **614** that converts the frequency domain equalizer coefficient **611**, as may have been modified by multiplier **612**, to produce time domain equalizer coefficients. Tap ordering block **616** tap orders the time domain equalizer coefficients and produces the tap ordered time domain equalizer coefficients to time domain equalizer **616**. The time domain equalizer **616** is operable to equalize the time domain data symbols of the time domain signal **602** using the time domain equalizer coefficients to produce equalized time domain symbols **640**.

The channel estimation block **604** may also perform cluster path processing operations as were previously described with reference to **604** may produce the time domain channel estimate based upon the cluster path processed time domain training symbols. The MMSE weight calculation block **610** may perform MMSE algorithm on the frequency domain equalizer coefficients to produce the frequency domain equalizer coefficients.

**700** commences with operations for each of at least two diversity paths (Step **702**). As was previously described with reference to **302**-**308**, each servicing a respective diversity path. Thus, referring again to **704**-**708** are performed for each diversity path. In particular, for each diversity path, the baseband processing module receives a corresponding time domain signal that includes corresponding time domain training symbols and corresponding time domain data symbols.

With reference to a first diversity path, operation includes receiving a first time domain signal that includes first time domain training symbols and first time domain data symbols. Operation then includes processing the first time domain training symbols to produce a first time domain channel estimate (Step **706**). Further, operation includes converting the time domain channel estimate to the frequency domain to produce a first frequency domain channel estimate (Step **708**).

With respect to a second diversity path, operation includes receiving a second time domain signal that includes second time domain training symbols and second time domain data symbols (Step **704**). Operation for the second diversity path further includes processing the second time domain training symbols to produce a second time domain channel estimate (Step **706**). Further, operation includes converting the second time domain channel estimate to the frequency domain to produce a second frequency domain channel estimate (Step **708**).

When the operations of Steps **702**-**708** have been completed for each diversity path, operation proceeds to Step **710** where frequency domain equalizer coefficients are produced for each of a plurality of diversity paths. For the particular example of the structure of **710** includes producing first frequency domain equalizer coefficients and second frequency domain equalizer coefficients based upon the first frequency domain channel estimate and the second frequency domain channel estimate. Operation then includes converting the frequency domain equalizer coefficients to time domain equalizer coefficients (Step **712**). For the particular case of a first and a second diversity path, the operation at Step **712** would include converting the first frequency domain equalizer coefficients to the time domain to produce first time domain equalizer coefficients and converting the second frequency domain equalizer coefficients to the time domain to produce second time domain equalizer coefficients.

Operation then includes, for each diversity path, time domain equalizing respective time domain data symbols (Step **714**). For the particular case of a first and a second diversity path, the operations of Step **714** include equalizing the first time domain data symbols using the first time domain equalizer coefficients and equalizing the second time domain data symbols using the second time domain equalizer coefficients. Finally, operation includes combining the equalized time domain data symbols from a plurality of diversity paths (Step **716**). For the particular case of the first and second diversity paths, the operation of Step **716** includes combining the equalized first time domain data symbols and the second equalized time domain data symbols to produce composite time domain data symbols.

The operations **702**-**716** are repeated each time new equalizer coefficients are produced based upon received physical layer frames that include training symbols. In many RF receivers, the operations **700** of

The operations at Step **706** may include cluster path processing as has been previously described. When cluster processing is performed, the time domain channel estimate include cluster path processed time domain training symbols. Fast Fourier transformations are employed in converting from the time domain to the frequency domain while Inverse Fast Fourier transformations are to employed to convert from the frequency domain to the time domain. The operations at Step **710** may include using an MMSE algorithm to produce the frequency domain equalizer coefficients based upon the channel estimates received. The operations of

**800** includes first receiving a time domain signal that includes time domain training symbols and time domain data symbols (Step **802**). Operation continues with processing the time domain training symbols to produce a time domain channel estimate (Step **804**). Operation continues with converting the time domain channel estimate to the frequency domain to produce a frequency domain channel estimate (Step **806**).

Operation further includes producing frequency domain equalizer coefficients based upon the frequency domain channel estimate produced at Step **806** (Step **808**). Then, operation includes converting the frequency domain equalizer coefficients to the time domain to produce time domain equalizer coefficients (Step **810**). Operation concludes with equalizing the time domain data symbols using the time domain equalizer coefficients produced at Step **810** (Step **812**). From Step **812** operation ends. Of course, the operations **800** of **800** of

**902** and a dominate interfering signal **906** are operated upon by an RF signal. With the model of **914** that is the combination of the desired signal that has been operated upon by desired signal channel **904**, the interfering signal **906** that has been operated upon by interfering signal channel **908**, and noise **910**, all of these additive (represented by adder **912**). According to the present invention, equalizer coefficients are generated that substantially/full cancel the component of the RX signal **914** caused by the interfering signal **906**.

In a one dominated interferer case, the signal model at the k-subcarrier of the RX signal **914** in the frequency domain is given by:

*Y[k]=H* _{d} *[k]S+H* _{1} *[k]I+N* (Eq. 11)

Structures and methodologies for removing some/all of the interfering signal by an equalizer constructed according to embodiments of the present invention will be described further herein with reference to

**222** perform equalization operations according to the present invention. Of course, a baseband processing module **222** would include additional components in addition to as those illustrated in

The components of the baseband processing module **222** of

The first diversity path component includes a desired signal cluster path processor/channel estimation block **1004**, an interfering signal cluster path processor/channel estimation block **1042**, a Fast Fourier Transform (FFT) block **1006**, a Fast Fourier Transform (FFT) block **1044**, multiplier **1012**, Inverse Fast Fourier Transform (IFFT) block **1014**, tap ordering block **1016**, and time domain equalizer **1018**. The second diversity path components include desired signal cluster path processing/channel estimation block **1024**, interfering signal cluster path processing/channel estimation block **1048**, FFT block **1026**, FFT block **1050**, multiplier **1030**, IFFT block **1032**, tap ordering block **1034**, and time domain equalizer **1036**.

The shared processing blocks of the RF receiver of **1056**, joint scrambling and code tracking for interference block **1054**, a Minimum Mean Square Error (MMSE) weight calculation block **1010**, a noise variance estimation block **1002**, and a combiner **1038**. Operations regarding the joint DLL **1056** will be describe further below. Generally, the joint DLL **1056** is controlled by CPP operations and set the sampling positions of the CPP/channel estimation blocks **1004**, **1042**, **1024**, and **1048**. The joint scrambling and code tracking for interference block **1054** provides scrambling and code tracking information to the interfering signal CPP/channel estimation bocks **1042** and **1048**.

In its operations, the first diversity path operates upon a first time domain signal **1002**. The first time domain signal **1002** includes desired signal time domain training symbols and data symbols and interfering signal time domain training symbols and data symbols. As is generally known, frames of transmitted symbols in an RF system typically include a preamble that has training symbols and a payload portion that carries data symbols. The training symbols are used by channel estimation operations to produce equalizer coefficients that are then used for equalization of the data symbols. The desired signal CPP/channel estimation block **1004** is operable to process desired signal time domain training symbols of the first time domain signal **1002** to produce a first desired signal time domain channel estimate. The interfering signal CPP/channel estimation block **1042** is operable to process interfering signal time domain training symbols of the first time domain signal **1002** to produce a first interfering signal time domain channel estimate. In producing the their respective channel estimates, the CPP/channel estimation blocks **1004** and **1042** may receive estimates of energies of the desired and interfering signals from desired signal energy estimation block **1056** and interfering energy estimation block **1054**, respectively. The CPP/channel estimation blocks **1004** and/or **1042** is/are operable to perform cluster path processing. The FFT block **1006** is operable to convert the first desired signal time domain channel estimate to the frequency domain to produce a first desired signal frequency domain channel estimate **1008**. The FFT block **1044** is operable to convert the first interfering signal time domain channel estimate to the frequency domain to produce a first desired signal frequency domain channel estimate **1046**.

Likewise, the second diversity path operates upon a second time domain signal **1022**. The second time domain signal **1022** includes desired signal time domain training symbols and data symbols and interfering signal time domain training symbols and data symbols. The desired signal CPP/channel estimation block **1024** is operable to process desired signal time domain training symbols of the second time domain signal **1022** to produce a second desired signal time domain channel estimate. The interfering signal CPP/channel estimation block **1048** is operable to process interfering signal time domain training symbols of the second time domain signal **1022** to produce a second interfering signal time domain channel estimate. In producing the their respective channel estimates, the CPP/channel estimation blocks **1024** and **1048** may receive estimates of energies of the desired and interfering signals from desired signal energy estimation block **1056** and interfering energy estimation block **1054**, respectively. The CPP/channel estimation blocks **1024** and/or **1048** is/are operable to perform cluster path processing. The FFT block **1026** is operable to convert the second desired signal time domain channel estimate to the frequency domain to produce a second desired signal frequency domain channel estimate **1028**. The FFT block **1050** is operable to convert the second interfering signal time domain channel estimate to the frequency domain to produce a second desired signal frequency domain channel estimate **1052**.

The MMSE/weight calculation block **1010** is operable to receive noise variance estimation parameters from noise variance estimation block **1002** and to produce first frequency domain equalizer coefficients **1011** and second frequency domain equalizer coefficients **1013** based upon the first desired signal frequency domain channel estimate **1008**, the first interfering signal frequency domain channel estimate **1046**, the second desired signal frequency domain channel estimate **1028**, and the second interfering signal frequency domain channel estimate **1052**. The operations of the noise variation estimation block **1002** will be described further herein with reference to

Referring again to the first diversity path, the multiplier **1012** is operable to multiply an output of FFT block **1006** with the first frequency domain equalizer coefficients **1011**. However, in another embodiment, the multiplier **1018** simply passes through the first frequency domain equalizer coefficients **1011**. Then, the IFFT block **1014** is operable to convert the first frequency domain equalizer coefficients **1011**, as operated upon by multiplier **1012**, to the time domain to produce first time domain equalizer coefficients. Next, the tap ordering block **1016** is operable to order the first time domain equalizer coefficients to produce tap ordered time domain equalizer coefficients to the time domain equalizer **1018**. Time domain equalizer **1018** is operable to equalize the first time domain data symbols using the first time domain equalizer coefficients received from tap ordering block **1016**.

Referring again to the second diversity path, the multiplier **1030** is operable to multiply the second frequency domain equalizer coefficients **1013** with an output received from FFT block **1026**. In another embodiment, the multiplier block **1030** is operable to simply pass through the second frequency domain equalizer coefficients **1013**. The IFFT block **1032** is operable to convert its input from the frequency domain to the time domain to produce second time domain equalizer coefficients. The tap ordering block **1034** is operable to tap order the second time domain equalizer coefficients to produce an output of time domain equalizer. The time domain equalizer **1036** is operable to equalize the second time data symbols using the second time domain equalizer coefficients. Finally, combiner **1038** is operable to combine the equalized first time domain data symbols received from the first time domain equalizer **1018** and the second equalized time domain data symbols received from time domain equalizer **1036** to produce a composite time domain data symbols **1040**.

**1100** commences with operations for each of at least two diversity paths (Step **1102**). As was previously described with reference to **302**-**308**, each servicing a respective diversity path. Thus, referring again to **1104**-**1114** are performed for each diversity path. In particular, for each diversity path, the baseband processing module receives a corresponding time domain signal that includes desired signal time domain training symbols, desired signal time domain data symbols, interfering signal time domain training symbols, and interfering signal time domain data symbols.

With reference to a first diversity path, operation includes receiving a first time domain signal. The first diversity path then estimates energies of a desired signal and at least one interfering signal present in the first time domain signal (Step **1106**). Operation then includes processing the first interfering signal (dominant interferer) time domain training symbols to produce a first interfering signal time domain channel estimate (Step **1108**). Further, operation includes converting the first interfering signal time domain channel estimate to the frequency domain to produce a first desired signal frequency domain channel estimate (Step **1110**). Operation then includes processing the first desired signal time domain training symbols to produce a first desired signal time domain channel estimate (Step **1112**). Further, operation includes converting the first desired signal time domain channel estimate to the frequency domain to produce a first desired signal frequency domain channel estimate (Step **1114**).

With reference to a second diversity path, operation includes receiving a second time domain signal. The second diversity path then estimates energies of a desired signal and at least one interfering signal present in the second time domain signal (Step **1106**). Operation then includes processing the second interfering signal (dominant interferer) time domain training symbols to produce a second interfering signal time domain channel estimate (Step **1108**). Further, operation includes converting the second interfering signal time domain channel estimate to the frequency domain to produce a second desired signal frequency domain channel estimate (Step **1110**). Operation then includes processing the second desired signal time domain training symbols to produce a second desired signal time domain channel estimate (Step **1112**). Further, operation includes converting the second desired signal time domain channel estimate to the frequency domain to produce a second desired signal frequency domain channel estimate (Step **1114**).

When the operations of Steps **1104**-**1114** have been completed for each diversity path, operation proceeds to Step **1116** where frequency domain equalizer coefficients are produced for each of a plurality of diversity paths. For the particular example of the structure of **1116** includes producing first frequency domain equalizer coefficients and second frequency domain equalizer coefficients based upon the first frequency domain channel estimate and the second frequency domain channel estimate. Operation then includes converting the frequency domain equalizer coefficients to time domain equalizer coefficients (Step **1118**). For the particular case of a first and a second diversity path, the operation at Step **1118** would include converting the first frequency domain equalizer coefficients to the time domain to produce first time domain equalizer coefficients and converting the second frequency domain equalizer coefficients to the time domain to produce second time domain equalizer coefficients.

Operation then includes, for each diversity path, time domain equalizing respective time domain data symbols (Step **1120**). For the particular case of a first and a second diversity path, the operations of Step **1120** include equalizing the first time domain data symbols using the first time domain equalizer coefficients and equalizing the second time domain data symbols using the second time domain equalizer coefficients. Finally, operation includes combining the equalized time domain data symbols from a plurality of diversity paths (Step **1122**). For the particular case of the first and second diversity paths, the operation of Step **1122** includes combining the equalized first time domain data symbols and the second equalized time domain data symbols to produce composite time domain data symbols.

The operations **1102**-**1122** are repeated each time new equalizer coefficients are produced based upon received physical layer frames that include training symbols. In many RF receivers, the operations **1100** of

The operations at Steps **1108** and **1112** may include cluster path processing as has been. previously described. When cluster processing is performed, the time domain channel estimates include cluster path processed time domain training symbols. Fast Fourier transformations are employed in converting from the time domain to the frequency domain while Inverse Fast Fourier transformations are to employed to convert from the frequency domain to the time domain. The operations at Step **1116** may include using an MMSE algorithm to produce the frequency domain equalizer coefficients based upon the channel estimates received. The operations of

**222** are operable to receive a time domain signal **1202** from an RF front end such as was illustrated in **1202** includes desired signal time domain training symbols and data symbols and interfering signal time domain training symbols and data symbols. The components of **1204**, interfering signal channel estimation block **1242**, an FFT block **1206**, an FFT block **1244**, a weight calculator block **1210**, an IFFT block **1214**, a tap ordering block **1216**, and a time domain equalizer **1218**. The channel estimation block **1204** is operable to process the desired signal time domain training symbols of the time domain signal **1202** to produce a time domain channel estimate **1203**. The interfering signal estimation block **1204** is operable to process the interfering signal time domain training symbols of the time domain signal **1202** to produce a time domain channel estimate **1203**. The FFT block **1206** is operable to convert the desired signal time domain channel estimate **1203** to the frequency domain to produce a desired signal frequency domain channel estimate **1208**. The FFT block **1244** is operable to convert the interfering signal time domain channel estimate to the frequency domain to produce an interfering frequency domain channel estimate **1246**. The weight calculation block **1210** is operable to produce frequency domain equalizer coefficients based upon the desired signal frequency domain channel estimate **1208**, the interfering signal frequency domain channel estimate **1246**, and noise variation estimation received from noise variation estimation block **1202**. Multiplier **1212** receives the frequency domain equalizer coefficient **1212** and the receiving input from the FFT block **1206**. The multiplier **1212** produces an output to IFFT block **1214** that converts the frequency domain equalizer coefficient **1212**, as may have been modified by multiplier **1212**, to produce time domain equalizer coefficients. Tap ordering block **1216** tap orders the time domain equalizer coefficients and produces the tap ordered time domain equalizer coefficients to time domain equalizer **1216**. The time domain equalizer **1216** is operable to equalize the time domain data symbols of the time domain signal **1202** using the time domain equalizer coefficients to produce equalized time domain symbols **1240**.

The channel estimation blocks **1204** and/or **1242** may also perform cluster path processing operations as were previously described with reference to **1204** and **1242** may produce the time domain channel estimates based upon the cluster path processed time domain training symbols. The MMSE weight calculation block **1210** may perform MMSE algorithm on the frequency domain equalizer coefficients to produce the frequency domain equalizer coefficients.

**1300** includes first receiving a time domain signal that includes desired signal time domain training symbols and data symbols and interfering signal time domain training symbols and data symbols (Step **1302**). Operation continues with estimating energies of a desired signal and at least one interfering signal present in the time domain signal (Step **1304**). Operation next includes processing the time domain training symbols to produce an interfering signal time domain channel estimate (Step **1306**). Operation continues with converting the interfering signal time domain channel estimate to the frequency domain to produce an interfering signal frequency domain channel estimate (Step **1308**). Operation next includes processing the time domain training symbols to produce a desired signal time domain channel estimate (Step **1312**). Operation continues with converting the desired signal time domain channel estimate to the frequency domain to produce a desired signal frequency domain channel estimate (Step **1314**).

Operation further includes producing frequency domain equalizer coefficients based upon the frequency domain channel estimates produced at Steps **1308** and **1314** (Step **13** **16**). Then, operation includes converting the frequency domain equalizer coefficients to the time domain to produce time domain equalizer coefficients (Step **1318**). Operation concludes with equalizing the time domain data symbols using the time domain equalizer coefficients produced at Step **1318** (Step **1320**). From Step **1320** operation ends. Of course, the operations **1300** of

The operations of **1010** and **1210** and implemented at steps **1116** and **1316** may operate according to the following. The MMSE optimum weights computed at each sub-carrier as:

Then, by using IFFT operations, the coefficient of the time domain equalizer can be achieved. The interference suppression capability is demonstrated by:

In more detail, we have

By ignoring the subscription k, (Eq. 12) can be written as:

By defining

The equations simplify to

In order to simply the 2-by-2 matrix Direct Matrix Inversion, the simplified weight calculation method is derived as follows:

By defining

the FDE-IS weight at each subcarrier is given as

In (Eq. 12), we assume that the transmission power of the dominant interference is equal to that of the desired signal and also that the power allocation of the CPICH of the dominant interferer is same as that of the desired signal. These estimations affect the operation of the system of the present invention. Thus, robustness for the impact due to the estimation offset of transmission power and the CPICH power allocation of the dominant interference is addressed according to aspects of the present invention. By define the offset factors γ and ζ as the estimation offset of the transmission power and the CPICH power allocation, respectively. Then, the (Eq. 12) can be written as

In general, we have γ<=1 by setting P_{l}=P_{S}=4.0 at all times, where there is a estimation offset in WCDMA compress mode or discontinuous HSDPA transmission pattern of interference, and 0.4<ζ^{2}<2.5 by assuming the estimated E_{C—}CPICH/Ior of interference equal to −10 dB and the estimation offset is within ±4 dB. By ignoring the subscription k, (Eq. 16 can be written as)

By defining

In following:

By defining

the FDE-IS weight at each sub-carrier is given as

Approximately at high SNR range, we have:

The timing of the CPP processing operations (e.g., blocks **1004**, **1042**, **1024**, and **1048**) according to the present invention is also important, e.g., alignment of the CPP operations with the desired signal and the interfering signal. Consider that a scanner module **418** of the baseband processing module **222** provides timing references to the CPP/Ch_Est blocks **1004**, **1024**, **1042**, and **1048** for the desired signal and the interfering signal around NTc and MTc, respectively. With these assumption, a signal model in the time domain can be represented by:

By assuming:

*t−p*1*=T* _{s}τ_{d} *=NTc+p*1;τ_{i} *=MTc+p*2

(Eq. 18) can be written as

With these assumptions, because we only compensate for the collected energy of the desired signal and its corresponding channel response at a sampling position of both the desired and interfering signals, the optimum time for desired with maximum energy output can be achieved by using a Maximum Energy Delay Locked Loop for the CPP operations. The sampling position of the interfering signal is then slaved to the desired signal DLL. By using on time sampling information from the desired DLL, the channel response for both the desired signal and the interfering signal at the same sampling position can be achieved.

Further, by adjusting scrambling code N chip, we can get channel estimation of the desired signal at a sampling phase P**1**. Similarly, at the interfering signal CPP processing, by adjusting scrambling code M chip, we can get channel estimation of interference at same sampling phase P**1**. Furthermore, by considering that the DLL can track the main path that can shift over 7 chips during scanner update period and the desired and interference can shift in different direction. We have to add an extra DLL to lock the main path of interference. The extra DLL only provide the SC code phase of interference. For example, the SC of interference shift one chip, say m->m+1, because the sampling position slaved to desired DLL. The channel profile of interference is totally shift by 1 chip. Considering we use IIR filter to measure the power. We must use the previous and adjacent finger (L+1) as current finger L's history value.

**1402** of the system includes multiple antennas **1404** and **1406**, each of which transmits an information signal. These information signals S_{1 }and S_{2 }may be Space Time Transmit Diversity (STTD) signals, the formation and transmission of which is generally known and will not be described further other than as they relate to the present invention. In some operations, the first information signal S_{1 }carries the same data as the second information signal S_{2}. The information signals S_{1 }and S_{2 }transmitted via antennas **1404** and **1406** are operated upon by channel **1408** and received together as a merged information signal via antenna **1412** by RF receiver **1410**. The RF receiver **1410** operates upon this merged information signal, as will be described further with reference to

**1452** of the system includes multiple antennas **1454** and **1456**, each of which transmits a respective information signal. These information signals S_{1 }and S_{2 }carry differing data, i.e., a first information signal S_{1 }carries differing data than does second information signal S_{2}. The information signals S_{1 }and S_{2 }transmitted via antennas **1454** and **1456** are operated upon by channel **1458** and received together as merged information signals by antennas **1462** and **1464** by RF receiver **1460**. Because of the operations of the channel **1458**, each of the antennas receives a respective merged information signal that is a combination of the transmitted signals S_{1 }and S_{2}. The RF receiver **1460** operates upon these merged information signals S_{1 }and S_{2}, as will be described further with reference to

**222** are operable to receive a merged information **1502** from an RF front end such as was illustrated in **1502** includes a first information signal and a second information signal, each of the first and second information signals having time domain training symbols and data symbols and is, of course, in the time domain upon receipt. The baseband processing module includes at least one channel estimator **1504** and **1508** operable to process the first information signal time domain training symbols and the second information signal time domain training symbols to produce a first information signal time domain channel estimate and a second information signal time domain channel estimate. The baseband processing module **222** further includes at least one Fast Fourier Transform module **1506** and **1510** operable to transform the first information signal time domain channel estimate and the second information signal time domain channel estimate to the frequency domain to produce a first information signal frequency domain channel estimate and a second information signal frequency domain channel estimate, respectively. The baseband processing module **222** further includes a weight calculator **1512** operable to produce first frequency domain equalizer coefficients and second frequency domain equalizer coefficients based upon the first information signal frequency domain channel estimate, the second information signal frequency domain channel estimate, and noise variance information received from noise variance estimation block **1514**. The baseband processing module **222** also includes at least one Inverse Fast Fourier Transform module **1516** and **1522** operable to transform the first frequency domain equalizer coefficients and the second frequency domain equalizer coefficients to the time domain to produce first time domain equalizer coefficients and second time domain equalizer coefficients, respectively. Further, the baseband processing module further includes at least one equalizer **1520** and **1526** operable to equalize the merged information signal time using the first time domain equalizer coefficients to produce equalized first information signal data symbols and to equalize the merged information signal time using the first time domain equalizer coefficients to produce equalized second information signal data symbols.

In particular, channel estimation block **1504** is operable to process the first information signal time domain training symbols of the merged information **1502** to produce a first information signal time domain channel estimate. The second information signal estimation block **1508** is operable to process the second information signal time domain training symbols of the merged information signal **1502** to produce a second information signal domain channel estimate. FFT block **1506** is operable to convert the first information signal time domain channel estimate to the frequency domain to produce a first information signal frequency domain channel estimate. FFT block **1510** is operable to convert the second information signal time domain channel estimate to the frequency domain to produce a second information signal frequency domain channel estimate. The weight calculation block **1512** is operable to produce first and second frequency domain equalizer coefficients based upon the first information signal frequency domain channel estimate, the second information signal frequency domain channel estimate, and noise variation estimation received from noise variation estimation block **1514**. The operations of the noise variation estimation block **1514** will be described further herein with reference to

IFFT block **1516** converts the first frequency domain equalizer coefficients to the time domain to produce first time domain equalizer coefficients. Tap ordering block **1518** tap orders the first time domain equalizer coefficients and produce tap ordered first time domain equalizer coefficients to time domain equalizer **1520** The time domain equalizer **1520** is operable to equalize the merged information signal **1502** using the time domain equalizer coefficients to produce equalized first information signal data symbols.

IFFT block **1522** converts the second frequency domain equalizer coefficients to the time domain to produce second time domain equalizer coefficients. Tap ordering block **1524** tap orders the second time domain equalizer coefficients and produce tap ordered second time domain equalizer coefficients to time domain equalizer **1526** The time domain equalizer **1526** is operable to equalize the merged information signal **1502** using the time domain equalizer coefficients to produce equalized second information signal data symbols.

Despreader **1530** is operable to despread the equalized first information signal data symbols. Despreader **1528** is operable to despread the equalized second information signal data symbols. STTD decoder **1532** is operable to STTD decode the despread equalized first information signal data symbols and the despread equalized second information signal data symbols.

The channel estimation blocks **1504** and/or **1508** may also perform cluster path processing operations as were previously described with reference to **1504** and **1508** may produce the time domain channel estimates based upon the cluster path processed time domain training symbols. The MMSE weight calculation block **1512** may perform MMSE algorithm on the frequency domain equalizer coefficients to produce the frequency domain equalizer coefficients.

**1600** includes first receiving a merged information signal that includes a first information signal and a second information signal, each of the first and second information signals having time domain training symbols and data symbols (Step **1602**). Operation continues with estimating energies of the information signals and optionally, the energy of at least one interfering signal present in the merged information signal (Step **1604**). Operation next includes processing the first information signal time domain training symbols and the second information signal time domain training symbols to produce a first information signal time domain channel estimate and a second information signal time domain channel estimate (Step **1606**). Operation continues with converting the first information signal time domain channel estimate and the second information signal time domain channel estimate to the frequency domain to produce a first information signal frequency domain channel estimate and a second information signal frequency domain channel estimate, respectively (Step **1608**). Operation next includes producing first frequency domain equalizer coefficients and second frequency domain equalizer coefficients based upon the first information signal frequency domain channel estimate and the second information signal frequency domain channel estimate (Step **1610**). Operation continues with converting the first frequency domain equalizer coefficients and the second frequency domain equalizer coefficients to the time domain to produce first time domain equalizer coefficients and second time domain equalizer coefficients, respectively (Step **1612**).

Then, operation includes equalizing the merged information signal time using the first time domain equalizer coefficients to produce equalized first information signal data symbols and equalizing the merged information signal time using the first time domain equalizer coefficients to produce equalized second information signal data symbols (Step **1614**). Operation continues with despreading and combining the equalized data symbols (Step **1616**). From Step **1616** operation ends. Of course, the operations **1600** of

The operations of Step **1612** may be performed on an STTD signal. In such case, the first information signal data symbols and the second information signal data symbols carry common data. In this case, the method **1600** includes STTD decoding the equalized first information signal data symbols and the equalized second information signal data symbols. The operations of Step **1616** may include despreading the equalized first information signal data symbols and despreading the equalized second information signal data symbols prior to STTD decoding the despread equalized first information signal data symbols and the despread equalized second information signal data symbols as was illustrated in

Referring now to all of **15**, and **16**, a baseband processing module **222** and operations of the present invention may implement the following MISO equations and techniques. In particular, MISO operations **1600** of **222** (in particular the MMSE weight calculation block **1010** of ^{th}-subcarrier in the frequency domain may be modeled as:

*Y[k]=H* _{1} *[k]S* _{1} *+H* _{2} *[k]S* _{2} *+N* (Eq. 19)

By treating the second information signal S_{2 }as interference to the first information signal S_{1}, the FDE-IS for the first information signal S_{1 }with MMSE optimum weight at each sub-carrier is given as

Similarly, by treating the first information signal S_{1 }as interference to the second information signal S_{2}, the FDE-IS for the second information signal S_{2 }with MMSE optimum weight at each sub-carrier is given as:

Then, using IFFT operations, the coefficient of the time domain equalizers (at Step **1112**) is achieved.

**222** perform equalization operations according to one or more embodiments the present invention. Of course, a baseband processing module **222** would include additional components in addition to as those illustrated in

The components of the baseband processing module **222** of

The first diversity path component includes a first information signal cluster path processor/channel estimation block **1704**, a second desired signal cluster path processor/channel estimation block **1708**, a Fast Fourier Transform (FFT) block **1706**, a Fast Fourier Transform (FFT) block **1712**, Inverse Fast Fourier Transform (IFFT) block **1716**, tap ordering block **1718**, time domain equalizer **1720**, Inverse Fast Fourier Transform (IFFT) block **1722**, tap ordering block **1724**, and time domain equalizer **1726**. The second diversity path components include first information signal cluster path processor/channel estimation block **1734**, a second desired signal cluster path processor/channel estimation block **1738**, a Fast Fourier Transform (FFT) block **1736**, a Fast Fourier Transform (FFT) block **1740**, Inverse Fast Fourier Transform (IFFT) block **1748**, tap ordering block **1750**, time domain equalizer **1752**, Inverse Fast Fourier Transform (IFFT) block **1742**, tap ordering block **1744**, and time domain equalizer **1746**.

The shared processing blocks of the RF receiver of **1730**, a Minimum Mean Square Error (MMSE) weight calculation block **1713**, a noise variance estimation block **1714**, combiners **1754** and **1756**, despreaders **1758** and **1760**, and STTD decoder **1762**. Generally, the joint DLL **1730** is controlled by CPP operations that set the sampling positions of the CPP/channel estimation blocks **1704**, **1708**, **1734**, and **1738**.

In its operations, the first diversity path operates upon a first time domain signal (first merged information signal) **1702**. The first time domain signal **1702** includes first information signal time domain training symbols and data symbols and second information signal time domain training symbols and data symbols. As is generally known, frames of transmitted symbols in an RF system typically include a preamble that has training symbols and a payload portion that carries data symbols. The training symbols are used by channel estimation operations to produce equalizer coefficients that are then used for equalization of the data symbols. The first information signal CPP/channel estimation block **1704** is operable to process first information signal time domain training symbols of the first time domain signal **1702** to produce a first information signal time domain channel estimate. The second information signal CPP/channel estimation block **1708** is operable to process second information signal time domain training symbols of the first time domain signal **1702** to produce a second information signal time domain channel estimate. In producing the their respective channel estimates, the CPP/channel estimation blocks **1704** and **1708** may receive estimates of energies of the information and second information signals from an information signal energy estimation block and an interfering energy estimation block, respectively (not shown). The CPP/channel estimation blocks **1704** and/or **1708** is/are operable to perform-cluster path processing. The FFT block **1706** is operable to convert the first information signal time domain channel estimate to the frequency domain to produce a first information signal frequency domain channel estimate. The FFT block **1712** is operable to convert the second information signal time domain channel estimate to the frequency domain to produce a second information signal frequency domain channel estimate.

Likewise, the second diversity path operates upon a second time domain signal (second merged information signal) **1732**. The second time domain signal **1732** includes first information signal time domain training symbols and data symbols and second information signal time domain training symbols and data symbols. The first information signal CPP/channel estimation block **1734** is operable to process the first information signal time domain training symbols of the second time domain signal **1732** to produce a second information signal time domain channel estimate. The second information signal CPP/channel estimation block **1738** is operable to process second information signal time domain training symbols of the second time domain signal **1732** to produce a second information signal time domain channel estimate. In producing their respective channel estimates, the CPP/channel estimation blocks **1734** and **1738** may receive estimates of energies of the information and second information signals from first information signal energy estimation block and interfering energy estimation block, respectively (not shown). The CPP/channel estimation blocks **1734** and/or **1738** is/are operable to perform cluster path processing. The FFT block **1736** is operable to convert the second information signal time domain channel estimate to the frequency domain to produce a second information signal frequency domain channel estimate. The FFT block **1740** is operable to convert the second information signal time domain channel estimate to the frequency domain to produce a second information signal frequency domain channel estimate.

The MMSE/weight calculation block **1713** is operable to produce first frequency domain equalizer coefficients, second frequency domain equalizer coefficients, third frequency domain equalizer coefficients (first frequency domain equalizer coefficients for the second diversity path), and fourth frequency domain equalizer coefficients (second frequency domain equalizer coefficients for the second diversity path) based upon the first information signal frequency domain channel estimate and the first second information signal frequency domain channel estimate received from the first diversity path and the first information signal frequency domain channel estimate, the second information signal frequency domain channel estimate received from the second diversity path, and noise variance estimation parameters received from noise variance estimation block **1714**. The operations of the noise variation estimation block **1714** will be described further herein with reference to

Referring again to the first diversity path, the IFFT block **1716** is operable to convert the first frequency domain equalizer coefficients to the time domain to produce first time domain equalizer coefficients. Next, the tap ordering block **1718** is operable to order the first time domain equalizer coefficients to produce tap ordered first time domain equalizer coefficients to the time domain equalizer **1720**. Time domain equalizer **1720** is operable to equalize the first merged information signal (time domain signal **1702**) using the tap ordered first time domain equalizer coefficients received from tap ordering block **1718**. The IFFT block **1722** is operable to convert the second frequency domain equalizer coefficients to the time domain to produce second time domain equalizer coefficients. Next, the tap ordering block **1724** is operable to order the second time domain equalizer coefficients to produce tap ordered second time domain equalizer coefficients to the time domain equalizer **1726**. Time domain equalizer **1726** is operable to equalize the first merged information signal (time domain signal **1702**) using the tap ordered second time domain equalizer coefficients received from tap ordering block **1724**.

Referring again to the second diversity path, the IFFT block **1742** is operable to convert the third frequency domain equalizer coefficients (first frequency domain equalizer coefficients for the second diversity path) to the time domain to produce third time domain equalizer coefficients (first time domain equalizer coefficients for the second diversity path). Next, the tap ordering block **1744** is operable to order the third time domain equalizer coefficients to produce tap ordered third time domain equalizer coefficients to the time domain equalizer **1746**. Time domain equalizer **1746** is operable to equalize the second merged information signal (time domain signal **1732**) using the tap ordered third time domain equalizer coefficients received from tap ordering block **1744**. The IFFT block **1748** is operable to convert the fourth frequency domain equalizer coefficients (second frequency domain equalizer coefficients for the second diversity path) to the time domain to produce fourth time domain equalizer coefficients (second time domain equalizer coefficients for the second diversity path). Next, the tap ordering block **1750** is operable to order the fourth time domain equalizer coefficients to produce tap ordered fourth time domain equalizer coefficients to the time domain equalizer **1752**. Time domain equalizer **1752** is operable to equalize the second merged information signal (time domain signal **1732**) using the tap ordered fourth time domain equalizer coefficients received from tap ordering block **1750**.

Combiner **1754** is operable to combine the outputs of time domain equalizers **1726** and **1752** while combiner **1756** is operable to combine the outputs of time domain equalizers **1720** and **1746**. Despreader **1758** is operable to despread the output of combiner **1754** while despreader **1760** is operable to despread the output of combiner **1756**. Further, in some embodiments, when STTD is employed, STTD decoder is operable to STTD decode the outputs of despreaders **1758** and **1760**.

**1800** commences with operations for each of at least two diversity paths (Step **1802**). As was previously described with reference to **5**, and **17**, a radio may include a plurality of RF front ends **302**-**308**, each servicing a respective diversity path. Thus, referring again to **1804**-**1814** are performed for each diversity path. In particular, for each diversity path, the baseband processing module receives a merged information signal that includes a first information signal and a second information signal, each of the first and second information signals having time domain training symbols and data symbols.

With reference to a first diversity path, operation includes receiving a first time domain information signal (first merged information signal). The first diversity path then estimates energies of first and second information signals and, in some cases, one or more interfering signals present in the time domain merged information signal (Step **1806**). Operation then includes processing the first information signal (dominant interferer) time domain training symbols to produce a first information signal time domain channel estimate (Step **1808**). Further, operation includes converting the first information signal time domain channel estimate to the frequency domain to produce a first information signal frequency domain channel estimate (Step **1810**). Operation then includes processing the second information signal time domain training symbols to produce a second information signal time domain channel estimate (Step **1812**). Further, operation includes converting the second information signal time domain channel estimate to the frequency domain to produce a second information signal frequency domain channel estimate (Step **1814**).

With reference to a second diversity path, operation includes receiving a second time domain signal (second merged information signal). The second diversity path then estimates energies of first and second information signals and, in some cases, one or more interfering signals present in the time domain merged information signal (Step **1806**). Operation then includes processing the first information signal time domain training symbols to produce a first information signal time domain channel estimate (Step **1808**). Further, operation includes converting the first information signal time domain channel estimate to the frequency domain to produce a first information signal frequency domain channel estimate (Step **1810**). Operation then includes processing the second information signal time domain training symbols to produce a second information signal time domain channel estimate (Step **1812**). Further, operation includes converting the second information signal time domain channel estimate to the frequency domain to produce a second information signal frequency domain channel estimate (Step **1814**).

Steps **1804** through **1814** may be performed for more than two diversity paths. When the operations of Steps **1804**-**1814** have been completed for each diversity path, operation proceeds to Step **1816** where one or more sets of frequency domain equalizer coefficients are produced for each of the plurality of diversity paths. For the particular example of the structure of **1816** includes producing first and second frequency domain equalizer coefficients for each diversity path based upon the frequency domain channel estimates. Operation then includes converting the frequency domain equalizer coefficients to time domain equalizer coefficients (Step **1818**). Operation then includes, for each diversity path, time domain equalizing respective time domain data symbols (Step **1820**). Finally, operation includes combining at least some of the equalized time domain data symbols from the plurality of diversity paths (Step **1822**). These operations may include STTD combining operations.

The operations **1802**-**1822** are repeated each time new equalizer coefficients are produced based upon received physical layer frames that include training symbols. In many RF receivers, the operations **1800** of

The operations at Steps **1808** and **1812** may include cluster path processing as has been previously described. When cluster processing is performed, the time domain channel estimates include cluster path processed time domain training symbols. Fast Fourier transformations are employed in converting from the time domain to the frequency domain while Inverse Fast Fourier transformations are to employed to convert from the frequency domain to the time domain. The operations at Step **1816** may include using an MMSE algorithm to produce the frequency domain equalizer coefficients based upon the channel estimates received. The operations of

The operations of **222** of

*Y[k]=H* _{1} *[k]S* _{1} *+H* _{2} *[k]S* _{2} *+N* (Eq. 22)

By treating the second information signal S_{2 }signal as interference to the first information signal S_{1}, the frequency domain equalizer coefficients for the first information signal S_{1 }signal with MMSE optimum weight at each sub-carrier is given as:

Similarly, by treating the first information signal S_{1 }signal as interference to the second information signal S_{2}, the frequency domain equalizer coefficients for the second information signal S_{2 }with MMSE optimum weight at each sub-carrier is given as:

Then, using IFFT operations, the coefficient of the time domain equalizer are achieved. In more detail, we have

By ignoring the subscript k, we have

After weight simplification processing, we have:

After weight simplification processing, we have:

**1900** supporting multiple RF carriers **1902**, **1904**, and **1906**. The WCDMA RF band(s) **1900** extend across a frequency spectrum and include WCDMA RF carriers **1902**, **1904**, and **1906**. According to one aspect of the present invention, the cell searcher module **316** of the baseband processing module **222** of an RF transceiver that supports WCDMA operations according to the present invention is operable to scan the WCDMA RF band(s) **1900** to identify WCDMA RF energy of at least one WCDMA carrier **1902**, **1904**, or **1906**. During initial cell search operations, the cell searcher module **316** will, in combination with other components of the baseband processing module **222**, identify a strongest WCDMA carrier, e.g., **1904**. Then, the cell searcher module **316** synchronizes to WCDMA signals within the WCDMA carrier **1904**. These WCDMA signals correspond to a particular base station cell or sector. In these initial cell search synchronization operations, the cell searcher module **316** preferably synchronizes to a strongest cell/sector.

WCDMA signals transmitted from multiple base stations/sectors may use a common WCDMA RF carrier **1904**. Alternately, the WCDMA signals from differing base stations/sectors may use differing WCDMA carriers, e.g., **1902** or **1906**. According to the present invention, the cell searcher module **316** and the baseband processing module **222** are operable to synchronize to WCDMA signals from differing cells/sectors operating in one or more of the WCDMA RF bands **1902**, **1904**, or **1906**. Such synchronization operations occur not only for initial cell search but for neighbor cell search or detected cell search operations.

The CPICH carries pre-defined symbols with a fixed rate (30 kbps, hence 10 symbols per time slot) and spreading factor of 256. The channelization code for CPICH is fixed to the 0^{th }code. According to aspects of the present invention that are described further herein with reference to

**2000** of **222**, such as that illustrated in **222** includes hardware that performs operations according to embodiments of the present invention. This hardware may be dedicated hardware, general purpose hardware running software instructions, and/or a combination of general and specialized hardware.

Referring again to **2000** commences with receiving a composite information signal (Step **2002**). The composite signal is a time domain signal that includes a first information signal, a second information signal, and a pilot signal. The first information signal includes information intended for the RF receiver. In one embodiment, the second information signal is an interfering signal that does not carry data intended for the RF receiver. However, in another embodiment, the second information signal also carries data intended for the RF receiver. As was previously illustrated with reference to

Operation continues with determining channel estimates for the information signals (Step **2004**). As is generally known, a channel estimate may be determined for the first information signal based upon training symbols of the first information signal. Likewise, the channel estimate for the second information signal may be determined based upon training symbols of the second information signal. Then, operation **2000** includes determining channel powers based upon the channel estimates determined at Step **2004** (Step **2006**). The operation of Steps **2004** and **2006** are performed for at least two information signals. However, the operations **2004** and **2006** could be performed for more than two information signals. The reader should recall that the second information signal could include data intended for the RF receiver or may be an interfering signal.

Operation **2000** proceeds with determining a composite signal power of the composite signal (Step **2008**). Then, operation **2000** includes determining a noise variance of the composite signal based upon the first information signal channel power, the second information signal channel power, and the composite signal power (Step **2010**). The operations of Step **2004**-**2010** will be further described herein with reference to particular structures illustrated in **2000** includes determining a plurality of equalizer coefficients based upon the information signal channel estimates and the determined noise variance (Step **2012**). The manner in which these equalizer coefficients are determined has been previously described herein with reference to single received path structures and multiple received path structures. Further, as described herein, the calculation of the plurality of equalizer coefficients may include both frequency domain and time domain operations that have been previously described with reference to

**2102** that is the composite signal is operated on by a plurality of components. A first information signal channel estimate block **2104** receives the time domain signal **2102** and not only produces a channel estimate but also produces a first information signal channel power (Ch_{—}1_pwr). Likewise, a second information signal channel estimate block **2106** receives the time domain signal **2102**, estimates the second information signal channel based upon training symbols, and produces a second information signal channel power (Ch_{—}2_pwr). Composite signal power estimation block **2108** receives time domain signal **2102** and produces a composite signal power estimate (I_{O}). Noise variance estimation block **2110** receives the first information signal channel power, the second information signal channel power, the composite signal power estimate, and an automatic gain control (AGC) estimate from AGC block **2112**. Based on these inputs, the noise variance estimation block **2110** produces a noise variance (Prior_AGC_Nvar) that has not been AGC automatic gain control adjusted. One particular embodiment of the composite signal power estimation block **2108** and the noise variance estimation block **2110** are described further with reference to **2114** based upon input received from AGC magnitude block **2116**. The output of multiplier **2114** is a post AGC variance (Post_AGC_Nvar). Then, this post AGC noise variance is adjusted by pilot signal power to interference ratio scaling by multiplier **2118** based upon input received by pilot signal to interference block **2120**. The output of multiplier **2118** is the noise variance (P_{N}) that is used in equalizer coefficient determination operations.

**2108** receives the time domain signal **2102**. A magnitude block **2202** of the composite signal power block **2108** produces a power representation of the time domain signal **2102**. The power representation of the time domain signal **2102** is accumulated over a plurality of chips by adder **2204** and then normalized over the plurality of chips by divider **2206**. The summing block **2204** sums the output of magnitude determination block **2202** for a number of chips that may be **512** chips, for example. The summation block **2204** is reset as is required based upon the averaging operations. The output of the composite signal power block **2108** is the composite signal power (I_{O}) of the time domain signal **2102** (composite signal).

The noise variance estimation block **2110** receives the first information signal channel power (Ch_{—}1_Pwr) and the second information signal channel power (Ch_{—}2_Pwr). Scalers **2208** and **2210** scale the first and second information signal channel powers based upon a scaling factor. The output of scalers **2208** and **2210** is summed by summing block **2212** and then AGC normalized with a presumed AGC factor 4.0 by multiplier **2214**. Then, the output of multiplier **2214** is acted upon along with the composite signal power received from composite signal power **2108** by subtraction block **2216**. The output of subtraction block **2216** is averaged over a time slot of the time domain signal **2102**. For example, with the example of the WCDMA system of **2218** produces the magnitude of the AGC operator **2212**. The output of the averaging block **2216** is AGC adjusted by divider operator **2220** based upon an input received from AGC magnitude block **2218**. The output of the divider operator **2220** is filtered by infinite impulse response filter **2222**, which produces the Prior_AGC_Nvar that was previously indicated in

**1720** and **1726** illustrated in **2302** combines the input signals and outputs a signal that includes I and Q components. Blocks **2304** and **2306** determine the power of the I and Q signals received from the combiner **2302**. The outputs of block **2304** and **2306** are summed by summing block **2308** over a plurality of chips. With the example of **2304** and **2306** are summed over 2,560 chips, which corresponds to 10 symbols of the pilot signal (CPICH). Then, the divider block **2310** determines an average of the power of the pilot signal and its interference over a particular slot to produce a signal P_{RX}.

Another portion of the structure of **2311** despreads the I and Q components received from combiner **2302** and produces despread I and Q components. Divider block **2312** then divides the signal produced by the CPICH despreader **2311** based upon a number of chips in a CPICH symbol, e.g., example 256 chips. Then, imaginary determination block **2314**, power determination block **2316**, and averaging normalizing components **2318** and **2320** produce an interference (noise) component of the pilot signal (P_{N}). Further, the real component block **2322**, averaging components **2324** and **2326**, and power determination component **2328** produce a signal P_{S }that represents the power of the pilot signal (P_{S}).

The signal P_{N }that has been normalized by multiplier block **2320** is subtracted from the signal P_{RX }by subtraction block **2330**. Then, the output of subtraction block **2330** is used to divide signal P_{S }to produce a signal representative of the pilot signal power to interference ratio. The output of divider block **2332** is normalized by multiplier block **2334** based upon the factor **1**-α. Then, summation block **2336**, slot delay block **2338**, and normalization block **2340** operate further upon the power to interference ratio using the operator α. The outputs of adder block **2336** is the pilot signal to interference ratio (CPICH_E_{C}/I_{or}). This value is used by block **2120** of **2114**.

The operations of

*Y* _{1} *=H* _{d1} *S+H* _{I1} *I+N* _{1} (Eq. 43)

*Y* _{2} *=H* _{d2} *S+H* _{I2} *I+N* _{2} (Eq. 44)

The signal variance at both sides of Equations (43) and (44) can be given as

*D*(*Y* _{1})=|*H* _{d1}|^{2} *D*(*S*)+*|H* _{I1}|^{2} *D*(*I*)+σ_{N1} ^{2}

*D*(*Y* _{2})=|*H* _{d2}|^{2} *D*(*S*)+*|H* _{I2}|^{2} *D*(*I*)+σ_{N2} ^{2}

Accordingly, the equivalent equations about the signal variance model in time domain can be derived as

*D*(*y* _{1})=Σ|*h* _{d1}|^{2} *D*(*s*)+Σ|*h* _{i1}|^{2} *D*(*i*)+σ_{n1} ^{2} (Eq. 45)

*D*(*y* _{2})=Σ|*h* _{d2}|^{2} *D*(*s*)+Σ|*h* _{i2}|^{2} *D*(*i*)+σ_{n2} ^{2} (Eq. 46)

With the assumption that the transmit power of interference is constant, D(s)=D(i)=4.0. Furthermore, the combined signal of the outputs of two frequency equalizers is given as

It is demonstrated that the interference I is suppressed very well, only noise items in Equation (47) includes interference information. Thus, from the Equations (44) and (45), the power of noise at each antenna (i.e., Pn**1** and Pn**2**) can be achieved. Also, from Equation (47), the power allocation of CPICH at the desired signal (i.e., E_{C—}CPICH/Ior) also can be calculated. In some operations, the estimated channel is updated every 2 CPICH symbols while the AGC figures are updated every slot. The noise power at dual antenna and E_{C—}CPICH/Ior, which is very important input elements for FDE-IS, is updated as is appropriate.

**2400** commences with the RF receiver, e.g., baseband processing module **22** of the RF receiver, receiving the composite signal (**2402**). The composite signal includes the first information signal, the second information signal, and a pilot signal, and may include other components as well. Operation continues with equalizing the composite signal using multiple equalizers to produce multiple equalized information signals (**2404**). In one particular embodiment where two equalizers operate upon the composite signal, the equalizing operations of Step **2404** include equalizing the first composite signal by a first equalizer to produce a first equalized information signal and equalizing the composite signal by a second equalizer to produce a second equalized information signal.

Operation continues with combining the outputs of the multiple equalizers (Step **2406**). Then, operation **2400** includes determining a signal power of the composite signal (Step **2410**). Then, operation **2400** includes determining pilot signal interference power (Step **2410**). Some of these operations **2404**-**2416** may be performed by the structure of **2410**). Operation then includes determining composite channel interference (Step **2414**). Then, operation includes determining a ratio of the composite signal interference to pilot signal interference (Step **2416**). Operation next includes determining at least one noise variance parameter based upon the ratio of composite signal interference to pilot signal interference. Then, operation may include determining a plurality of equalizer coefficients based upon a first information signal channel estimate, a second information signal channel estimate, and the noise variance of the composite signal (Step **2420**). The manner in which the operations of Step **2402**-**2418** may be determined or described further herein with reference to the structure of **2402**-**2418** may be performed by the structures previously described with reference to

In one particular embodiment of the operation **2418** of **2418**. Generally, as will be described further with reference to _{OR}/I_{OC }is determined and, based upon this ratio, a fixed diagonal loading value (noise variance) is determined that is used in subsequent operations to determine equalizer coefficients.

TABLE 1 | |||

Adaptive Fixed Diagonal Loading Values | |||

Fixed Diagonal Loading | |||

IOR/IOC range (dB) | Value | ||

less than 0 | 9000 | ||

(0–2) | 8000 | ||

(2–3) | 5000 | ||

(3–5) | 2000 | ||

(5–9) | 1000 | ||

(9–10) | 900 | ||

Greater than 10 | 800 | ||

The noise variance may include a first noise variance corresponding to the first information signal and a second noise variance corresponding to the second information signal. These first and second noise variances may be separately gained scaled using gain scaling operations. Calculations of the equalizer coefficients based upon the information signal channel estimates and the determined noise variances of Step **2420** may be performed according to techniques previously described herein using both time domain and frequency domain operations.

**2502** that receives outputs from multiple equalizers. Combiner **2502** produces I and Q components of equalized signals, e.g., first and second desired equalized information signals, each of which may a pilot signal. Power determination blocks **2504** and **2506**, summing block **2508**, and chip time accumulator (summation block) **2508** combine the power of the I and Q components of the equalized signals. The output of summation block **2508** is divided by divider block **2510** based upon a number of chips in a slot, for example **2560**. The output of divider block **2510** is an estimate of the received power of the equalized signals from combiner **2502**, Prx.

CPICH despreader block **2511** receives the output from combiners **2502** and despreads the received signals using a despreading code corresponding to the pilot signal CPICH. The output of CPICH despreader **2511** is normalized across a number of chips in the pilot signal by divider block **2512**. Imaginary component determination block **2514**, power determination block **2516**, summation block **2518** and normalization multiplier **2520** determines the pilot signal interference power Ioc. Real component determination block **2522**, summation block **2524**, normalization block **2526**, and power determination block **2528** receive the output of divider **2512** and produce the pilot signal power P_{S}.

Subtraction block **2530** receives the signal Ioc and subtracts that signal from the composite signal power Prx to produce interference power component Ior. Then, divider block **2536** receives inputs Ior and Ioc and operates upon these signals to produce the ratio Ior/Ioc, which is a ratio of the interference of an equalized composite signal to the interference of the pilot signal interference power. The ratio of the composite signal interference power to the pilot signal interference power (Ior/Ioc) is received by noise variance table **2538**. The noise variance table **2538** produces two outputs that are normalized by multiplication block **2540** and division block **2542** based upon the gain ratio G**1**/G**2**. These multiplication block **2540** and division block **2542** produce noise variance estimations Pn**1** and Pn**2**.

Divider block **2532** receives as input the pilot signal power and the composite signal interference power I_{0r }and produces a ratio of the pilot signal power to the composite signal interference. This output is filtered by IIR filter **2534** which produces the ratio of the pilot signal to interference ratio CPICH_EC/IOR.

As one of ordinary skill in the art will appreciate, the terms “operably coupled” and “communicatively coupled,” as may be used herein, include direct coupling and indirect coupling via another component, element, circuit, or module where, for indirect coupling, the intervening component, element, circuit, or module does not modify the information of a signal but may adjust its current level, voltage level, and/or power level. As one of ordinary skill in the art will also appreciate, inferred coupling (i.e., where one element is coupled to another element by inference) includes direct and indirect coupling between two elements in the same manner as “operably coupled” and “communicatively coupled.”

The present invention has also been described above with the aid of method steps illustrating the performance of specified functions and relationships thereof. The boundaries and sequence of these functional building blocks and method steps have been arbitrarily defined herein for convenience of description. Alternate boundaries and sequences can be defined so long as the specified functions and relationships are appropriately performed. Any such alternate boundaries or sequences are thus within the scope and spirit of the claimed invention.

The present invention has been described above with the aid of functional building blocks illustrating the performance of certain significant functions. The boundaries of these functional building blocks have been arbitrarily defined for convenience of description. Alternate boundaries could be defined as long as the certain significant functions are appropriately performed. Similarly, flow diagram blocks may also have been arbitrarily defined herein to illustrate certain significant functionality. To the extent used, the flow diagram block boundaries and sequence could have been defined otherwise and still perform the certain significant functionality. Such alternate definitions of both functional building blocks and flow diagram blocks and sequences are thus within the scope and spirit of the claimed invention.

One of average skill in the art will also recognize that the functional building blocks, and other illustrative blocks, modules and components herein, can be implemented as illustrated or by discrete components, application specific integrated circuits, processors executing appropriate software and the like or any combination thereof.

Moreover, although described in detail for purposes of clarity and understanding by way of the aforementioned embodiments, the present invention is not limited to such embodiments. It will be obvious to one of average skill in the art that various changes and modifications may be practiced within the spirit and scope of the invention, as limited only by the scope of the appended claims.

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Referenced by

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---|---|---|---|---|

US8965756 * | Mar 14, 2011 | Feb 24, 2015 | Adobe Systems Incorporated | Automatic equalization of coloration in speech recordings |

US20120239391 * | Mar 14, 2011 | Sep 20, 2012 | Adobe Systems Incorporated | Automatic equalization of coloration in speech recordings |

Classifications

U.S. Classification | 375/229 |

International Classification | H03H7/30 |

Cooperative Classification | H04L2025/03605, H04L2025/03426, H04L25/03006 |

European Classification | H04L25/03B |

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