|Publication number||US7901358 B2|
|Application number||US 11/592,741|
|Publication date||Mar 8, 2011|
|Filing date||Nov 2, 2006|
|Priority date||Nov 2, 2005|
|Also published as||CA2628100A1, CA2628100C, CA2935422A1, CN101351724A, CN101351724B, EP1952175A2, EP1952175B1, US20070239001, USRE46185, WO2007056104A2, WO2007056104A3, WO2007056104A9|
|Publication number||11592741, 592741, US 7901358 B2, US 7901358B2, US-B2-7901358, US7901358 B2, US7901358B2|
|Inventors||James Mehi, Ronald E. Daigle, Laurence C. Brasfield, Brian Starkoski, Jerrold Wen, Kai Wen Liu, Lauren S. Pflugrath, F. Stuart Foster, Desmond Hirson|
|Original Assignee||Visualsonics Inc., Sunnybrook Health Science Centre|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (289), Non-Patent Citations (48), Referenced by (32), Classifications (32), Legal Events (7)|
|External Links: USPTO, USPTO Assignment, Espacenet|
This application claims the benefit of U.S. Provisional Patent Application No. 60/733,091 filed Nov. 2, 2005, and the benefit of U.S. Provisional Patent application No. 60/733,089 filed Nov. 2, 2005, both of which are fully incorporated herein and made a part hereof.
Ultrasound echography systems using an arrayed transducer have been used in human clinical applications where the desired image resolution is in the order of millimeters. Operating frequencies in these clinical systems are typically below 10 MHz. With these low operating frequencies, however, such systems are not appropriate for imaging where higher resolutions are needed, for example in imaging small animals such as mice or small tissue structures in humans.
Moreover, small animal imaging applications present several challenging requirements which are not met by currently available imaging systems. The heart rate of an adult mouse may be as high as 500 beats per minute, so high frame rate capability may be desired. The width of the region being imaged, the field of view, should also be sufficient to include the entire organ being studied.
Ultrasound systems for imaging at frequencies above 15 MHz have been developed using a single element transducer. However, arrayed transducers offer better image quality, can achieve higher acquisition frame rates and offer other advantages over single element transducer systems. The embodiments according to the present invention overcome many of the challenges in the current art, including those described above.
Provided herein is a system and method for acquiring an ultrasound signal comprised of a signal processing unit adapted for acquiring a received ultrasound signal from a ultrasound transducer having a plurality of elements. The system can be adapted to receive ultrasound signals having a frequency of at least 15 megahertz (MHz) with a fixed transducer having a field of view of at least 5.0 millimeters (mm) at a frame rate of at least 20 frames per second (fps). The signal processing unit can further produce an ultrasound image from the acquired ultrasound signal. The transducer can be, but is not limited to, a linear array transducer, a phased array transducer, a two-dimensional (2-D) array transducer, or a curved array transducer. The system can include such a transducer or be adapted to operate with such a transducer.
Also provided herein is a system and method for acquiring an ultrasound signal comprising a processing unit for acquiring received ultrasound signals from an ultrasound transducer operating at a transmit and receive frequency of at least 15 MHz, wherein the processing unit comprises a signal sampler that uses quadrature sampling to acquire the ultrasound signal.
Additional advantages of the invention will be set forth in part in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. The advantages of the invention will be realized and attained by means of the elements and combinations particularly pointed out in the appended claims. It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory only and are not restrictive of the invention, as claimed.
The accompanying drawings, which are incorporated in and constitute a part of this specification, illustrate several embodiments according to the invention and together with the description, serve to explain the principles of the invention:
The present invention may be understood more readily by reference to the following detailed description of the invention and the Examples included therein and to the Figures and their previous and following description.
Before the present compounds, compositions, articles, devices, and/or methods are disclosed and described, it is to be understood that this invention is not limited to specific methods, specific components, or to particular computer architecture, as such may, of course, vary. It is also to be understood that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting. As used in the specification and the appended claims, the singular forms “a,” “an” and “the” include plural referents unless the context clearly dictates otherwise. Thus, for example, reference to “a processing unit,” or to “a receive channel” includes two or more such processing units or receive channels, and the like.
Ranges may be expressed herein as from “about” one particular value, and/or to “about” another particular value. When such a range is expressed, another embodiment includes from the one particular value and/or to the other particular value. Similarly, when values are expressed as approximations, by use of the antecedent “about,” it will be understood that the particular value forms another embodiment. It will be further understood that the endpoints of each of the ranges are significant both in relation to the other endpoint, and independently of the other endpoint.
“Optional” or “optionally” means that the subsequently described event or circumstance may or may not occur, and that the description includes instances where said event or circumstance occurs and instances where it does not.
Aspects of the exemplary systems disclosed herein can be implemented via a general-purpose computing device such as one in the form of a computer 101 shown in
The system bus 113 represents one or more of several possible types of bus structures, including a memory bus or memory controller, a peripheral bus, an accelerated graphics port, and a processor or local bus using any of a variety of bus architectures. By way of example, such architectures can include an Industry Standard Architecture (ISA) bus, a Micro Channel Architecture (MCA) bus, an Enhanced ISA (EISA) bus, a Video Electronics Standards Association (VESA) local bus, and a Peripheral Component Interconnects (PCI) bus also known as a Mezzanine bus. This bus, and all buses specified in this description can also be implemented over a wired or wireless network connection. The bus 113, and all buses specified in this description can also be implemented over a wired or wireless network connection and each of the subsystems, including the processor 103, a mass storage device 104, an operating system 105, application software 106, data 107, a network adapter 108, system memory 112, an Input/Output Interface 110, a display adapter 109, a display device 111, and a human machine interface 102, can be contained within one or more remote computing devices 114 a,b,c at physically separate locations, connected through buses of this form, in effect implementing a fully distributed system.
The computer 101 typically includes a variety of computer readable media. Such media can be any available media that is accessible by the computer 101 and includes both volatile and non-volatile media, removable and non-removable media. The system memory 112 includes computer readable media in the form of volatile memory, such as random access memory (RAM), and/or non-volatile memory, such as read only memory (ROM). The system memory 112 typically contains data such as data 107 and/or program modules such as operating system 105 and application software 106 that are immediately accessible to and/or are presently operated on by the processing unit 103.
The computer 101 may also include other removable/non-removable, volatile/non-volatile computer storage media. By way of example,
Any number of program modules can be stored on the mass storage device 104, including by way of example, an operating system 105 and application software 106. Each of the operating system 105 and application software 106 (or some combination thereof) may include elements of the programming and the application software 106. Data 107 can also be stored on the mass storage device 104. Data 104 can be stored in any of one or more databases known in the art. Examples of such databases include, DB2®, Microsoft® Access, Microsoft® SQL Server, Oracle®, mySQL, PostgreSQL, and the like. The databases can be centralized or distributed across multiple systems.
A user can enter commands and information into the computer 101 via an input device (not shown). Examples of such input devices include, but are not limited to, a keyboard, pointing device (e.g., a “mouse”), a microphone, a joy stick, a serial port, a scanner, and the like. These and other input devices can be connected to the processing unit 103 via a human machine interface 102 that is coupled to the system bus 113, but may be connected by other interface and bus structures, such as a parallel port, game port, or a universal serial bus (USB). In an exemplary system of an embodiment according to the present invention, the user interface can be chosen from one or more of the input devices listed above. Optionally, the user interface can also include various control devices such as toggle switches, sliders, variable resistors and other user interface devices known in the art. The user interface can be connected to the processing unit 103. It can also be connected to other functional blocks of the exemplary system described herein in conjunction with or without connection with the processing unit 103 connections described herein.
A display device 111 can also be connected to the system bus 113 via an interface, such as a display adapter 109. For example, a display device can be a monitor or an LCD (Liquid Crystal Display). In addition to the display device 111, other output peripheral devices can include components such as speakers (not shown) and a printer (not shown) which can be connected to the computer 101 via Input/Output Interface 110.
The computer 101 can operate in a networked environment using logical connections to one or more remote computing devices 114 a,b,c. By way of example, a remote computing device can be a personal computer, portable computer, a server, a router, a network computer, a peer device or other common network node, and so on. Logical connections between the computer 101 and a remote computing device 114 a,b,c can be made via a local area network (LAN) and a general wide area network (WAN). Such network connections can be through a network adapter 108. A network adapter 108 can be implemented in both wired and wireless environments. Such networking environments are commonplace in offices, enterprise-wide computer networks, intranets, and the Internet 115. The remote computer 114 a,b,c may be a server, a router, a peer device or other common network node, and typically includes all or many of the elements already described for the computer 101. In a networked environment, program modules and data may be stored on the remote computer 114 a,b,c. The logical connections include a LAN and a WAN. Other connection methods may be used, and networks may include such things as the “world wide web” or Internet.
For purposes of illustration, application programs and other executable program components such as the operating system 105 are illustrated herein as discrete blocks, although it is recognized that such programs and components reside at various times in different storage components of the computing device 101, and are executed by the data processor(s) of the computer. An implementation of application software 106 may be stored on or transmitted across some form of computer readable media. Computer readable media can be any available media that can be accessed by a computer. By way of example, and not limitation, computer readable media may comprise “computer storage media” and “communications media.” Computer storage media include volatile and non-volatile, removable and non-removable media implemented in any method or technology for storage of information such as computer readable instructions, data structures, program modules, or other data. Computer storage media includes, but is not limited to, RAM, ROM, EEPROM, flash memory or other memory technology, CD-ROM, digital versatile disks (DVD) or other optical storage, magnetic cassettes, magnetic tape, magnetic disk storage or other magnetic storage devices, or any other medium which can be used to store the desired information and which can be accessed by a computer. An implementation of the disclosed method may be stored on or transmitted across some form of computer readable media.
The processing of the disclosed method can be performed by software components. The disclosed method may be described in the general context of computer-executable instructions, such as program modules, being executed by one or more computers or other devices. Generally, program modules include computer code, routines, programs, objects, components, data structures, etc. that perform particular tasks or implement particular abstract data types. The disclosed method may also be practiced in grid-based and distributed computing environments where tasks are performed by remote processing devices that are linked through a communications network. In a distributed computing environment, program modules may be located in both local and remote computer storage media including memory storage devices.
Aspects of the exemplary systems shown in the Figures and described herein, can be implemented in various forms including hardware, software, and a combination thereof. The hardware implementation can include any or a combination of the following technologies, which are all well known in the art: discrete electronic components, a discrete logic circuit(s) having logic gates for implementing logic functions upon data signals, an application specific integrated circuit having appropriate logic gates, a programmable gate array(s) (PGA), field programmable gate array(s) (FPGA), etc. The software comprises an ordered listing of executable instructions for implementing logical functions, and can be embodied in any computer-readable medium for use by or in connection with an instruction execution system, apparatus, or device, such as a computer-based system, processor-containing system, or other system that can fetch the instructions from the instruction execution system, apparatus, or device and execute the instructions.
Aspects of the exemplary systems can be implemented in computerized systems. Aspects of the exemplary systems, including for instance the computing unit 101, can be operational with numerous other general purpose or special purpose computing system environments or configurations. Examples of well known computing systems, environments, and/or configurations that may be suitable for use with the system and method include, but are not limited to, personal computers, server computers, laptop devices, and multiprocessor systems. Additional examples include set top boxes, programmable consumer electronics, network PCs, minicomputers, mainframe computers, distributed computing environments that include any of the above systems or devices, and the like.
Aspects of the exemplary systems can be described in the general context of computer instructions, such as program modules, being executed by a computer. Generally, program modules include routines, programs, objects, components, data structures, etc. that perform particular tasks or implement particular abstract data types. The system and method may also be practiced in distributed computing environments where tasks are performed by remote processing devices that are linked through a communications network. In a distributed computing environment, program modules may be located in both local and remote computer storage media including memory storage devices.
Among many possible applications, the described embodiments enable in vivo visualization, assessment, and measurement of anatomical structures and hemodynamic function in longitudinal imaging studies of small animals. The systems can provide images having very high resolution, image uniformity, depth of field, adjustable transmit focal depths, multiple transmit focal zones for multiple uses. For example, the ultrasound image can be of a subject or an anatomical portion thereof, such as a heart or a heart valve. The image can also be of blood and can be used for applications including evaluation of the vascularization of tumors. The systems can be used to guide needle injections.
The described embodiments can also be used for human clinical, medical, manufacturing (e.g., ultrasonic inspections, etc.) or other applications where producing an image at a transmit frequency of 15 MHz or higher is desired.
Embodiments according to the described systems can comprise one or more of the following, which are described in greater detail herein: an array transducer that can be operatively connected to a processing system that may be comprised of one or more of signal and image processing capabilities; digital transmit and receive beamformer subsystems; analog front end electronics; a digital beamformer controller subsystem; a high voltage subsystem; a computer module; a power supply module; a user interface; software to run the beamformer; a scan converter, and other system features as described herein.
An arrayed transducer used in the system can be incorporated into a scanhead that, in one embodiment, may be attached to a fixture during imaging which allows the operator to acquire images free of the vibrations and shaking that usually result from “free hand” imaging. A small animal subject may also be positioned on a heated platform with access to anesthetic equipment, and a means to position the scanhead relative to the subject in a flexible manner. The scanhead can be attached to a fixture during imaging. The fixture can have various features, such as freedom of motion in three dimensions, rotational freedom, a quick release mechanism, etc. The fixture can be part of a “rail system” apparatus, and can integrate with the heated mouse platform.
The systems can be used with platforms and apparatus used in imaging small animals including “rail guide” type platforms with maneuverable probe holder apparatuses. For example, the described systems can be used with multi-rail imaging systems, and with small animal mount assemblies as described in U.S. patent application Ser. No. 10/683,168, entitled “Integrated Multi-Rail Imaging System,” U.S. patent application Ser. No. 10/053,748, entitled “Integrated Multi-Rail Imaging System,” U.S. patent application Ser. No. 10/683,870, now U.S. Pat. No. 6,851,392, issued Feb. 8, 2005, entitled “Small Animal Mount Assembly,” and U.S. patent application Ser. No. 11/053,653, entitled “Small Animal Mount Assembly,” which are each fully incorporated herein by reference.
Small animals can be anesthetized during imaging and vital physiological parameters such as heart rate and temperature can be monitored. Thus, an embodiment of the system may include means for acquiring ECG and temperature signals for processing and display. An embodiment of the system may also display physiological waveforms such as an ECG, respiration or blood pressure waveform.
Provided herein are embodiments of a system for acquiring ultrasound signals comprising a signal processing unit adapted for acquiring a received ultrasound signal from an ultrasound transducer having a plurality of elements. The system can be adapted to receive ultrasound signals having a frequency of at least 15 megahertz (MHz) with a transducer having a field of view of at least 5.0 millimeters (mm) at a frame rate of at least 20 frames per second (fps). In other embodiments, the ultrasound signals can be acquired at an acquisition rate of 50, 100, or 200 (fps). Optionally, ultrasound signals can be acquired at an acquisition rate of 200 frames per second (fps) or higher. In other examples, the received ultrasound signals can be acquired at a frame rate within the range of about 100 fps to about 200 fps. In some exemplary aspects, the length of the transducer is equal to the field of view. The field of view can be wide enough to include organs of interest such as the small animal heart and surrounding tissue for cardiology, and full length embryos for abdominal imaging. In one embodiment, the two-way bandwidth of the transducer can be approximately 50% to 100%. Optionally, the two-way bandwidth of the transducer can be approximately 60% to 70%. Two-way bandwidth refers to the bandwidth of the transducer that results when the transducer is used both as a transmitter of ultrasound and a receiver—that is, the two-way bandwidth is the bandwidth of the one-way spectrum squared.
The processing unit produces an ultrasound image from the acquired ultrasound signal(s). The acquired signals may be processed to generate an ultrasound image at display rate that is slower than the acquisition rate. Optionally, the generated ultrasound image can have a display rate of 100 fps or less. For example, the generated ultrasound image has a display rate of 30 fps or less. The field of view can range from about 2.0 mm to about 30.0 mm. When a smaller field of view is utilized, the processing unit can acquire the received ultrasound signals at an acquisition rate of at least 300 frames per second (fps). In other examples, the acquisition rate can be 50, 100, 200 or more frames per second (fps).
In one embodiment, in which a 30 MHz center frequency transducer is used, the image generated using the disclosed systems may have a lateral resolution of about 150 microns (μm) or less and an axial resolution of about 75 microns (μm) or less. For example, the image can have an axial resolution of about 30 microns (μm). Furthermore, embodiments according to the present invention transmit ultrasound that may be focused at a depth of about 1.0 mm to about 30.0 mm. For example, the transmitted ultrasound can be focused at a depth of about 3.0 mm to about 10.0 mm. In other examples, the transmitted ultrasound can be focused at a depth of about 2.0 mm to about 12.0 mm, of about 1.0 mm to about 6.0 mm, of about 3.0 mm to about 8.0 mm, or of about 5.0 mm to about 30.0 mm.
In various embodiments, the transducer can be, but is not limited to, a linear array transducer, a phased array transducer, a two-dimensional (2-D) array transducer, or a curved array transducer. A linear array is typically flat, i.e., all of the elements lie in the same (flat) plane. A curved linear array is typically configured such that the elements lie in a curved plane. The transducers described herein are “fixed” transducers. The term “fixed” means that the transducer array does not utilize movement in its azimuthal direction during transmission or receipt of ultrasound in order to achieve its desired operating parameters, or to acquire a frame of ultrasound data. Moreover, if the transducer is located in a scanhead or other imaging probe, the term “fixed” may also mean that the transducer is not moved in an azimuthal or longitudinal direction relative to the scan head, probe, or portions thereof during operation. The described transducers, which are fixed as described, are referred to throughout as an “array,” a “transducer,” an “ultrasound transducer,” an “ultrasound array,” an “array transducer,” an “arrayed transducer,” an “ultrasonic transducer” or combinations of these terms, or by other terms which would be recognized by those skilled in the art as referring to an ultrasound transducer. The transducers as described herein can be moved between the acquisition of ultrasound frames, for example, the transducer can be moved between scan planes after acquiring a frame of ultrasound data, but such movement is not required for their operation. As one skilled in the art would appreciate however, the transducer of the present system can be moved relative to the object imaged while still remaining fixed as to the operating parameters. For example, the transducer can be moved relative to the subject during operation to change position of the scan plane or to obtain different views of the subject or its underlying anatomy.
Arrayed transducers are comprised of a number of elements. In one embodiment, the transducer used to practice one or more aspects of the present invention comprises at least 64 elements. In one aspect, the transducer comprises 256 elements. The transducer can also comprise fewer or more than 256 elements. The transducer elements can be separated by a distance equal to about one-half the wavelength to about two times the wavelength of the center transmit frequency of the transducer (referred to herein as the “element pitch.”). In one aspect, the transducer elements are separated by a distance equal to about the wavelength of the center transmit frequency of the transducer. Optionally, the center transmit frequency of the transducer used is equal to or greater than 15 MHz. For example, the center transmit frequency can be approximately 15 MHz, 20 MHz, 30 MHz, 40 MHz, 50 MHz, 55 MHz or higher. In some exemplary aspects, the ultrasound transducer can transmit ultrasound into the subject at a center frequency within the range of about 15 MHz to about 80 MHz. In one embodiment according to the present invention, the transducer has a center operating frequency of at least 15 MHz and the transducer has an element pitch equal to or less than 2.0 times the wavelength of sound at the transducer's transmitted center frequency. The transducer can also have an element pitch equal to or less than 1.5 times the wavelength of sound at the transducers transmitted center frequency.
By non-limiting example, one transducer that may be used with the described system can be, among others, an arrayed transducer as described in U.S. patent application Ser. No. 11/109,986, entitled “Arrayed Ultrasonic Transducer,” filed Apr. 20, 2005 and published on Dec. 8, 2005 as U.S. Patent Application Publication No.: US 2005/0272183 A1, which is fully incorporated herein by reference and made a part hereof. The transducer may also comprise an array of piezoelectric elements which can be electronically steered using variable pulsing and delay mechanisms. The processing system according to various embodiments of the present invention may include multiple transducer ports for the interface of one or more transducers or scanheads. As previously described, a scanhead can be hand held or mounted to rail system and the scanhead cable can be flexible.
Whether the system includes a transducer, or is adapted to be used with a separately acquired transducer, each element of the transducer can be operatively connected to a receive channel of a processing unit. Optionally, the number of transducer elements is greater than the number of receive channels. For example, the transducer may comprise at least 64 elements that are operatively connected to at least 32 receive channels. In one aspect, 256 elements are operatively connected to 64 receive channels. In another aspect, 256 elements are operatively I connected to 128 receive channels. In yet another aspect, 256 elements are operatively connected to 256 receive channels. Each element can also be operatively connected to a transmit channel.
The system can further comprise one or more signal samplers for each receive channel. The signal samplers can be analog-to-digital converters (ADCs). The signal samplers can use direct sampling techniques to sample the received signals. Optionally, the signal samplers can use bandwidth sampling to sample the received signals. In another aspect, the signal samplers can use quadrature sampling to sample the received signals. Optionally, with quadrature sampling, the signal samplers comprise sampling clocks shifted 90 degrees out of phase. Also with quadrature sampling the sampling clocks also have a receive period, and the receive clock frequency can be approximately equal to the center frequency of a received ultrasound signal but may be different from the transmit frequency. For example, in many situations, the center frequency of the received signal has been shifted lower than the center frequency of the transmit signal due to frequency dependent attenuation in the tissue being imaged. For these situations the receive sample clock frequency can be lower than the transmit frequency.
An acquired signal can be processed using an interpolation filtration method. Using the interpolation filtration method a delay resolution can be used, which can be less than the receive clock period. In an exemplary aspect, the delay resolution can be, for example, 1/16 of the receive clock period.
The processing unit can comprise a receive beamformer. The receive beamformer can be implemented using at least one field programmable gate array (FPGA) device. The processing unit can also comprise a transmit beamformer. The transmit beamformer can also be implemented using at least one FPGA device.
In one aspect, 512 lines of ultrasound are generated, transmitted into the subject and received from the subject for each frame of the generated ultrasound image. In a further aspect, 256 lines of ultrasound can also be generated, transmitted into the subject and received from the subject for each frame of the generated ultrasound image. In another aspect, at least two lines of ultrasound can be generated, transmitted into the subject and received from the subject at each element of the array for each frame of the generated ultrasound image. Optionally, one line of ultrasound is generated, transmitted into the subject and received from the subject at each element of the array for each frame of the generated ultrasound image.
The ultrasound systems described herein can be used in multiple imaging modes. For example, the systems can be used to produce an image in B-mode, M-mode, Pulsed Wave (PW) Doppler mode, power Doppler mode, color flow Doppler mode, RF-mode and 3-D mode. The systems can be used in Color Flow Imaging modes, including directional velocity color flow, Power Doppler imaging and Tissue Doppler imaging. The systems can also be used with Steered PW Doppler, with very high pulse repetition frequencies (PRF). The systems can also be used in M-Mode, with simultaneous B-Mode, for cardiology or other applications where such techniques are desired. The system can optionally be used in Duplex and Triplex modes, in which M-Mode and PW Doppler and/or Color Flow modes run simultaneously with B-Mode in real-time. A 3-D mode in which B-Mode or Color Flow mode information is acquired over a 3-dimensional region and presented in a 3-D surface rendered display can also be used. A line based image reconstruction or “EKV” mode, can be used for cardiology or other applications, in which image information is acquired over several cardiac cycles and recombined to provide a very high frame rate display. Line based image reconstruction methods are described in U.S. patent application Ser. No. 10/736,232, now U.S. Pat. No. 7,052,460 issued May 30, 2006 and entitled “System for Producing an Ultrasound Image Using Line Based Image Reconstruction,” which is incorporated fully herein by reference and made a part hereof. Such line based imaging methods image can be incorporated to produce an image when a high frame acquisition rate is desirable, for example when imaging a rapidly beating mouse heart. In the RF acquisition mode, raw RF data can be acquired, displayed and made available for off-line analysis.
In one embodiment, the transducer can transmit at a pulse repetition frequency (PRF) of at least 500 hertz (Hz). The system can further comprise a processing unit for generating a color flow Doppler ultrasound image from the received ultrasound. Optionally, the PRF is between about 100 Hz to about 150 KHz. In M-Mode or RF Mode the PRF is between about 100 Hz and about 10 KHz. For Doppler modes, the PRF can be between about 500 Hz and about 150 KHz. For M-Mode and RF mode, the PRF can be between about 50 Hz and about 10 KHz.
Exemplary Arrayed Transducer
Referring now to
Referring now in particular to
The interposer can further comprise a dielectric layer disposed thereon a portion of the top surface of the interposer about the central opening of the piece. In this aspect, and referring also to
Referring now to
The circuit board further comprise a plurality of board electrical traces formed thereon the top surface of the Kapton™ layer, each board electrical trace having a proximal end adapted to couple to an electrical trace of the transducer and a distal end adapted to couple to a connector, such as, for example, a cable for communication of signals therethrough. In one aspect, the length of the circuit forming each electrical trace has a substantially constant impedance.
The circuit board also comprises a plurality of vias that pass though the Kapton™ layer and are in communication with the underlying ground layer so that signal return paths, or signal ground paths, can be formed. Further, the circuit board comprises a plurality of ground pins. Each ground pin has a proximal end that is coupled to the ground layer of the circuit board (passing through one of the vias in the Kapton layer) and a distal end that is adapted to couple to the connector.
Referring now to
In one aspect, the gold ball bumps are applied directly onto the circuit board. Each ball bump is positioned in communication with one electrical trace of the circuit board. When the PZT stack is applied, it is aligned with the electrical traces of the circuit board and electrical continuity is made via the ball bumps. The PZT stack is secured to the circuit board by, for example and not meant to be limiting, a) use of an underfill, such as a UV curable; b) use of an ACF tape; c) by electroplating pure Indium solder onto the electrodes of either the PZT or the circuit board and reflowing the Indium to provide a solder joint between the signal electrode on the PZT and the gold ball bump on the circuit board, and the like.
An arrayed transducer can be operatively connected to the processing unit of the system using the flex circuit as shown in
The connection can comprise a cable or bundle of cables. The cable can connect each element of the array to the processing unit in a one-to-one relationship; that is, each element can be electrically connected with its own signal and a ground lead to a designated connection point in the processing unit whereby the plurality of individual element connections are bundled together to form the overall cable. Optionally, each individual electrical connection can be unbundled and not physically formed into a cable or cable assembly.
Suitable cables can be coaxial cables, twisted pairs, and copper alloy wiring. Other connection means can be via non-physically connected methods such as RF links, infrared links, and similar technologies where appropriate transmitting and receiving components are included.
The individual element connections can comprise coaxial cable of a type typically used for connection array elements to processing units. These coaxial cables can be of a low loss type. The coaxial cables typically comprise a center conductor and some type of outer shielding insulated from the center conductor and encased in an outer layer of insulation. These coaxial cables can have nominal impedances appropriate for use with an array. Example nominal impedances can be 50 ohms or more, including 50 ohms, 52 ohms, 73 ohms, 75 ohms or 80 ohms.
An exemplary medical cable for use with one or more of the ultrasound imaging systems described herein comprises a minimum of 256 coaxial cables of 40 AWG with a nominal impedance of about 75 ohms with coaxial cable lengths of about 2.0 m. The length can be less than 2.0 m or greater than 2.0 m. The medical cable jacket length can accommodate the cable length, can include additional metal sheaths for electrical shielding and can be made of PVC or other flexible materials.
Cables and the connections for connecting an array transducer to the processing unit, including those described herein can be fabricated by companies such as Precision Interconnect-Tyco Electronics (Tyco Electronics Corporation, Wilmington, Del.).
The exemplary cable, at the proximal end, can further comprise of flex/strain relief, 12 PCBs interfacing between the coaxial cables and the ZIF™ pins, a 360 Pin ITT Cannon ZIF™ connector and actuation handle (DLM6-360 type), and a shielded casing around the connector. The exemplary cable, at the distal end, can comprise of a flex/strain relief cable terminated to two PCBs, interfacing between the coaxial cables and the flex circuit board, wherein each PCB has 1 BSH-090-01-L-D-A Samtec Connector (Samtec, Inc., New Albany, Ind.) and each PCB has 75 Ohm characteristic impedance traces with cables terminated from both sides of the PCB in a staggered layout.
The cable can use a “flex circuit” method of securing and connecting a plurality of coax cables which comprise the large cable. In an exemplary embodiment, the array has 256-elements. The array is mounted in the central region of a flex circuit. The flex circuit has two ends such that the odd numbered elements 1,3,5,7 . . . 255 are terminated on the left end of the flex with a BTH-090 connector labeled J1, and that the even numbered elements 2,4,6,8 . . . 256 are terminated on the right end of the flex with a BTH-090 connector labeled J3. For both ends, the elements are terminated in sequence along the upper and bottom rows of their respective connectors with GND (signal return) pins evenly dispersed across the connector in a repeated pattern.
The repeat pattern is defined from the outer edge of the flex towards the central region of the flex and is as follows:
2 signal pins, GND
3 signal pins, GND
2 signal pins, GND
3 signal pins . . .
3 signal pins, GND
2 signal pins, GND
2 signal pins, GND.
A schematic showing a side view of the folded flex circuit, with the array mounted in the central array of the flex is shown in
The flex circuit can be connected to the exemplary cable described above. The flex circuit can be connected to a Precision Interconnect-Tyco Electronics medical cable assembly. The electrical, for example, connection from the flex to the ZIF™ connector can be made through two scanhead PCBs followed by a coax cable bundle and 12 short PCBs each with a 2×15 connector inserted into ZIF™ pins.
Each scanhead PCB (total of two) can comprise one BSH-090 connector, 128 traces (all traces with controlled impedance of for example 75 Ohms at 30 MHz) and can be terminated with 128 (40 AWG 75 Ohm) coax cables. The PCB can have outer dimensions of 0.525″ by 2.344.″
An exemplary medical cable, as partially shown in
The 12 individual PCBs used to connect to the ZIF connector have coax cables connected on one or both sides of the board. One edge of the PCB can have a connector suitable for insertion into the ZIF connector (Samtec SSW or equivalent) and each PCB shall have the appropriate traces and vias required to connect the correct coaxial cable to the correct ZIF pin. Each PCB can have a Samtec SSW, or equivalent, connector with two rows of 15 pins, although the number of coax cables may differ on some of the 12 PCBs as defined in the
One of the 12 PCBs requires provisions in the trace layout to include an EEPROM as defined in
Various connection methods can be used including connectors of various styles. For these various connection methods, the impedance can be 75 Ohms at a center frequency of 30 MHz.
TABLE 1 The layout of connections on the connector end of the ZIF PCB that plugs into the ITT Connector. General Pattern Signal Signal Signal GND Signal Signal GND Signal Signal Signal Signal Signal Signal GND Signal Signal GND Signal Signal Signal Signal Signal Signal GND Signal Signal GND Signal Signal Signal
An exemplary embodiment of an ultrasound system 1600 according to the present invention is shown in
The exemplary system 1600 comprises an array transducer 1601, a cable 1619, and a processing unit 1620. The cable 1619 connects the processing unit 1620 and the array transducer 1601. The processing unit may comprise software and hardware components. The processing unit can comprise one or more of a multiplexer(MUX)/front end electronics 1602, a receive beamformer 1603, a beamformer control 1604, a transmit beamformer 1605, a system control 1606, a user interface 1607, a scan converter 1608, a video processing display unit 1609, and processing modules including one or more of a M-mode processing module (not shown), a PW Doppler processing module 1611, a B-mode processing module 1612, a color flow processing module 1613, a 3-D mode processing module (not shown), and a RF mode processing module 1615. The center frequency range of the exemplary system can be about 15-55 MHz or higher. When measured from the outside edge of the bandwidths, the frequency range of the exemplary system can be about 10-80 MHz or higher.
The array transducer 1601 interfaces with the processing unit 1620 at the MUX/front end electronics (MUX/FEE) 1602. The MUX portion of the MUX/FEE 1602 is a multiplexer which can electronically switch or connect a plurality of electrical paths to a lesser number of electrical paths. The array transducer 1601 converts electrical energy to ultrasound energy and vice versa and is electrically connected to the MUX/FEE 1602.
The MUX/FEE 1602 comprises electronics which generate a transmit waveform which is connected to a certain subset of the elements of the array, namely the elements of the active aperture. The subset of elements is called the active aperture of the array transducer 1601. The electronics of the MUX/FEE 1602 also connects the active aperture of the array to the receive channel electronics. During operation, the active aperture moves about the array transducer 1601, in a manner determined by components described herein.
The MUX/FEE 1602 switchably connects the elements of the active aperture to transmit and receive channels of the exemplary system. In an exemplary 256-element array transducer embodiment of the invention, there are 64 transmit channels and 64 receive channels that can be switchably connected to the active aperture of up to 64 elements. The up to 64 elements of the active aperture are contiguous. In certain embodiments of the invention, there is a separate transmit MUX and a separate receive MUX. Other embodiments of the invention share the MUX for both the transmit channels and the receive channels.
During a transmit cycle of the exemplary ultrasound system 1600, the front end electronics portion of the MUX/FEE 1602 supply a high voltage signal to the elements of the active aperture of the array transducer 1601. In one aspect, the front end electronics can also provide protection circuitry for the receiver channels to protect them from the high voltage transmit signal, as the receive channels and the transmit channels have a common connection point at the elements of the array transducer 1601. The protection can be in the form of isolation circuitry which limits the amount of transmit signal that can leak or pass into the receive channel to a safe level which will not cause damage to the receive electronics. Characteristics of the MUX/FEE 1602 include a fast rise time on the transmit side, and high bandwidth on the transmit and receive channels.
The MUX/FEE 1602 passes signals from the transmit beamformer 1605 to the array transducer 1601. In an exemplary embodiment, the transmit beamformer 1605 generates and supplies separate waveforms to each of the elements of the active aperture. In an exemplary embodiment, the waveform for each element of the active aperture is the same. In another aspect, the waveforms for each element of the active aperture are not all the same and in some embodiments have differing center frequencies.
In one exemplary embodiment, each separate transmit waveform has a delay associated with it. The distribution of the delays for each element's waveform is called a delay profile. The delay profile is calculated in a way to cause the desired focusing of the transmit acoustic beam to the desired focal point. In certain embodiments, the transmit acoustic beam axis is perpendicular to the plane of the array 1601, and the beam axis intersects the array 1601 at the center of the active aperture of the array transducer 1601. The delay profile can also steer the beam so that it is not perpendicular to the plane of the array 1601. In an exemplary aspect of the present invention, a delay resolution of 1/16 can be used. Or, in other words, 1/16 of the period of the center frequency of the transmit center frequency, though other delay resolutions are contemplated within the scope of this invention. For example at a 50 MHz center frequency, the period is 20 nanoseconds, so 1/16 of that period is 1.25 nanoseconds, which is the exemplary delay resolution used to focus the acoustic beam. It is to be appreciated that the delay resolution may be different than 1/16th of a period, for example delay resolutions less than 1/16th (e.g., 1/24, 1/32, etc) as well as delay resolutions greater than 1/16 (e.g., 1/12, ⅛, etc.) are contemplated within the scope of this invention.
The receive beamformer 1603, can also be connected to elements of the active aperture of the array transducer 101 via the MUX/FEE 1602. During transmit an acoustic signal penetrates into the subject and generates a reflected signal from the tissues of the subject. The reflected signal is received by the elements of the active aperture of the array transducer 1601 and converted into an analog electrical signal emanating from each element of the active aperture. The electrical signal is sampled to convert it from an analog to a digital signal in the receive beamformer 1603. Embodiments of the invention use quadrature sampling for digitization of the received signal. During the receive cycle of the system 1600, the array transducer 1601 also has a receive aperture that is determined by the beamformer control 1604, which tells the receive beamformer 1603 which elements of the array to include in the active aperture and what delay profile to use. The receive beamformer 1603 of the exemplary embodiment is a digital beamformer.
The receive beamformer 1603 introduces delays into the received signal of each element of the active aperture. The delays are collectively called the delay profile. The receive delay profile can be dynamically adjusted based on time-of-flight—that is, the length of time that has elapsed during the transmission of the ultrasound into the tissue being imaged. The time-of-flight is used to focus the receive beamformer to a point of focus within the tissue. In other words, the depth of the receive beam is adjusted using a delay profile which incorporates information pertaining to the time-of-flight of the transmitted beam.
The received signal from each element of the active aperture is summed wherein the sum incorporates the delay profile. The summed received signal flows along the receive channel from the receive beamformer 1603 to one or more of the processing module(s) 1611, 1612, 1613, and/or 1615, including those not shown in
The beam former control 1604 is connected to the MUX/FEE 1602 through the transmit beamformer 1605 and the receive beamformer 1603. It is also connected to the system control 1606. The beamformer control 1604 provides information to the MUX/FEE 1602 so that the desired elements of the array transducer 1601 are connected to form the active aperture. The beamformer control 1604 also creates and sends to the receive beamformer 1603 the delay profile for use with the reception of a particular beam. In embodiments of the invention, the receive delay profile can be updated repeatedly based upon the time of flight. The beamformer control 1604 also creates and sends to the transmit beamformer 1605 the transmit delay profile.
The system control 1606 operates in a manner known to one of ordinary skill in the art. It takes input from the user interface 1607 and provides the control information to the various components of the system 1600 in order to configure the system 1600 for a chosen mode of operation. The scan converter 1608 operates in a manner known in the art and takes the raw image data generated from the one or more of the processing modules and converts the raw image data into an image that can be displayed by the video processing/display 1609. For some processing modes of operation, the image can be displayed without using the scan converter 1608 if the video characteristics of the image are the same as those of the display.
The processing modules, except as noted herein, function in a manner known to one of ordinary skill in the art. For the PW Doppler module 1611 and the color flow processing module 1613, the pulse repetition frequency (PRF) can be high due to the high center frequencies of embodiments of this invention. The maximum unaliased velocities which may be measured are proportional to the PRF and inversely proportional to the transmit center frequency. The PRFs required to allow for the unaliased measurement of specific velocities given specific transmit center frequencies may be calculated in a method known to one of ordinary skill in the art. Given that the transmit center frequencies used are in the range of 15 to 55 MHz, or higher, and the blood flow velocities can be as high as 1 m/s and in some cases greater than 1 m/s unaliased measurement of the Doppler signal resulting from those velocities will require the PRF for PW Doppler to be up to 150 KHz. Embodiments of the invention have a PW Doppler mode which supports PRFs up to 150 KHz, which for a center frequency of 30 MHz allows for unaliased measurement of blood velocities up to 1.9 m/s in mice with a zero degree angle between the velocity vector of the moving target and the ultrasound beam axis.
In certain embodiments, the RF module 1615 uses interpolation. If the sampling method used is quadrature sampling, then the RF signal may be reconstructed from the quadrature baseband samples by zero padding and filtering, as would be known to one of ordinary skill in the art. If Nyquist sampling is used, then no reconstruction is required since the RF signal is sampled directly. In certain embodiments, the RF module 1615 reconstructs the RF signal from the quadrature samples of the receive beamformer output. The sampling takes place at the center frequency of the receive signal, but in quadrature, giving a baseband quadrature representation of the signal. The RF signal is created by first zero padding the quadrature sampled data stream, with the number of zeros determined by the desired interpolated signal sample rate. Then, a complex bandpass filter is applied to the zero padded data stream, which rejects the frequency content of the zero padded signal that is outside the frequency band from fs/2 to 3 fs/2, where fs is the sample frequency. The result after filtering is a complex representation of the original RF signal. The RF signal is then passed on to the main computer unit for further processing such as digital filtering and envelope detection and display. The real part or the complex representation of the RF signal may be displayed. For example, the RF data acquired for a particular scan line may be processed and displayed. Alternatively, RF data from a certain scan line averaged over a number of pulse echo returns can be displayed, or RF data acquired from a number of different scan lines can be averaged and displayed. The scan lines to be used for acquisition of the RF data can be specified by the user based on evaluation of the B-Mode image, by placing cursor lines overlaid on the B-Mode image. A Fast Fourier Transform (FFT) of the RF data can also be calculated and displayed. The acquisition of RF data and the acquisition of B-Mode data can be interleaved so as to allow for the display of information from both modes concurrently in real time. The acquisition of physiological signals such as the ECG signal can also occur concurrently with the acquisition of RF data. The ECG waveform can be displayed while the RF data is acquired. The timing of the acquisition of RF data can be synchronized with user defined points within the ECG waveform, thereby allowing for the RF data to be referenced to specific times during a cardiac cycle. The RF data can be stored for processing and evaluation at a later time.
In one embodiment, as shown in
The embodiment of the system of
As another example of mode-dependent interface controls, a color flow imaging user interface may have image format controls that may include color flow mode select (e.g., color flow velocity, Power Doppler, Tissue Doppler); trackball; steering angle; color box position/size select (after selection trackball is used to adjust position or size); preset recall; preset menu and invert color map. Transmit controls may include transmit power (transmit amplitude); transmit focal zone location and transmit frequency. Image optimization controls may include; color flow gain; gate size; PRF (alters velocity scale); clutter filter select; frame rate/resolution control; preprocessing select; persistence; dynamic range (for Power Doppler only) and color map select.
Yet another example of a user interface is a PW Doppler user interface which may have PW Doppler format controls that may include PW Doppler mode select; trackball; activate PW cursor (trackball is used to adjust sample volume position); sample volume size; Doppler steering angle; sweep speed; update (selects either simultaneous or interval update imaging); audio volume control and flow vector angle. Transmit controls may include transmit power (transmit amplitude) and transmit frequency. Spectral Display optimization controls may include PW Doppler gain; spectral display size; PRF (alters velocity scale); clutter filter select; preprocessing and dynamic range.
An exemplary M-Mode user interface may have image format controls including M-Mode cursor activation; trackball (used to position cursor); strip size and sweep speed. Transmit controls may include transmit power (transmit amplitude); transmit focal zone location; transmit frequency and number of cycles. Image optimization controls may include M-Mode gain; preprocessing; dynamic range and post-processing.
An exemplary RF Mode user interface may have, for example, RF line acquisition controls that may include RF line position; RF gate; number of RF lines acquired; RF region activate; RF region location; RF region size; number of RF lines in region; averaging; and B-Mode interleave disable. Transmit controls may include transmit power (transmit amplitude); transmit focal zone location; transmit f-number; transmit frequency; number of cycles; acquisition PRF and steering angle. Receive processing controls may include RF Mode gain; filter type, order; window type and number of lines averaged.
The digital samples of the received signal are processed at a rate which is generally different from the rate at which the data is acquired. Such processing is referred to herein as “asynchronous signal processing.” The processing rate is the rate at which data is displayed, typically about 30 frames per second (fps.) As one would recognize, however, the data can be displayed at a rate up to the acquisition rate or can be displayed at less than about 30 fps. The data can be acquired at much faster frame rates, in certain embodiments of the invention at about 300 frames per second, or at a speed necessary to acquire the diagnostic information desired. For example, image date of a rapidly moving anatomical structures such as a heart valve can be acquired using a faster frame rate and then can be displayed at a slower frame rate. Data acquisition rates can be less than 30 fps, 30 fps, or more than 30 fps. For example, data acquisition rates can be 50, 100, 200, or 300 or more fps.
The display rate can be set such that it does not exceed that which the human eye can process. Some of the frames which can be acquired can be skipped during display, although all of the data from the receive beamformer output is stored in an RF data buffer such as the RF cine buffer 1713. The data is sometimes referred to as RF data or by the sampling method used to acquire the data, (for instance in the case of quadrature sampling, the data can also be referred to as baseband quadrature data). The quadrature or RF data is processed prior to display. The processing may be computationally intensive, so there are advantages to reducing the amount of processing used, which is accomplished by processing only the frames which are to be displayed at the display rate, not the acquisition rate. The frames that were skipped over during display can be viewed when live imaging stops or the system is “frozen.” The frames in the RF buffer 1713 can be retrieved, processed, and played back at a slower rate, e.g., if the acquisition rate is 300 frames per second, the play back of every frame at 30 frames per second would be 10 times slower than normal, but would allow the operator to view rapid changes in the image. The playback feature is usually referred to as the “Cineloop” feature by persons of ordinary skill in the art. Images can be played back at various rates, or frame by frame, backwards and forwards.
The system 1600 shown in
The multiplexing of the elements of the array transducer 1601 for the receive cycle can be carried out by a RX switch 1817 as shown in an exemplary diagram (
The exemplary front end 1816 shown in
The received signal from the selected array transducer element passes into the low noise amplifier (LNA) 1804. From the LNA 1804, the then amplified signal passes into time gain control (TGC) 1805. Since elapsed time is proportional to the depth of the received reflected signals, this is also referred to as a depth dependent gain control. In an ultrasound system, as time goes by from the transmission of an ultrasound wave, the signal passes deeper into the tissue and is increasingly attenuated; the reflected signal also suffers this attenuation. The TGC 1805 amplifies the received signal according to a time varying function in order to compensate for this attenuation. The factors which can be used to determine the time varying TGC gain are time of flight, tissue characteristics of the subject or subject tissue under study, and the application (e.g. imaging modality). The user may also specify gain as a function of depth by adjusting TGC controls on the user interface panel 1607. Embodiments may use, for example, an Analog Devices (Norwood, Mass.) AD8332 or similar device to perform the LNA 1804 and TGC 1805 functions. From the TGC 1805, the receive signal passes into the receive beamformer 1803 where it is sampled by a sampler, in this embodiment, the analog-to-digital converters 1807 and 1808. In other embodiments according to the invention only one analog-to-digital converter is used if sampling is done at a rate greater than the Nyquist rate; for instance at 2 or 3 times the Nyquist rate, where the Nyquist rate involves sampling the ultrasound signals from the individual elements at a rate which is at least twice as high as the highest frequency in the signal.
In other embodiments of the invention, quadrature sampling is employed and two analog-to-digital converters are used, namely the “I” and the “Q” sampler. In the exemplary embodiment of the receive beamformer 1803, the receive signal is digitized in blocks 1807 and 1808 using quadrature sampling analog-to-digital converters (ADC); two ADCs are required per channel, with sampling clocks shifted 90° out of phase. The sample rate used can be the center frequency of the receive signal. For comparison, direct sampling would use a sampling rate in theory of at least twice the highest frequency component in the receive signal, but practically speaking at least three times the sampling rate is preferred. Direct sampling would use one ADC per channel.
Once sampled, the now digitized received signal passes into a Field Programmable Gate Array (FPGA) in which various functions associated with receive beamforming are implemented. Within the FPGA, the digitized received signal can undergo a correction for the DC offset of the ADC. This is implemented by the subtraction of a value equal to the measured DC offset at the ADC output. Each ADC may have a different DC offset correction value. The DC offset may be determined by averaging a number of digital samples appearing at the output of the ADC with no signal present at the receive channel input, for example, during a calibration period at system start up. The digitized signal next passes into a FIFO buffer 1822 where each sample is stored for an appropriate duration so that the appropriate delay profile can be implemented. The delay can be implemented in both coarse and fine manners. A coarse delay can be implemented by shifting the signal by one or more sample points to obtain the desired delay. For instance, if the desired delay is one sample period, then shifting by one sample in the appropriate direction provides a signal with the appropriate delay. However, if a delay of a value not equal to the sample period is desired, a fine delay can be implemented using an interpolation filter 1809.
From the FIFO buffer 1822, the digitized received signal passes into the interpolation filter 1809 for the calculation of any fine delay. The interpolation filter 1809 is used in a system where the sample period is greater than the appropriate fine delay resolution. For instance, if the sample rate is the center frequency of the ultrasound signal and is 50 MHz, the sample rate is one sample every 20 nanoseconds. However, a delay resolution of 1.25 nanoseconds ( 1/16 of 20 nanoseconds) is used in certain embodiments to provide the desired image quality, though other delay resolutions are contemplated within the scope of this invention. The interpolation filter 1809 is used to calculate a value for the signal at points in time other than the sampled point. The interpolation filter 1809 is applied to the in-phase and quadrature portions of the sampled signal. Embodiments of the interpolation filter 1809 comprise a finite impulse response (FIR) filter. The coefficients of each filter can be updated dynamically by the beamformer control module based on the time of flight, sample by sample. After processing by the interpolation filter, a phase rotation can be applied by a multiplier 1811 multiplying the in-phase and quadrature components by the appropriate coefficients. The phase rotation is used to incorporate into the interpolated sample the correct phase relative to the ADC sample frequency. The RX controller 1810 controls the FIFO modules and the interpolation filters. The receive delay is updated dynamically, so the interpolation filter coefficients at each channel need to change at certain intervals. The delay implemented by the FIFO also needs to change at certain intervals. Also, the receive aperture size is adjusted dynamically, so each channel becomes active at a specific time during the reception of the ultrasound signal; a channel is activated by multiplying by 1 instead of 0 at the “multiply” module 1811. The multiply module 1811 can also apply a “weight” which is a value between 0 and 1, independently to each channel in the receive aperture. This process, which is known as apodization, is known to one skilled in the art. The value by which the interpolated sample is multiplied by may vary with time, so as to implement an apodized receive aperture which expands dynamically during the reception of the ultrasound signal.
The configuration for each receive line is stored in the Line Memory 1830. Each line configuration in the Line Memory 1830 contains the Aperture Select Index, the Mode Select, and the Aperture Enable. The Aperture Select index is used to determine the Aperture to Channel mapping. The Mode Select is used to access multiple delay profiles. The Aperture Enable index controls the initial aperture size. The aperture select look-up table (AP_SEL LUT) 1832 is a method to reduce the number of possible configurations and therefore number of bits required to store in the line memory. The AP_SEL LUT 1832 is re-programmable.
The Memory Control 1834 is a state machine that decodes the line configuration. The state machine is configured by the Control and Status memory 1836. It is configured differently for different modes (e.g. B-Mode, Color Flow Mode, PW Doppler Mode, etc.). The Memory Control 1834 controls the loading of the aperture memory into the RXBF Buffer 1826 and generates the SOL_delayed and FIFO_WEN signals. The pulse SOL_delayed is used to transfer the initial delay parameters into the RX Phase Rotation and RX Apodization block 1809 in a single RXCLK period. The dynamic receive parameters are then transferred in each subsequent RXCLK period. The FIFO_WEN signal starts the receive ADC data acquisition into the FIFO for the RX interpolation filter.
The Control and Status Memory 1836 also contains common parameters such as the Receive Length. The Receive Length parameter determines how many receive samples to collect for each line.
It is to be appreciated that increasing the number of receive channels allows for larger receive apertures, which can benefit deep imaging by improving lateral resolution and penetration. The synthetic aperture mode allows for apertures greater than 64 to be used, but at the expense of a reduction in frame rate. With an increase in the number of receive channels, this can be done without a frame rate penalty.
In one embodiment according to the present invention, the receive beamformer 1803 allows for multi-line beamforming. Multi-line beamforming allows for higher frame rates by processing multiple receive lines in parallel. Frame rate increases by a factor equal to the number of parallel receive lines. Since beamforming occurs simultaneously for multiple receive apertures, higher data processing rates through the interpolation filters 1809 are used. The amount of data transferred from the receive beamformer to a host CPU would increases by a factor equal to the number of parallel receive lines. The transmit beam is broadened so that it overlaps the multiple receive lines.
The signal from each receive beamformer 1803 is then summed by summers 1815. The summed signal represents a received signal at a given time that is reflected from a given depth. The summed received signal is then routed through modules described earlier and shown in
During the transmit operation cycle of the system 1600, selected transmit output stages are connected to the transmit channel in order to form the active aperture. In this aspect, the multiplexing is done prior to the transmit output stage. For example, as previously described, transmit channel 1 1801 can be switchably connected to the transmit output stages corresponding to elements numbered 1, 65, 129, and 193 in
Optionally, the transmit multiplexing can be done after the transmit output stage using multiplexing circuitry able to accommodate a higher voltage bipolar signal.
Referring back to
During the process of transmit beamforming, one or more of each of the transmit channels within the active transmit aperture can produce a transmit waveform which can be delayed relative to a reference control signal. The number of transmit channels determines the maximum transmit aperture size. The benefit of increasing the number of transmit channels is improved lateral resolution and penetration for deep imaging. In various embodiments, the array transducer has 64 transmit channels or may have 96 or 128 transmit channels. The delays can vary from channel to channel, and collectively the delays are referred to as the transmit delay profile. Transmit beamforming may also include the application of a weighting function to the transmit waveforms, a process known to one of ordinary skill in the art as “apodization.” Transmit apodization uses independent control of the amplitude of the transmitted waveform at each channel. The benefit to image quality is improved contrast resolution due to a reduction in spurious lobes in the receive beam profile, which can be either side lobes or grating lobes. Each transmitter output stage can have an independently controlled supply voltage, and control hardware.
Transmit waveshaping involves the generation of arbitrary waveforms as the transmit signal, i.e., the modulation of amplitude and phase within the transmit waveform. The benefit is an improvement to axial resolution through shaping of the transmit signal spectrum. Techniques such as coded excitation can be used to improve penetration without loss of axial resolution.
The transmit beamformer 1812 described herein may be implemented in one embodiment with an FPGA device. A typical implementation of a transmit beamformer 1812 which provides a delay resolution of, for example, 1/16 the transmit clock period may require a clock which is 16 times the transmit clock frequency. For the frequency range of the system described here, this would imply a maximum clock frequency of 16 times 50 MHz, or 800 MHz, and a typical FPGA device may not support clock frequencies at that rate. However, the transmit beamformer 1812 implementation described below uses a clock frequency within the FPGA of only eight (8) times the transmit clock frequency.
Each channel of the transmit beamformer is comprised of a TX controller 1814 and a Tx pulse generator 1813. The TX controller 1814 uses a parameter called, for example, an ultrasound line number (also known as a ray number), to select the active transmit aperture through the appropriate configuration of the transmit multiplexer. The ray number value identifies the origin of the ultrasound scan line with respect to the physical array. Based on the ray number, a delay value is assigned to each transmit channel in the active transmit aperture. The TX pulse generator 1813 generates a transmit waveform for each transmit channel using waveform parameters and control signals as described herein.
The delay profile and transmit waveform for the entire aperture is stored in the TX Aperture Memory 1838. Multiple delay profiles can be stored in the TX Aperture Memory 1838. Multiple delay profiles are required for B-Mode imaging in which multiple focal zones are used, and PW Doppler and Color Flow Imaging modes in which the Doppler mode focal depth and transmit waveforms are different than those used for B-Mode. In this exemplary embodiment, the TX Aperture Memory 1838 contains delay profile and transmit pulse wave shape data for a 64 channel aperture. On each Channel Board there are 16 transmit channels, each of which can be connected to one of four different array elements through a transmit output stage. A 64:16 crosspoint switch 1840 is used to route the correct transmit waveform data sets to each of the 16 channels. The control of the other 48 channels is implemented on the other 3 channel boards. The TXBF buffer 1842 temporarily stores the TX pulse generator data before the start of line (SOL) trigger. The TX_TRG trigger moves the data from the TXBF Buffer 1842 into the TX Pulse generator 1813 in one TXCLK period.
The configuration for each transmit line is stored in the Line Memory 1844. Each line configuration in the Line Memory 1844 contains the following information: Aperture Select Index, Mode Select, Aperture Enable Index, and Element Select Index. The Aperture Select index is used to determine the Aperture to Channel mapping. The Mode Select is used to access multiple delay profiles. The Aperture Enable index controls the aperture size. The Element Select index controls which element is active in the case that there are more array elements than transmit channels or receive channels. The indexing of the Aperture Select, Aperture Enable and Element Select look-up tables (AP_SEL LUT 1846, AP_EN LUT 1848, ES LUT 1850) is a method to reduce the number of possible configurations and therefore number of bits required to store in the line memory 1844. The look-up tables are all re-programmable.
The Control and Status memory 1852 contains common parameters such as the number of transmit cycles (TX Cycles), the number of lines in the frame, and also configures the state machine in the Memory Control block 1854. Memory Control 1854 is a state machine that decodes the Aperture Select, Aperture Enable and Element Select line information.
Referring now to
Transmit pulse generation begins when a TXTRG pulse 2206 is received from the channel control board 1814. The TXTRG signal 2206 is sent to the transmit beamformer channels, and is the signal which the transmit beamformer delays are referenced to. The TXTRG pulse 2206 begins the counting of ½ intervals of the transmit frequency clock cycle denoted by TXCLK×2 2246. The current hardware implementation uses a clock of 2 times the transmit clock. The coarse delay 2210 is implemented by a Coarse Delay counter 2248 which is clocked by a clock, TXCLK×2 2246. The signal TXTRG 2206 causes the count to begin.
A COARSE DONE signal 2208 is generated when the number of clock cycles of TXCLK×2 2246 has reached the coarse delay input variable value 2210. The COARSE DONE signal 2208 enables the byte select circuit composed of multiplexers 2250 and 2252, Pulse Inversion select Circuit composed of multiplexers 2254 and 2256, and the 8:1 parallel-to-serial circuits 2212 and 2213. The 16 bit waveshape words 2202 and 2203 are transferred into 16 bit registers 2216 and 2217. The output of the A waveshape register 2216 is composed of the Partial Waveshapes: Partial_Waveshape_A(7:0) 2260 and Partial_Waveshape_A(15:8) 2261. Partial_Waveshape_A(7:0) 2260 is transferred to the either 8:1 parallel-to-serial circuit 2212 or 8:1 parallel-to-serial circuit 2213 through the Pulse Inversion Circuit composed of multiplexers 2254 and 2256. Following the transfer of Partial_Waveshape_A(7:0) 2260, Partial_Waveshape_A(15:8) 2261 is transferred to the either 8:1 parallel-to-serial circuit 2212 or 8:1 parallel-to-serial circuit 2213 through the Pulse Inversion Circuit composed of multiplexers 2254 and 2256. The Byte Select signal 2214 controls which of Partial_Waveshape A(7:0) 2260 or Partial_Waveshape_A(15:8) 2261 is multiplexed through to the Pulse Inversion Circuit. In this way, the full 16 bits of Waveshape_A 2202 is transferred to the 8:1 parallel-to-serial circuits for serialization into a one bit data stream.
As can be seen from
The 8:1 parallel-to-serial circuit 2212 and 2213 have double data rate (DDR) outputs. COARSE DONE 2208 begins the count of the number of output pulses. When the pulse number counter finishes counting the number of pulses, the Enable signal 224 goes low causing the registers 2216 and 2217 to stop outputting the Partial Waveshapes. The 16-bit waveshape of the “A” phase 2202 is converted to 1 serial bit in two TXCLK×2 2246 cycles. The 16-bit waveshape of the “B” phase 2203 is also converted to 1 serial bit in two TXCLK×2 2246 cycles. Pulse inversion may be achieved by swapping the “A” and “B” phases before the signals are sent to the parallel-to-serial circuits. The signal swap occurs if the Pulse Inversion signal 2258 is enabled on the Pulse Inversion MUX circuit 2254 and 2256.
The 8:1 parallel-to-serial circuit with double data rate (DDR) output is clocked with TXCLK×8 2266 which is at a frequency of 8 times the transmit clock. With DDR output, the waveshape is shifted out at a rate of 16 times the transmit clock frequency. The signals from 8:1 parallel-to-serial circuit 2212 or 8:1 parallel-to-serial circuit 2213 are transferred out of the FPGA using the LVDS standard before it is re-synchronized by clock TXCLK×16 2236.
The “A” phase signal is re-synchronized by a low jitter positive emitter coupled logic (PECL) flip-flop 2234 and a low jitter clock, TXCLK×16 2236, at 16 times the transmit frequency. This can eliminate jitter added by the circuit inside the FPGA. The “B” phase signal is also re-synchronized by flip-flop 2235.
Both the “A” and “B” signals go to respective driver circuits 2238, 2240 to increase their current drive capability. The output of the drivers become signals TXB 2004 and TXA 2002 and connect to the transmit multiplexers in the front end circuit 2000.
Re-sending of the waveshape data 2202 and 2203 continues until the pulse number counter 2242 has reached the number specified by the pulse count input variable 2204 and the enable signal 2244 changes state.
The 16 bit word which constitutes Waveshape_A 2202 may change from one transmit cycle to the next. The same applies to Waveshape_B 2203. This allows for the generation of transmit waveforms with arbitrarily specified pulse widths from one cycle to the next. Waveshape_A 2202 and Waveshape_B 2203 are specified independently. For example, either odd or even transmit waveforms may be generated.
The TXPower signal (shown as “TX High Voltage” in
The data transfer to computer unit 1903 transfers data from the beamformer control board 1900 to the computer unit 1905. Embodiments can use a PCI express bus 1904, as is known in the art, for this transfer, or similar buses.
In the exemplary schematic shown in
Receive switching is effected by QTDP 2008, QTDN 2006, QLSH 2028, QLSL 2030, and SW1 2020. SW1 2020 is a receive multiplexing switch which can be a single pole single throw (SPST) or a single pole double throw (SPDT) switch of a type such as a GaAs PHEMT (gallium arsenide pseudomorphic high electron mobility transistor). Alternatively, the receive multiplexing switch may be implemented with other types of field effect transistors or bipolar transistors. If SW1 2020 is a SPDT switch it is configured as shown in
During receive intervals, the receive multiplexing switch is configured such that there is a connection between the array element and the receive channel. The pulser drive MOSFETs, QTDN 2006 and QTDP 2008, are both turned on during receive, while QLSH 2028, QLSL 2030, QTXMUXN 2016 and QTXMUXP 2014 are held off. This causes LTXS 2038 to present mainly its leakage inductance as an impedance in series with the receive signal. For received signals too small to forward bias D1 2010 or D2 2012, these diodes present high shunt impedance, dominated by their junction capacitance. L1 2040 and the leakage inductance LTXS 2038 are used to level the receive mode input impedance, compensating for the junction capacitance of D1 2010, D2 2012 and the capacitance of the ganged switches forming the receive multiplexer.
In an alternative implementation of the front end circuit, and as shown in
The pulser employs a center-tapped transformer and NMOS FETs, together with a switch-selectable level supply, to generate nominally square pulses. In order to control the delivered spectrum when connected to the transducer element thru a controlled impedance coax cable, it employs series and shunt resistances. These serve to reduce the time-variation of source impedance during operation of the pulser and provide back termination of the transducer during the interval immediately following transmit pulses. Not shown in the schematic is the drive circuit for the final stage MOSFETs. This circuit, (which is on the far side of a multiplexer as describe below), may be either a discrete switching MOSFET pulse amplifier or a collection of CMOS buffers sufficient to provide the required drive.
The transformer needed for the pulser is built as windings printed on the PCB augmented by small ferrite slabs fastened onto both sides of the PCB, around the windings. This technique is amenable to automated assembly provided the ferrite slabs can be packaged appropriately.
The following examples are put forth so as to provide those of ordinary skill in the art with a complete disclosure and description of how the articles, devices and/or methods claimed herein are made and evaluated, and are intended to be purely exemplary of the invention and are not intended to limit the scope of what the inventors regard as their invention. Efforts have been made to ensure accuracy with respect to numbers (e.g., amounts, temperature, etc.), but some errors and deviations should be accounted for.
The processing unit 2306 is partitioned into two major subsystems. The first is the front end 2308, which includes the beamformer, the front end electronics, the beamformer controller and the signal processing module. The second is the computer unit 2310, or back end. The front end subsystem 2308, is concerned with transmit signal generation, receive signal acquisition, and signal processing. The back end 2310, which can be an off-the-shelf PC motherboard, is concerned with system control, signal and image processing, image display, data management, and the user interface. Data can be transferred between the front and back end sub-systems by, for example, a PCI express bus, as is known in the art to one of ordinary skill.
The module which processes the receive signals is the receive beamformer, as previously described herein. The subsystem which generates the transmit signals is the transmit beamformer, also as previously described herein. Each channel of the transmit and receive beamformers is connected to a separate element in the array 2302. By altering the delay and amplitude of the individual transmit or receive signals at each element, the beamformer is able to adjust the focal depth, aperture size and aperture window as a function of depth. The exemplary system of
TABLE 2 Modes Supported B-Mode M-Mode PW Doppler Color Flow (Velocity) Doppler Power Doppler Tissue Doppler 2nd Harmonic Triplex EKV ECG triggered imaging 3-D imaging 3-D real-time (4 Hz) RF Mode Anatomical M-Mode
Exemplary specifications of the system shown in
TABLE 3 System Specifications Number of transducer elements supported Up to 256 Transmit channels (active aperture) 64 Receive channels 64 Transducers supported Linear, curved linear Center frequency range 15 to 55 MHz Data acquisition method Quadrature sampling BF sampling frequency range 15 to 62 MHz Receive BF fine delay implementation Interpolation filter Receive delay resolution T/16 ADC number of bits 10 Transmit delay resolution T/16 TGC yes Synthetic Aperture yes Maximum transmit voltage 80 Vpp Transmit power control yes Multiple Transmit focal zones yes Transmit cycle adjustment 1-32 B-mode frame rate max 200 CFI frame rate max 160 PW Doppler maximum PRF 150 KHz CFI maximum PRF 75 KHz Doppler beam steering yes Cine buffer size 300 frames Physiological signal acquisition yes Transducer connectors One or more
The system or portions thereof may be housed in a portable configuration such as, for example, a cart, including beamformer electronics 2316, a computer unit 2310, and a power supply unit 2312. The user interface includes an integrated keyboard 2318 with custom controls, trackball, monitor, speakers, and DVD drive. The front panel 2320 of the cart has connectors 2322 for connecting an array-based transducer 2302 and mouse physiological information such as ECG, blood pressure, and temperature. The rear peripheral panel 2314 of the cart allows the connection of various peripheral devices such as remote monitor, footswitch, and network 2324. The cart has a system of cooling fans 2326, air guides, and air vents to control the heat of the various electronics.
In one embodiment the computer unit 2310 may be an off-the-shelf Intel architecture processor running an operating system such as, for example, Microsoft Windows XP. The computer unit 2310 may be comprised of, for example, an Intel 3 GHz CPU (Xeon Dual Processor or P4 with Hyperthreading); 2 GB DDR memory; PCI Express x4 with cable connector; 100 Mbps Ethernet; USB 2.0; Graphics controller capable of 1024×768×32 bpp@100 Hz; Audio output (stereo); 2×120 GB 7200 RPM Hard disk drives (one for O/S+software; one for user data) and 300 W ATX power supply with power-factor correction.
In one embodiment the power supply unit 2312 may be comprised of the following: a universal AC line input (100, 120, 220-240 VAC, 50 or 60 Hz), where the AC input is provided by a detachable cable that connects to a system AC input terminal block and has AC distribution using IEC terminal blocks. In one embodiment, the inrush current is limited to 6 A or less during the first 100 ms of power up. The system cart of
Filtered ambient air is provided through the use of fans 2326 to the system cart electronics which include, for example, the beamformer electronics (i.e., the beamformer card cage 2316, power supply unit 2312, and computer unit 2310. The cooling system supports, for example, in one embodiment an ambient operating temperature range of +10 to +35° C., and the exhaust air temperature is kept below 20° C. above the ambient air temperature, though other ambient operating ranges are contemplated within the scope of this invention.
Electro-Magnetic Interference (EMI) Shielding
In one embodiment, the exemplary system is provided with a contiguous EMI shield in order to prevent external electromagnetic energy from interfering with the system operation, and to prevent electromagnetic energy generated by the system from emanating from the system.
The system shielding extends to the transducer cable 2304 and the array 2302, and the transducer connector 2322. The computer 2310 and power supply units 2312 may be housed in separate shielded enclosures within the system. All shields are maintained at approximately ground potential, with very low impedance between them. There is a substantially direct connection between the chassis ground of the system and earth ground. Also, in one embodiment the AC supply may be isolated from the system power supply by an isolation transformer as part of the power supply unit 2312.
An overview of an embodiment of the electronics for an exemplary system according to the invention is shown in
Exemplary channel boards are shown, and have been previously described, in reference to
As can be seen in
The transmit channels and transmit output stages generate bipolar pulses at a specified frequency ranging from about 15 to about 55 MHz, with a specified cycle count and amplitude. The transmit waveforms generated by each channel have a specific delay relative to the other channels with a resolution equal to approximately 1/16 of the period of the transmit frequency. The delay profile across the active transmit aperture is controlled by the transmit beamformer controller. A low jitter master clock is used to generate the transmit burst signals. The transmit output stage includes a means of adjusting the peak to peak voltage on a per channel basis, in order to create an apodized transmit aperture.
The receive channels provide variable gain adjustment, filtering and digitization of the receive signals, and receive beamforming. The gain is implemented with a variable gain amplifier which also acts as the preamplifier. Gain is varied throughout the acquisition of the ultrasound line according to a predetermined gain profile known as the TGC curve. Anti-aliasing filters precede the ADC (analog-to-digital converter) to prevent aliasing and to limit the noise bandwidth.
As shown in
The receive beamformer is setup via the RX Control Bus. The transmit beamformer is setup via the TX Control Bus. The control parameters are updated before the start of each ultrasound line. The control parameters are TX aperture, TX delay profile (coarse and fine delay), RX aperture, RX delay profile (initial, coarse and fine delay), RX phase, and RX apodization. When all the control parameters are set and the system is ready—a start-of-line (SOL) signal is sent to begin a transmit/receive cycle.
Transmit Output Stage
Multiplexing of the transmit channels occurs prior to the transmit output stage. Since the transmit beamformer can work with arrays with up to 256 elements, there are 256 transmit output stages, one per element. As shown and described in reference to
Each transmit channel is multiplexed to four output stages as can be seen in
As can be seen in
Front End Circuit
For a transducer array comprised of 256 elements, there are 256 front end circuit sections, one dedicated to each array element. As can be seen in reference to
Also as can be seen in reference to
The receive beamformer, such as the one shown in
Channel Board Configuration
As shown in the exemplary system of
Beamformer Control Board
The beamformer control board 2404 of the exemplary system of
The TX/RX controller 2412 provides master timing using start of frame and start of line synchronization signals to the transmit beamformer and receive beamformer. It sets up the beamformer parameters in memory via a custom local bus. All the low-jitter clock frequencies for beamforming are generated on the beamformer control board 2404.
The RF partial sum data from each channel board 2406 is summed 2414 together with synthetic aperture data 2416. Then the ray line data goes into a first-in-first-out (FIFO) memory 2418 where it sits temporarily before being copied to the RF Cine buffer 2420. The RF Cine buffer 2420 stores full frames of RF data and is randomly accessible. Data is read from the RF Cine buffer 2020 and copied to the host CPU via the PCI Express link 2410. Alternatively, the data can be processed by the signal processor module 2422 before being sent to the main computer unit. The data is then buffered, processed further and displayed by the application software and application user interface that runs on the main computer unit.
The data traffic control and reading/writing of control parameters is facilitated by the embedded CPU 2424. The embedded CPU 2424 itself is accessible by the host CPU via the PCI Express link 2410. Other functions provided by the beamformer control board 2404 are the physiological acquisition system and power supply monitoring.
Transmit Beamformer Control:
The transmit (TX) beamformer control updates the transmit beamformer parameters each transmit line. The parameters include number of coarse delay cycles at the transmit center frequency (fc), number of fine delay cycles (at 16×fc), transmit waveshape (at 16×fc), number of transmit cycles, transmit select, and transmit voltage. The transmit beamformer control also schedules the updating of parameters for duplex mode, triplex mode, or multiple focal zones.
Receive Beamformer Control:
The receive beamformer control controls the receive delay profile, aperture size and apodization for each channel. The delay control consists of coarse and fine delays, which are controlled by the dual port RAM read pointer and the interpolation filter coefficient selector bit, respectively.
The aperture control signal controls the aperture size dynamically by specifying when the output of each channel becomes active. This is done by controlling the clear signal of the final output register of the interpolation filters. Dynamic receive apodization is controlled by five bits of apodization data with which the signal in each channel is multiplied. The receive control signals are read out from a control RAM at the input sample clock rate as shown in
A block diagram of transmit/receive synchronization is shown in
The phase difference between transmit 2708 and receive clocks 2710 is fixed as long as the TX_Sync_Period 2712 is calculated correctly. The TX_Sync_Period 2712 is the minimum number of transmit clock cycles required to achieve synchronization. For example, if the transmit clock frequency is 30 MHz and the receive clock frequency is 25 MHz, TX_Sync_Period 2712 will be 6 cycles of the transmit clock.
The clock generator 2428 provides the appropriate clock frequencies for transmit and receive beamforming. It comprises a low-jitter master clock, a programmable divider, clock buffers and re-synchronization circuits. The frequencies are: transmit frequency (fc)—25 to 50 MHz; receive frequency—20 to 50 MHz in-phase and quadrature; digital clocks—fc×2, ×4, ×8, ×16. The fastest clock used in this exemplary embodiment can be 800 MHz (50 MHz×16).
PCI Express Bridge
The PCI Express bridge 2426 connects the host CPU and the embedded CPU 2424 via a PCI bus 2410. This allows DMA transfers from the RF cine buffer 2420 to the host processor memory and vice versa. PCI Express builds on the communication models of PCI and PCI-X buses. PCI Express uses the same memory mapped address space as PCI and PCI-X with single or burst read/write commands. However, PCI Express is a point-to-point serial interconnect using a switch to connect different devices, whereas PCI and PCI-X are parallel multi-drop buses. PCI Express can be used as a chip-to-chip or board-to-board communication link via card edge connectors or over a cable.
The bandwidth of the PCI Express link may be, for example: Uplink—210 MB/s burst and 140 MB/s sustained rate for RF Data, MIS Data, and diagnostics; Downlink—20 MB/s burst and <1 MB/s sustained rate for writing control parameters.
Synthetic Aperture FPGA
The partially summed beamformer RF data from the channel boards 2416 is first processed in the synthetic aperture FPGA. The processing comprises beamformer final summation, synthetic aperture and write to FIFO.
RF Cine Buffer
Functionally, the RF cine buffer 2420 is, for example, a 1 GByte dual port RAM. The RF cine buffer 2420 is a random access memory block that stores RF data organized in lines and frames. The data can be input and output at different rates to support asynchronous signal processing. The data stream is made up of interleaved I and Q beamformed data. The FIFO buffer provides storage of the beamformer data while the memory is being read by the CPU for the next display period.
In one embodiment, buffer specifications may include, for example: Storage—300 Full Size Frames (512 ray lines×1024 samples/line×32 bits I&Q data); Buffer Size—>629 M bytes; Input rate—140 Mbytes/sec; Output rate—140 Mbytes/sec (RF Data Rate) 32 Mbytes/sec (Video Rate).
Asynchronous Signal Processing
According to an embodiment of the described exemplary ultrasound system, it is capable of very high acquisition frame rates in some modes of operation, in the range of several hundred frames per second. The display rates do not have to be equivalent to the acquisition rates. The human eye has a limited response time, and acts as a low pass filter for rapid changes in motion. It has been demonstrated that frame rates above 30 fps have little benefit in adding to perceived motion information. For this reason, displayed ultrasound image information can be processed at a rate of 30 fps or lower, even when the acquisition rates are much higher. To uncouple acquisition from signal processing, a large RF buffer memory is used to store beamformer output data. An exemplary structure for buffering the beamformer output date is shown in
During active acquisition, the beamformer summation output is written by the Write Controller 2802 to the next available frame storage area, which is typically the storage area immediately following that pointed to by the “last frame” pointer. As the data in each new frame is acquired, the “first frame” and “last frame” pointers are updated so that the data is written to the correct address in the buffer. When acquisition is stopped (freeze), the buffer then contains the last N frames, with the “first frame” pointer indicating the oldest frame in the buffer.
The signal processing module 2422 has access to the RF memory buffer 2420. It accesses one acquisition frame at a time, at the display frame rate to produce the displayed estimate data. While the system is scanning, a timer signals the signal processing module that a display frame is required. At that time, the signal processing module 2422 will check to see if a new acquisition RF frame is available, and if so, will read the data and begin processing it. If the acquisition rate is faster than the display rate, the acquisition frames will be decimated prior to processing and display. After the system has been put in freeze, the RF frames stored in the memory buffer can be processed at any desired rate, up to the original acquisition rate.
Signal Processing Module
The beamformer control board 2404 comprises a signal processor 2422 in the data path to reduce the data load and/or computation load on the host CPU. The processor 2422 may be, for example, a FPGA with a sufficient number of multipliers and memory, or a CPU such as, for example, a 970 PPC or a general purpose DSP. The signal processing functions performed are divided between the signal processing module 2422 on the beamformer control board 2404, and the main computer unit (i.e., host computer). They include post-beamforming gain control, B-Mode amplitude detection and log compression, PW Doppler spectral estimation, color flow clutter filter and frequency/power estimation, asynchronous signal processing or frame averaging. Factors that may be considered in deciding where the processing takes place include the processing speed required, the complexity of the process, and the data transfer rates required.
B-Mode Signal Processing
For B-Mode imaging, the signal processing module 2422 performs processes which may include line interpolation, detection and compression.
Color Flow Imaging (CFI) Signal Processing
In one embodiment according to the present invention, Doppler color flow imaging is combined with B-Mode imaging such that the common blocks of the B-Mode signal path and the Doppler color flow signal path are time multiplexed to provide both types of processing. Typically, the B-Mode lines are acquired in between the CFI ensembles at rate of 1 or 2 lines for each ensemble, depending on the relative ray line densities of B-Mode and CFI (typical CFI images use half the ray line density of B-Mode), as is known to one of ordinary skill in the art.
For CFI, the Signal Processing Module 2422 performs processes that may include: ensemble buffering; clutter filter; velocity estimate calculation; power estimate calculation and variance estimate calculation.
After the I and Q waveforms from the receive beamformer summed output have passed through the clutter filter, the various parameters of the Doppler signal are estimated by a Doppler frequency and power estimator in either the host computer or the CPU 2424 on the beamformer control board. The parameters estimated for each sample depth in the ensemble may include Doppler frequency, Doppler power, and the variance of the frequency estimates. These parameters may be used in a decision matrix to determine the probability that the frequency estimate is a true estimate of the Doppler spectrum, rather than a noise or clutter signal estimate. Color flow velocity estimates are derived from the Doppler frequency estimates. All of the estimates are derived using a 2-D autocorrelation method as is known to one of ordinary skill in the art.
PW Doppler Signal Processing
Pulsed Doppler acquisition may be either by itself, in duplex mode, or in triplex mode. In duplex mode, the PW Doppler transmit pulses are interleaved with the B-mode transmit pulses so that the B-mode image is updated in real time while the PW Doppler signal is acquired. The method of interleaving depends on the Doppler PRF selected. The components shared between B-Mode imaging and Pulsed Doppler processing are time multiplexed to accomplish both types of processing.
In triplex mode, Pulse Doppler is combined with B-Mode and color flow imaging. The simplest implementation of triplex mode is a time interleaving of either a B-Mode line or a CFI line, in a fixed sequence that eventually results in a full frame of B-Mode and CFI image lines. In this implementation, the PRFs for both Pulsed Doppler and CFI are reduced by half, compared with their normal single modes of operation.
The I and Q samples for each ray line are range gated (a selected range of I or Q signals are separated out from the full range available and averaged to produce a single I,Q pair), to select the region of interest for the Doppler sample volume. The length of the range gate can be varied, if desired, by the user to cover a range of depth. The resulting averaged I,Q pairs are sent to a spectral processor, as well as an audio processor, which converts the I, Q Doppler frequency data to two audio output streams, one for flow towards the transducer (forward) and the other for flow away from the transducer (reverse).
For PW Doppler imaging, the signal processing module 2422 performs processes including range gating (digital integration).
M-Mode Signal Processing
For M-Mode imaging, the signal processing module 2422 performs processes including detection and compression.
EKV Signal Processing
EKV is an acquisition method in which extremely high frame rate images are generated (1000 frames per second and higher) as a post processing operation using ECG (electro-cardio-graph) signals used as timing events. EKV imaging may be implemented with either a single element mechanically scanned transducer, or with a transducer array. EKV imaging involves the acquisition of ultrasound lines at a PRF of 1000 Hz or higher at each line position in the 2-D image over a time period. The time period over which ultrasound lines are acquired at each line position, referred to as the EKV Time Period, can be for example, 1 second, which is long enough to capture several cardiac cycles in a mouse or other small animal. The acquisition of each ultrasound line involves the firing of a single transmit pulse followed by acquisition of the returning ultrasound data. For example, if there are 250 lines in the 2-D image, a total of 250,000 ultrasound lines will be acquired in the EKV data set. Each frame of the EKV image is reconstructed by assembling the ultrasound lines which were acquired at the same time during the cardiac cycle.
In one embodiment, the sequence of acquisition of the EKV data set may be such that the ultrasound line position remains static while the ultrasound lines are acquired over the time period. For example, if the time period is 1 second, and the PRF is 1 kHz, 1000 ultrasound lines will be acquired at the first ultrasound line position. The line position can then be incremented, and the process repeated. In this way all EKV data for all 250 lines in the 2-D image will be acquired. The disadvantage of this method of sequencing is that length of time required to complete the full EKV data set can be relatively long. In this example the time would be 250×1 second=250 seconds.
In a preferred embodiment which makes use of an array, the method of interleaving allows for a reduction in the length of time required to complete the EKV data set. For example if the PRF is 1 kHz there is a 1 ms time period between pulses during which other lines can be acquired. The number of ultrasound lines which can be acquired is determined by the two-way transit time of ultrasound to the maximum depth in tissue from which signals are detected. For example, if the two-way transit time is 20 μsec, 50 ultrasound lines at different line positions may be interleaved during the PRF interval. If we label the line positions L1, L2 . . . L50, one exemplary interleaving method can be implemented as follows:
Position of acquired
. . .
. . .
. . .
. . .
. . .
. . .
The sequence in the above table is repeated until the EKV Time Period has elapsed, at which time there will be a block of data consisting of 1000 ultrasound lines acquired at 50 different line positions, from line 1 to line 50. The acquisition of the block of data is then repeated in this manner for the next 50 lines in the 2-D image, line 51 to line 100, followed by acquisition over lines 101 to 150, etc., until the full 250 line data set is complete.
The total time required for the complete data set over 250 lines is reduced by a factor equal to the number of lines interleaved, which in this example is 50. Therefore the total length of time required would be 5 seconds.
The embedded CPU 2422 on the beamformer control board 2404 is, in one embodiment, a 32-bit embedded microprocessor with a PCI interface 2426 and a DDR memory interface. The main function of the embedded CPU 2424 is data traffic control. It controls data flow from the receive beamformer FIFO 2418 to the RF cine buffer 2420, from the RF cine buffer 2420 to the signal processing module 2422, and from the signal processing module 2422 to the host PC.
The beamformer control and diagnostics information is memory mapped on the target PCI device as registers. The embedded CPU 2424 decodes the location of the registers and relays the information over the appropriate local bus. The local bus can be, for example, PCI, custom parallel (using GPIO), I2C serial, or UART serial, as each are known in the art.
Physiological Acquisition System
The physiological acquisition system 2430 (or “mouse acquisition system”) filters and converts analog signals from the mouse information system inputs 2438. These signals may include subject ECG, temperature, respiration, and blood pressure. After data conversion, the data is transferred to the embedded CPU 2424 memory via local bus, and then on to the host CPU for display via the PCI Express link 2410.
Power Supply Monitoring
The beamformer control board 2404 monitors the rack power supply 2432 and lower voltages generated on each board. For example, the rack power supply 2432 may provide +48 VDC to the backplane 2408. In one embodiment, two high voltage post regulators 2436 on each channel board 2406 supply the transmit portion of the front end circuit. The beamformer control board 2404 monitors these regulators for over-current or over-voltage situations
The backplane 2408 mounts to the instrument electronics card cage. In one embodiment it has blind mate edge connectors to allow each of the boards to plug in, though other connection schemes are contemplated within the scope of this invention. It provides interconnection between boards, and input/output connectors for signals outside the card cage. In one embodiment, the size of the backplane is 8 U high by 84 HP wide so that it may fit in a 8 U×19″ rackmount VME-style card cage. The card cage depth may be 280 mm in one embodiment.
An overview of an embodiment of system software 2330 is shown in
The framework 3018 which determines the overall structure of the components can be used to produce an application executable by the operating system of the processing platform of the computer unit 2310 and to interface with the operating system. For example, the framework 3018 may produce a Windows application and interface with the Windows operating system.
The application controller 3020 software component can be the state machine for the system software 3000. It may control the interaction between the operator, the system software 3000 and the front end 2308.
The application view 3022 software component can provide a foundation to support the presentation of the system software 3000 based on the state machine in the application controller 3020 software component as previously described herein.
The studies component 3002 may allow the operator to perform studies, review study data, edit content, and import or export study data. As previously described herein, there can be various operating modes supported by the system for acquiring data and can be managed by a modes 3004 software component of the system software 3000. The supported modes may include, for example, B-Mode, 3D Mode, M-Mode, PW-Doppler, Color Flow Doppler, etc. Each mode has adjustable parameters, cine loop acquisition, and main image display area, which may be managed by the modes 3004 software component. Some modes may operate simultaneously, e.g., PW-Doppler and B-Mode.
The beamformer control 3024 software component can generate the imaging parameters for the front end based on the settings in the system software 3000.
The user data manager 3006 software component may maintain user preferences regarding how the system is configured.
The measurements 3026 software component may allow the operator to make measurements and annotations on the mode data.
The calculations 3028 software component may allow the operator to perform calculation on measurements.
The utilities layer 3008 software component contains common utilities that are used throughout the application as well as third party libraries.
The hardware layer 3012 software component is used to communicate to the beamformer hardware via the PCI Express bus, as previously described herein.
The physiological 3030 software component can be used to control the physiological data collection through the hardware layer 3012 as previously described herein.
The data layer 3010 may contain a database of all the different sets of parameters required for operation. The parameters may be set depending on the current user configuration and mode of operation.
The message log 3014 and engineering configuration 3016 may be used for diagnostic reporting and troubleshooting.
Transducer Select Board
Referring back to
Another exemplary embodiment of the high frequency ultrasound imaging system comprises a modular, software-based architecture described below and as shown in
The embodiment of
The CPU/signal processing module 3106 is responsible for processing the RF data from the beamformer 3102 for image formation, or Doppler sensing. The signal processing module 3106 can comprise a CPU module with the processing tasks implemented in software executing in a general purpose computing environment. Alternatively, the signal processing module 3106 can be implemented with some signal processing functions in hardware or in software executing on dedicated processors, in which case an additional signal processing module can be implemented as a plug-in card to the system CPU 3108.
If a dedicated hardware solution is chosen for the signal processing module 3106, it can be implemented with high performance CPUs. Optionally it can be implemented with digital signal processing chips (DSPs). One type of DSP which may be used is of the floating point variety, as are known in the art, and can be controlled by the host CPU, as well as being “data driven.”
The system CPU 3108 can act as both a user interface/control system as well as a signal/image processing sub-system. System control information can be distributed using memory mapped I/O, wherein modules interface to the peripheral bus of the CPU module. Optionally, the system CPU 3108 can be physically separate from the beamformer module 3102 and can be connected via a PCI Express cable (or equivalent) 3110. An exemplary PCI Express cable 3110 is one that supports transfers up to 1 GB/sec and lengths of three meters. Some or all of the memory that exists on various modules can be mapped into the CPU's 3108 memory space, allowing for access to parameters and data.
The system CPU 3108 in the exemplary architecture can perform a number of real-time processing tasks, including signal processing, scan conversion, and display processing. These tasks can be managed in a manner that does not require a “hard” real-time operating system, allowing for expected system latencies in the scheduling of processing routines. In addition, the system CPU 3108 can be responsible for managing the user interface to the system, and providing setup and control functions to the other modules in response to user actions. The CPU motherboard and operating system can support multiple CPU's, with fast access to a high speed system bus, and near real-time task management.
The beamformer module 3102 of this exemplary system comprises a transmit beamformer. The transmit beamformer can provide functions which may include, for example, aperture control through selection of a subset of array elements, delay timing to start of transmit pulse, transmit waveform generation, and transmit apodization control. For this exemplary embodiment, a transducer array 3112 is utilized. In one embodiment, this transducer array 3112 contains up to 256 elements. To eliminate the need for high voltage switching of transmitter pulse drivers to transducer elements, the transmit beamformer component of the beamformer module 3102 may be comprised of a number of transmitters equivalent to the number of transducer array elements. For instance, in an exemplary array transducer having 256 elements the transmit beamformer comprises 256 transmitters. Optionally, the transmit beamformer can comprise less than 256 transmitters and a high voltage switching method to connect an individual transmitter to a specific element. High voltage multiplexers are used to select a linear subset of elements from a 256 element array.
Optionally, the transmit beamformer component of the beamformer module 3102 may comprise high voltage pulser drivers for all 256 elements of the exemplary array, and a switching mechanism which connects a subset of transmit waveform generators to the appropriate drivers/array elements. This optional embodiment uses 256 TX/RX switches for protection of the receiver inputs with low level multiplexing to select the subset of array elements for the receive aperture. The low level multiplexing can optionally be combined with the TX/RX switches and in some cases has less attenuation of the receive signals and faster switching, when compared with a high voltage MUX scheme.
Transmit delays of 1/16 wavelength can be used and provide appropriate focusing and side lobe reduction in the transmit beam. For desired steering and focus control, the maximum delay times, when measured in wavelengths, can be at least 0.7 times the largest transmit aperture. For example, with 128 transmitters and an array spacing of 1.5 wavelengths, the largest transmit aperture is 192 wavelengths. At 20 MHz center frequency, the maximum transmit delay times can be at least 6.72 microseconds.
For 1/16 wavelength accuracy, the highest center frequency of interest specifies the delay resolution. At 50 MHz, this gives a delay accuracy of 1.25 nsec, which uses the equivalent of an 800 MHz clock and a 13 bit counter to achieve the maximum delay time of 6.72 usec. Optionally, instead of a high frequency clock a four phase clock at 200 MHz can be used. This would allow selecting a specific transmit delay by selecting one of the four phases of the 200 MHz clock as input to an 11 bit counter, which is preloaded with the number of 200 MHz clocks in the delay time.
The transmit beamformer component of the beamformer module 3102 further comprises a bi-polar transmit pulser. This type of pulser drive is typically specified with three parameters: T1, which is a transmit frequency (duration of half cycle); T2, which is a half cycle on time (duration of either positive or negative half cycle pulse); and T3, which is a pulse duration (number of half cycle pulses in total transmit). These durations are shown in
The control of the half cycle pulse duration, T2, allows for closer approximation to a sine wave drive, with improved transducer output. It can also be used to obtain a somewhat crude apodization of the transmit pulse power, provided that sufficiently fine control of the duration is provided.
Transmit apodization can be used to reduce spurious lobes in the transmit beam, which can be either side lobes or grating lobes. Apodization of the transmit aperture results in reduced power output and worse lateral resolution, so it is not always desirable. Often a small amount of apodization capability, such as providing for only a few levels of power output, is sufficient to achieve a good compromise between spurious lobe reduction and lateral resolution. The pulse width modulation scheme mentioned above for transmit waveform generation is one possible means of providing limited transmit apodization. A second method is to provide not one, but possibly four or more levels of high voltage for the pulser drivers, with a means to select one of these levels on each pulser.
The beamformer module 3102 also comprises a receive beamformer component. There are several different receive beamforming implementations which can be used in the exemplary system. The digital methods discussed below have least one A/D converter for each element in the receive aperture. In this exemplary embodiment, the A/D converter bit depth is 10 bits, which gives the desired beamforming accuracy at −50 dB signal levels. The A/D dynamic range is chosen to reduce spurious lobes and thus provide contrast resolution as desired. Eight bit A/D converters can be used if appropriate. Embodiments of the exemplary system use 64 receive channels, combined using synthetic aperture to implement a 128 channel receive aperture for applications where maximum frame rate is not needed. One optional method for digital receive beamforming implementation samples the ultrasound signals from the individual elements at a rate which is at least twice as high as the highest frequency in the signal (often called the Nyquist rate.) For example, a 50 MHz, 100% bandwidth transducer the Nyquist sampling rate is 150 MHz or higher.
Another optional sampling method for the receive beamformer component of the beamformer module 3102 is bandwidth sampling. Sampling theory, as known to one skilled in the art, provides that if a continuous function only contains frequencies within a bandwidth, B Hertz, it is completely determined by its value at a series of points spaced less than 1/(2*B) seconds apart. Sampling a band-limited signal results in multiple copies of the signal spectrum appearing at a fixed relationship to the sampling spectrum. Provided these replicated spectra don't overlap, it is possible to reconstruct the original signal from the under-sampled data. For example, consider a signal with a maximum bandwidth of 20 MHz centered at 30 MHz, and sampled at a rate of 40 MHz. In this situation, the spectrum is replicated as shown in
Another form of bandwidth sampling, known as quadrature sampling, can optionally be used in an embodiment of the receive beamformer component of the beamformer module 3102. In this sampling method, two samples are taken at 90° phase with respect to the center frequency. These samples can be repeated at an interval which is consistent with the bandwidth of the signal. For example, if the quadrature samples are taken at every period of the center frequency, the sample rate supports a 100% bandwidth signal. The sample pair resulting from quadrature sampling is not a true complimentary pair, since the samples are taken at different times, however they are true samples of the analytic waveforms, and concurrent quadrature samples can be found by interpolating the samples of the two I and Q sampled waveforms to the same point in time. Quadrature sampling may be implemented with one high sample rate converter sampling at four times the center frequency or with two lower frequency converters each operating at the center frequency but with clocks differing in phase by 90° with respect to the center frequency
Optionally, yet another form of sampling can be used in the receive beamformer. This form of sampling is Nyquist sampling combined with bandwidth sampling. Normal Nyquist sampling is used for the lower transducer center frequencies and bandwidth sampling for the higher frequencies. Commercially available 10 bit A/Ds with maximum sample rates of 105 MSPS are available. With this sample rate capability, a 30 MHz center frequency transducer with 100% bandwidth can be sampled adequately at Nyquist rates. At 40 MHz, Nyquist sampling can be used for transducers with bandwidths up to approximately 60%, so for this center frequency or higher, bandwidth sampling can be used. If these higher sample rates are used, the beamformer processing circuitry also accommodates the higher clock rates and increased storage requirements.
A variation of quadrature sampling can be used to provide a higher bandwidth beamforming capability for those applications that can benefit from it (for example, harmonic imaging) In this method, two quadrature sample pairs may be acquired for every cycle of the center frequency. For example, consider the sampling of a signal which has a center frequency of 30 MHz and significant spectral content beyond 100% bandwidth, such that the spectrum extends to frequencies less than 15 MHz and/or greater than 45 MHz. Two A/D converters per channel may be used to acquire the RF signal at that channel, each sampling periodically at twice the center frequency, 60 MHz. The sample clock of the second A/D converter is delayed by ¼ the period of the 30 MHz center frequency relative to the sampling clock of the first A/D converter. Every second sample acquired by the A/D converters will be multiplied by −1. The sample stream originating from the first A/D converter will then be the down-converted quadrature (Q) sample stream, and that originating from the second A/D converter becomes the down-converted in-phase (I) sample stream. The fine delay required for receive beamforming may be implemented by interpolation of the quadrature samples. This method allows for accurate sampling of the RF signal over 200% bandwidth of the center frequency.
In an alternative method of providing higher bandwidth beamforming capability which requires one A/D converter per receive channel, the RF output of the beamformer can be formed using two acquisition pulses, similar to a synthetic aperture approach. For example, consider a 30 MHz signal spectrum with 100% bandwidth, so that the −6 dB spectrum extends from 15 to 45 MHz. In this case, the signal can be sampled at a 60 MHz sample rate, and the sign of every other sample flipped, to provide a down-converted sample stream that can be taken as the Q channel of a quadrature down-conversion scheme. On the next acquisition, the sampling clock is delayed by ¼ of the period of 30 MHz, providing (after flipping the sign of alternate samples) the I quadrature waveform. These two quadrature waveforms are then time shifted and combined after beamforming to reconstruct an RF signal that is accurate for 200% bandwidth of the 30 MHz center frequency. This is adequate to capture all the information from an ultrasound transducer with 100% bandwidth. The frame rate is reduced by half compared with single pulse ray line acquisition. Higher frame rates can be achieved over a region of interest by reducing the number of image lines.
In the exemplary embodiment of
Bandwidth sampling interpolation may be described using the following example. For an exemplary 30 MHz array using bandwidth sampling, the sample rate on all A/D converters can be set to 40 MHz, providing a 66.7% bandwidth. With 128 receive channels, about 10 micro-seconds is desired for maximum delay, thus implementation uses a programmable shift register of about 400 stages. At 40 MHz, the programmable interpolators need only calculate one of eleven intermediate sample values (for 1/16 wavelength accuracy), equally spaced between adjacent 40 MHz samples. The interpolators can be specifically designed for bandwidth sampling to provide for accurate signal reconstruction. Samples can be taken from the output of all channels' interpolators, and summed to produce the sampled RF waveform for the desired beamforming direction.
The signal reconstruction process for interpolating between bandwidth sampled data points is simplified for the example 30 MHz array given above. In this case, every odd sample can be taken as samples of the Q component of the quadrature baseband representation of the signal (with alternate sign), while even samples can be considered to be samples of the I component. A simple bandlimited interpolator can be used to find the I and Q signal values at the appropriate intermediate time point, which can then be combined to reconstruct the RF value. If desired, all of the bandwidth sampled data points can be down-converted by the interpolation filters, resulting in a baseband quadrature sampled beamformer output, which can simplify downstream signal processing.
Quadrature sampling interpolation may be described using the following example. In this example, the input signals for each channel are assumed to be quadrature sampled, at one quadrature pair per cycle of the transducer center frequency, providing an input bandwidth of 100% around the center frequency. The two samples in the pair are taken at 90 degrees phase difference with respect to the center frequency, which provides actual samples of the Q and I baseband signals, but the waveforms are sampled at different points in time. Before the Q and I data can be combined, this sampling offset is corrected using interpolation filters. The interpolation required for correcting the sample offsets can optionally be incorporated into the interpolation filters used for beamforming.
Since the quadrature sampling method proposed generates baseband I and Q signals, the interpolation filters are operating on these signals, rather than the RF waveforms. The samples for all channels are taken at the same time, which leads to I and Q waveforms with the same phase relative to an RF waveform common to all channels. This is equivalent to using mixers on all channels to derive I or Q signals, where the carrier frequency for the mixers all have the same phase. However, correct summation of the I and Q samples from different channels can be provided by adjusting the carrier phase on each channel to match the phase of the time delayed echo waveforms. This amounts to a phase rotation of the interpolated I, Q samples according to the interpolation point relative to 0 degrees phase of the RF center frequency period. This rotation can also be incorporated into coefficients of a FIR interpolation filter, to produce a corrected I and Q output from each channel that can be summed coherently.
As way of explanation of the quadrature sampling interpolation beamforming method, one can first consider a simpler conceptual model, rather than an actual implementation. In this model, interpolation will be implemented to 16 separate points over the period of the center frequency, providing 1/16 wavelength accuracy for beamforming. This level of accuracy has been shown to be sufficient to provide no significant degradation of beam profiles. Considering a quadrature sampled waveform as shown in
Typically, a four point FIR filter is sufficient for accurate interpolation. To interpolate the points 0-3, between Q and I samples, a window of eight samples can be used, as shown in
To interpolate the points 4-15, the window is moved forward by one sample, as shown in
Using these windows, a set of eight coefficients for each interpolation position can be computed, which when multiplied times the sample values in the window, yields the interpolated I and Q values. In the case of the first window, the interpolated I value would be the sum of the even numbered products (0,2,4,6) while the Q values would be the sum of the odd numbered products (1,3,5,7). In the case of the second window, the I and Q values would be reversed.
After applying the appropriate interpolation filters to each of the channel data shown above, and I and Q sample is obtained for each channel that corresponds to the appropriate delay for the range point. As previously described herein, these I and Q sample pairs can not be simply summed to derived a beamformed I,Q pair, since the phase of the I,Q sample from each channel is different. Before the I,Q pairs from each channel can be summed, each channel's I,Q pair is phase rotated to correspond the same phase with respect to the delay time implemented. For example, if two channels are receiving an echo return, where the path length difference to the range point corresponds to exactly ½ wavelength of the echo frequency, and these echo returns are sampled at the same times by our quadrature sampling scheme, the samples will fall on different points on the RF signals, and the resulting I,Q waveforms will be 180 degrees out of phase. This situation is illustrated in
Using these rotation equations on the interpolated I and Q samples allows the rotated I's and Q's to be summed coherently. The rotation of the I and Q samples can be incorporated into the 8 coefficients used for interpolation. For example, when using the first interpolation window, where the even samples are I samples, the sin (angle) in equation (1), above, can be multiplied by each of the I coefficients, and the cos (angle) term multiplied by each of the Q coefficients. The resulting FIR filter then provides the rotated Q value, when all product terms are added together. Similarly, another set of coefficients can be used to compute the rotated I value. In this scheme, the FIR filter operates twice per sample period, using different coefficients, to produce an output stream of rotated Q and I values. This stream can be summed with the stream of rotated Q and I values from other channels to produce the beamformer output, which in this case is interleaved I,Q data representing the down-converted summed RF. Alternatively, the interpolation of the Q and I values may be implemented with separate FIR filters, each with 4 coefficients. In this scheme, the phase rotation is implemented in a stage following the interpolation.
The sampling scheme in which two quadrature pairs are acquired for each period of the center frequency also requires a phase rotation after interpolation of the quadrature samples. In this scheme two A/D converters per channel may be used to acquire the RF signal at that channel, each sampling periodically at twice the center frequency. The sample clock of the second A/D converter is delayed relative to the sampling clock of the first A/D converter by ¼ of the period of the center frequency. Every second sample acquired by the A/D converters will be multiplied by −1. Interpolated values can be calculated for 16 separate points over the period of the center frequency, or for 8 points over the period of the sample clock. The interpolation points calculated over a span of two sample clock periods may be numbered from 0 to 15. The amount of phase rotation required is the interpolation point number multiplied by 2*pi/16. For example, when the interpolation point is located at ⅛ of a sample clock period after the start of odd numbered sample clock cycles, the amount of phase rotation will be 2*pi/16. When the interpolation point is located at ⅛ of a sample clock period after the start of even numbered sample clock cycles, the amount of phase rotation will be 2*pi*( 9/16). The interpolation points may be shifted by 1/32 of the center frequency so that the actual sample values don't fall on an interpolation point on order to ensure that the filter function inherent in the interpolation filter is applied to all points. After the phase rotation, the values can be summed to provide the beamformer output. The amplitude of the envelope of the received signal output from a quadrature beamformer may be determined by calculating the square root of the sum of the squares of the I and Q samples. A compression curve may then be applied to the envelope amplitude values. Doppler processing can use the summed I and Q sample stream directly to derive Doppler frequency estimates and/or compute FFT spectral data.
A possible implementation for interpolation filters operating on quadrature samples is described below. In one embodiment the interpolation filters and control logic can be implemented with an FPGA device. As provided above in reference to
At the start of an acquisition line, a write pointer 2504 and a read pointer 2506 in the dual port ram are reset to the top of the ram 2502. As each new sample comes in, the sample is written to the ram 2502 at the address of the write pointer 2504, which is then advanced to the next sequential location. When the write pointer 2504 reaches the end of the ram 2502, it wraps around to the beginning of the ram 2504 for the next write operation. The dual port rams 2502 are large enough to store samples for the maximum delay required by the steering and focusing needed for the acquisition line.
The input side of the dual port ram 2502, with the writing of each new sample and subsequent incrementing of the write pointer 2504, needs no channel unique control mechanism, since all channels can write their input data at exactly the same time and to the same addresses. The output side of the dual port ram 2502 uses independent channel control.
For dynamic focusing, the focal point is moved outward along the receive line at the half the speed of sound, so that focal point is always at the location of the echo being received. For a constant aperture, as the focal point moves out in range, the delay between the center and outer channels of the aperture decreases. With dynamic aperture, or constant f (i.e., focal length divided by the aperture size) number operation, the delay between inner and outer channels increases until the maximum aperture is reached, then the delay decreases.
Using dynamic aperture and dynamic focus with the dual port ram delay scheme, yields the following operation of the dual port ram pointers 2504, 2506: The center channel is delayed by the maximum delay amount (the amount for the full aperture) by letting the write pointer 2504 move ahead of the read pointer 2506 until the delay is achieved. At that point, the read pointer 2506 is moved ahead at the same rate as the write pointer 2504. An outer channel's initial delay is set by letting the write pointer 2504 move ahead of the read pointer 2506 by the appropriate amount. This initial delay offset can be less than the offset of the read 2506 and write pointers 2504 of the center channel. At this point, the read pointer 2506 is moved ahead at the same rate as the write pointer 2504 until the channel is made active in the aperture.
After a channel is made active in the aperture, its delay gradually increases with time to approach that of the center channel. This is accomplished by occasionally not moving the read pointer 2506 ahead when the write pointer 2504 is moved. This increases the offset between the read 2506 and write pointer 2504 gradually with time.
The above operation can be directed with only two binary state control signals as shown in
The second control signal (CE) 2608, 2608 a merely specifies when the channel's output becomes active, so that it participates in the summation of all active channels. This can be accomplished by the CE signal 2608, 2608 a controlling the ‘clear’ input of the final output register of the interpolators. A channel is made active in the aperture according to when its element sensitivity pattern allows it to receive the returning echoes with less than some threshold amount of attenuation. This time must be consistent with the initial delay time implemented by the first control signal. It should be noted that the CE signal 2608 specifies the time a channel becomes active in terms of the number of quadrature samples from the start of the acquisition line. This is because when a channel first participates in the sum of channels, it must contribute a quadrature pair. In the case of the Fc*2 sample clock, there are two clocks for every quadrature sample pair.
For the center channel, RPE 2606 is held low for the maximum delay time needed. This allows the write point 2504 to move ahead while the read point 2506 stays put. After the delay time is reached, RPE 2606 is set high (true) to allow the read pointer 2506 to advance at the same rate as the write pointer 2504. Since there is no dynamic focus required for the center channel, RPE 2606 remains high for the remainder of the acquisition line. The center channel CE signal 2608 brings the channel active shortly after the delay time is reached. The offset is to allow the shift register and register used for the interpolation filter to fill. The CE signal 2608 then removes the clear on the output register so the channel's data can enter the summation bus.
For the outer channel, RPE 2606 a is held low for only a short time, since its initial delay is much shorter than the center channel. Then RPE 2606 a is set high, allowing this delay to be maintained until the channel is made active. At that time, the RPE signal 2606 a is set low for a single clock cycle occasionally to implement the dynamic focus pattern. The CE signal 2608 a removes the clear on the output register when the channel can participate in the summation.
Referring back to
The input for the interpolation filters comes from the read address of the dual port ram 2502, which typically advances by one sample (I or Q) for each Sample Clk (Fc*2). When a read is performed of the dual port ram 2502, the sample is input to an eight sample shift register 2508, which holds the last eight samples read. If the read operation of the dual port ram 2502 is not enabled (RPE low), then no data enters the shift register 2508, and the read pointer 2506 is not advanced. The shift register 2508 still holds the last eight samples, and no samples are lost when the read pointer 2506 does not advance; the read pointer 2506 simply falls further behind the write pointer.
Every two sample clock cycles, the samples in the shift register 2508 are transferred in parallel to the inputs of the interpolation filter multipliers 2510. There they remain for the two multiply/accumulate operations that generate the I and Q outputs. When no dynamic focusing is occurring, the samples moved to the multiplier inputs shift forward in time by two samples for each center frequency period. The filter then outputs an I and Q sample for each period of the center frequency. With dynamic focus, occasionally the read cycle of the dual port ram is disabled, and the samples moved to the multiplier inputs shift forward by only one sample. This allows the interpolation point to move forward in time by less than a full period of the center frequency. With dynamic focus on an outer channel, the interpolation point is gradually moving back in time, towards the same time as the center channel.
The coefficients used by the interpolation filters are stored in a small ram 2512, which can be loaded by the system CPU. The ram 2512 can hold 32 sets of coefficients, 16 for the I interpolation point and 16 for the Q interpolation point. The coefficients are selected by five address lines, four of which are control lines that come from the control ram 2602. These four lines must provide a new address every other sample clock (Fc*2). The other line selects the I or Q coefficient set for the interpolation point chosen, and can be toggled with the operation of the filter, producing an I and Q sample every period of the center frequency. Finally, the output register 2514 for the interpolation filter holds the output samples before they enter the summation bus. This register's clear input is controlled by the CE control line. This allows a channel to be disabled from contributing to the sum bus until the interpolation output is valid.
Another way to implement the interpolation filters, phase rotation and dynamic apodization is shown in
The start of the read enable signals 2532, 2534 of each FIFO is delayed by a number of receive clock cycles equal to the initial coarse delay value 2536 required for each channel. If the read enable signal 2532, 2534 is held low while data is written into the FIFO 2528, 2530, the read out of the FIFO will be suspended and the coarse delay 2536 will increase. When the read enable signal 2532, 2534 goes high, the coarse delay 2536 that is applied remains constant. To align echoes in the signals from the center and the outer edge of the aperture, the center signals will be delayed a period of time before they can be summed with the signals from the outer edge. The delay value for sampled data at the center of the aperture is greater than that of the outer edges.
Dynamic receive focusing requires a control signal DF 2538 which goes high when the interpolation filter index 2540 needs to be changed. The interpolation filter index 2540 is a modulo 16 number ranging from 0 to 15. The interpolation filter index 2540 will decrease when the interpolation point has shifted by 1/16 wavelength. When the interpolation filter index 2540 decreases from 0 to 15, the FIFO read enable signal 2532, 2534 will go low for one clock cycle, to increase the coarse delay 2536 by one.
The fine delay is implemented by interpolation. In this example, the interpolation filters are implemented as systolic FIR filters with 4 taps 2542, 2544, 2546, 2548. There are 16 sets of coefficients for the 16 interpolation points. Each interpolation point has 4 coefficients 2550, 2552, 2554, 2556. By interleaving the I and Q samples and operating the filter at twice the receive clock frequency, the same interpolation filter can be used for both the I and Q samples. Different sets of coefficients are used for the I and Q interpolation, since the I and Q samples acquired by the ADCs are sampled at different points in time but are interpolated to the same point in time. To correct the sampling offset, the interpolation filter index for the Q samples will be offset from that of the I samples by 4. The coefficients used in the interpolation filter can alternate between I coefficients and Q coefficients by switching the address of the RAM 2558 which stores the coefficients. The interpolation filter indices are represented by the address counters 2560 for the coefficients. The address counters 2560 for the I and Q coefficients decrease by one when the DF signal 2538 goes high for one clock cycle. The output of the interpolation filter 2560 is I/Q interleaved.
The interpolated signals are fed to the phase rotation stage 2564, 2566 shown in
For dynamic apodization, the outputs of the phase rotation 2568, 2570 are multiplied by a factor which is dynamically changed during receive. Also if the multiplication factor in a channel is set to zero, the channel does not contribute to the aperture. This way, dynamic aperture updating is achieved. I and Q samples are interleaved through a multiplexer (MUX) 2572 to a common multiplier, which reduces the multiplier resources required.
Multi-Line Beamforming with Interpolation Filters
The use of interpolation filters for beamforming allows multi-line scanning. In multi-line scanning, several receive lines are reconstructed in the same transmit beam, as shown in
Since the adjacent receive scan lines in a multi-line scan have only small changes in the individual delays for each channel, the interpolation filter delay implementation allows all lines to be processed concurrently. This method works with bandwidth sampling, where the interpolation filters can be operated at a rate higher than the sample rate, as is shown in the exemplary conceptual implementation of the interpolation filter method for an individual channel in
For 3-1 multi-line scanning, as shown in
The output of the filter operations, as shown in
In the exemplary receive beamforming methods described above, the output is a digital data stream of samples representing the sampled RF data along a reconstruction line. This stream is derived by summing the data samples from all receive channels that participate in the receive active aperture. The RF data stream can be captured in a buffer with sufficient storage to hold an entire ray line. This same buffer can be used for synthetic aperture acquisitions, and can be summed with the RF data from the second half of the receive aperture as it exits from the summation circuitry.
For Nyquist or bandwidth sampling schemes with no down-conversion, the summed RF data stream exits the beamformer as a raw RF stream. This data stream can be converted to a different format using a pair of complimentary 90 degree phase difference filters, often referred to as Hilbert transform filters. These filters effectively band-pass the RF signal and down-convert it at the same time to baseband quadrature data streams. These baseband I and Q data streams can then be combined to provide echo amplitude data for 2D imaging, or processed further for Doppler blood flow detection. The Hilbert transform filters can also be used to selectively filter and process a portion of the received signal spectrum, as is needed for harmonic imaging, or frequency compounding. In the case of frequency compounding, the filters can be time multiplexed to produce interleaved output samples from different frequency bands of the spectrum.
Referring back again to
As is apparent from the above description, the control of the beamforming process can be complex, and a method of handling this complexity is to encode all the information prior to real-time scanning in memory blocks used to control the hardware. The beamformer controller's task is then reduced to ‘pointing’ to the appropriate portion of the memory block to retrieve the information needed for a beamformer event. Setting up the beamformer for a specific mode of operation is then accomplished by loading all the memory blocks with parameter information, then programming the various beamformer events with their respective pointers into the controller's state machine. To perform the scanning mode, the controller is then told to run, and steps through the events for an entire frame of acquisition data. At the end of the frame, the controller looks for a stop signal, and if none is found, repeats the whole sequence again.
Embodiments of the exemplary ultrasound system are capable of very high acquisition frame rates in some modes of operation, in the range of several hundred frames per second or higher. Just as with other embodiments according to the present invention, exemplary embodiments process displayed ultrasound image information at 30 fps or lower, even if the acquisition rates are much higher through the use of asynchronous processing as described in reference to
Also as previously described, the signal processing hardware/software has random access to the RF memory buffer, and accesses the RF data from a single acquisition frame to produce the displayed estimate data. In this exemplary embodiment, the maximum frame rate for signal processing and display is 30 fps, which is typically set by a timer, which signals the signal processing task every 1/30th of a second. When processing of a new display frame is complete, the signal processing/display task waits for the next 1/30 of a second time tick. At that time, the signal processing task reads the ‘Last Frame’ pointer from the Write Controller to see if a new frame is available. If the ‘Last Frame’ pointer has not advanced from the previously processed frame, signal processing does nothing, and waits for the next 1/30 of a second tick. If the ‘Last Frame’ pointer has changed, signal processing begins on the frame indicated by the pointer. In this manner, signal processing always starts on a 1/30 second tick, and always works on the most recent frame acquired. If acquisition is running much faster than 30 fps, then the ‘Last Frame’ pointer will advance several frames with each signal processing action.
After the system has been put in freeze, the RF frames stored in the memory buffer can be processed at any desired rate, up to the original acquisition rate. One simply calculates how many RF frames to advance in 1/30th of a second, which is computed as a floating point value that can vary from a fraction less than one to as many frames as occurred in 1/30th of a second during real time acquisition. With each 1/30th of second tick, signal processing accumulates the frame advance value, until an integer boundary, is crossed. At that time, signal processing processes the frame which is that integer boundary number of frames ahead of the last frame it processed.
Synthetic aperture beamforming is also supported by this memory buffer scheme. In this case, the various lines which make up the synthetic aperture are acquired into the memory buffer sequentially, so that the size of an RF storage frame increases. This is simply a different parameter for the Write Controller, which keeps track of how many lines are written per acquisition frame. For readout, signal processing then combines the multiple RF lines in a synthetic aperture to produce the final result.
The RF data for cineloop playback also provides for re-processing the data in different ways, bringing out new information. For example, the wall filters for color flow imaging can be changed during playback, allowing optimization for the specific flow conditions. Second, for the researcher who wants to work with RF data, the buffer memory can dumped to an external storage device, providing multiple frames of RF data to analysis. Finally, as a diagnostic tool, the buffer memory can be loaded with test RF data from the CPU, allowing debug, analysis and verification of the signal processing methods.
For the Nyquist sampled beamforming method, down-converted quadrature sampled data is derived from the RF data for amplitude detection and Doppler processing. This can be obtained with complimentary phase FIR filters that are designed to have a 90 degree phase difference over the frequencies in the pass band. These filters can also down-convert the sample stream to a lower sample rate, provided the output sample rate is still sufficient to sample the range of frequencies in the signal. To provide down-converted output samples, the filters operate on RF data that is shifted by an integral number of cycles of the center frequency of the spectrum. Alternately, different filters can be designed for non-integer number of cycle shifts to obtain smaller decimation ratios. A schematic design of an exemplary Hilbert filter, as are known in the art by one of ordinary skill, is shown in
The filters are designed by first computing a low pass filter designed using a windowing method. The filter length should be around 40 taps to insure a good response over a broad range of frequencies, and should be a multiple of the number of samples in the period of the center frequency of the RF data. For example, if the sample rate is 120 MHz and the center frequency is 30 MHz, there are 4 samples in the period of the center frequency and an appropriate filter length would be 40 taps (10 periods). The low pass coefficients are then multiplied by a sine and cosine function, whose frequency matches the center frequency. In the 30 MHz example, each period of the sine and cosine function has 4 samples.
To obtain down-converted samples, the filters are applied on samples that are shifted by an integral number of cycles of the center frequency. In the 30 MHz center frequency case (sampled at 120 MHz), the RF samples are shifted by 4 samples at a time, leading to a decimation ratio of 4 to 1. With this decimation ratio, the input signal is restricted to 100% bandwidth, otherwise, aliasing of the output samples will result.
To obtain smaller decimation ratios, the filters can use alternate coefficient sets to preserve the phase information. In the 30 MHz example, to achieve a decimation ration of 4 to 2, two sets of coefficients are used—one for 0 degrees phase, and another for 180 degrees phase.
These alternate coefficient sets are obtained by sampling the sine and cosine at the appropriate phase increments before multiplying with the low pass filter coefficients. In this case where the shift between output samplers is ½ the period of the center frequency, a simple method to provide the decimation rate is to leave the coefficients the same, and invert the sign of the filter output for the ½ period increments.
The pass band characteristics of the filters can be modified using different windowing functions. This may be desirable in applications such as harmonic imaging or tracking filters. Frequency compound can be achieved without additional filters for high decimation ratios, provided that the filters can operate at the input sample rate. For the 30 MHz example, two filters can be used with different center frequencies that operate on RF data at two sample shift increments. The filter block output a different filter result every two samples. The two interleaved I,Q samples from the different filters are then detected and summed together to produce a 4 to 1 decimated detected output.
An embodiment of the exemplary system interface to an array with up to 256 elements may be used to obtain ultrasound images. Table 4 shows exemplary depth range, field of view, frame rate in B-Mode and frame rate in color flow imaging (CFI) for acquiring images. These operating parameters can be used for the particular small animal imaging application described in the far left column. As would be clear to one skilled in the art, however, other combinations of operating parameters can be used to image other anatomic structures or portions thereof, of both small animal and human subjects.
A small animal subject is used and the animal is anesthetized and placed on a heated small animal platform. ECG electrodes are positioned on the animal to record the ECG waveform. A temperature probe is positioned on the animal to record temperature. The important physiological parameters of the animal are thereby monitored during imaging. The anesthetic used may be a for example isoflourane gas or another suitable anesthetic. The region to be imaged is shaved to remove fur. Prior to imaging, an ultrasound conducting gel is placed over the region to be imaged. The ultrasound array is placed in contact with the gel, such that the scan plane of the array is aligned with the region of interest. Imaging can be conducted “free hand” or by mounting the array onto a fixture to hold it steady.
B-Mode frame rates are estimated for the different fields of view indicated in Table 4. Higher frame rates are achievable with reduced field of view. Color flow imaging (CFI) frame rates are estimated for the indicated color box widths, with line density one-half that of B-mode, and with the B-mode image acquired concurrently.
Exemplary Depth Range, Field Of View, Frame Rate In B-Mode And
Frame Rate In Color Flow Imaging (CFI) For Acquiring Images
At least 180 fps
At least 60 fps
30 MHz center frequency
At least 100 fps
At least 30 fps
Mouse, shallow regions,
At least 190 fps
At least 80 fps
At least 70 fps
At least 25 fps
20 MHz center frequency
Unaliased velocities measurable with a 150 KHz PRF, for various center frequencies and angles are shown in Table 5 for Pulsed Wave (PW) Doppler.
Unaliased Velocities Measurable With 150 KHz PRF,
for Various Center Frequencies and Angles
A mouse heart rate may be as high as 500 beats per minute, or about 8 beats per second. As the number of frames acquired per cardiac cycle increases, the motion of the heart throughout the cardiac cycle can be more accurately assessed. The frame rate should be at least 10 frames per cardiac cycle, and preferably 20 for better temporal resolution. Therefore, in one embodiment frames are acquired at a rate of at least 160 frames per second, with a field of view large enough to include a long axis view of the mouse heart and surrounding tissue (10-12 mm). For example, using a 30 MHz linear array, the frame rate for a 12 mm field of view is about 180 frames per second. For smaller fields of view, the frame rates used are higher; (e.g., for a 2 mm field of view, with the 30 MHz linear array frame rates of over 900 frames per second can be used for viewing rapidly moving structures such as a heart valve).
The maximum velocities present in the mouse circulatory system (in the aorta) may be as high as 1 m/s in normal adult mice, but in pathological cases can be as high as 4-5 m/s. To acquire and display unaliased PW Doppler signals from the mouse aorta, the Pulse Repetition Frequency (PRF) for PW Doppler must be relatively high. In the exemplary system, PW Doppler mode PRFs as high as 150 KHz are used, which for a center frequency of 30 MHz and a Doppler angle of 60°, allows for unaliased measurement of blood velocities of 3.8 m/s.
The frame rate for B-Mode Imaging is determined by the two-way transit time of ultrasound to the maximum depth in tissue from which signals are detected, the number of lines per frame, the number of transmit focal zones, the number of lines processed for each transmit pulse and the overhead processing time between lines and frames. Images obtained with different transmit focal zone locations can be “stitched” together for improved resolution throughout the image at the expense of frame rate, which will decrease by a factor equal to the number of zones. Selection of lower or higher transmit center frequencies for increased penetration, or increased resolution, either user selectable or automatically linked to transmit focal zone location. Multi-line processing, which involves the parallel processing of ultrasound lines, can be used to increase frame rate.
PW Doppler features include a PRF range from about 500 Hz to about 150 KHz, alternate transmit frequency selection, the selection of range gate size and position, the selection of high-pass filter cut-off, and duplex mode operation in which a real-time B-Mode image is displayed simultaneously with the PW Doppler mode may be the same as the transmit frequency used in B-Mode, or it may be different. The ability to steer the PW Doppler beam is dependent on the frequency and pitch of the array used, and the directivity of the elements in the array, as would be appreciated by one skilled in the art. For an array with a pitch of 75 microns and operating in PW Doppler mode at a transmit frequency of 24 MHz, the beam may be steered up to approximately 20°. For this array, larger steering angles would result in unacceptably large grating lobes, which would contribute to the detection of artifactual signals.
Color flow imaging (CFI) can be used to provide estimates of the mean velocity of flow within a region of tissue. The region over which the CFI data is processed is called a “color box.” B-Mode data is usually acquired nearly simultaneously with the Color Flow data, by interleaving the B-Mode lines with Color Flow lines. The Color Flow data can be displayed as an overlay on the B-Mode frame such that the two data sets are aligned spatially. CFI includes a PRF range from about 500 Hz to about 25 to 75 KHz, depending on the type of array. With 40 MHz center frequency and 0° angle between ultrasound beam axis and velocity vector, maximum unaliased velocity will be about 0.72 m/s. Beam steering can depend on the characteristics of the array, (specifically the element spacing), the transmit frequency, and the capabilities of the beamformer; e.g., steering may not be available at the primary center frequency, but may be available at an alternate (lower) frequency. For an array with a pitch of 75 microns and operating in CFI mode at a transmit frequency of 24 MHz, the beam can be steered up to approximately 20°. Larger steering angles would result in unacceptable grating lobe levels. Color flow imaging features can include the selection of the color box size and position, transmit focal depth selection, alternate frequency selection, range gate size selection, and selection of high pass filter cut-off. Power Doppler is a variation of CFI which can be used to provide estimates of the power of the Doppler signal arising from the tissue within the color box. Tissue Doppler mode is a variation of CFI in which mean velocity estimates from moving tissue are provided. Multi-line processing is a method which may be applied to the CFI modes, in which more than one line of receive data is processed for each transmit pulse transmitted.
The beamformer may be capable of supporting modes in which 2-D imaging and Doppler modes are active nearly simultaneously, by interleaving the B-Mode lines with the Doppler lines. 3-D imaging, as known to one of ordinary skill in the art, utilizes mechanical scanning in elevation direction.
Throughout this application, various publications are referenced. The disclosures of these publications in their entireties are hereby incorporated by reference into this application in order to more fully describe the state of the art to which this invention pertains.
Unless otherwise expressly stated, it is in no way intended that any method set forth herein be construed as requiring that its steps be performed in a specific order. Accordingly, where a method claim does not actually recite an order to be followed by its steps or it is not otherwise specifically stated in the claims or descriptions that the steps are to be limited to a specific order, it is no way intended that an order be inferred, in any respect. This holds for any possible non-express basis for interpretation, including: matters of logic with respect to arrangement of steps or operational flow; plain meaning derived from grammatical organization or punctuation; and the number or type of embodiments described in the specification.
It will be apparent to those skilled in the art that various modifications and variations can be made in the present invention without departing from the scope or spirit of the invention. Other embodiments of the invention will be apparent to those skilled in the art from consideration of the specification and practice of the invention disclosed herein. It is intended that the specification and examples be considered as exemplary only, with a true scope and spirit of the invention being indicated by the following claims.
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|U.S. Classification||600/447, 702/176, 702/79, 702/75, 702/177|
|Cooperative Classification||A61B8/565, A61B8/56, G01S15/8915, G01S7/5202, G01S7/524, G10K11/346, G10K11/341, G01S7/52017, G01S7/52095, G01S15/8956, G01S15/8927, G01S7/52088, G01S7/52034, G01S7/526, G01S15/8997, G01S15/8959|
|European Classification||G10K11/34C4, G01S7/52S, G10K11/34C, G01S15/89D3C, G01S7/52S14B1, G01S15/89D11, G01S15/89D1C, G01S15/89D1C5, G01S7/52S14F, G01S7/52S2E|
|Jan 15, 2010||AS||Assignment|
Owner name: VISUALSONICS INC., CANADA
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:MEHI, JAMES;DAIGLE, RONALD E.;BRASFIELD, LAURENCE C.;ANDOTHERS;REEL/FRAME:023797/0344;SIGNING DATES FROM 20070115 TO 20070209
Owner name: VISUALSONICS INC., CANADA
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:MEHI, JAMES;DAIGLE, RONALD E.;BRASFIELD, LAURENCE C.;ANDOTHERS;SIGNING DATES FROM 20070115 TO 20070209;REEL/FRAME:023797/0344
|Feb 12, 2010||AS||Assignment|
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