|Publication number||USRE30586 E|
|Application number||US 06/008,775|
|Publication date||Apr 21, 1981|
|Filing date||Feb 2, 1979|
|Priority date||Feb 2, 1979|
|Publication number||008775, 06008775, US RE30586 E, US RE30586E, US-E-RE30586, USRE30586 E, USRE30586E|
|Inventors||Adrian P. Brokaw|
|Original Assignee||Analog Devices, Incorporated|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (2), Non-Patent Citations (2), Referenced by (19), Classifications (4)|
|External Links: USPTO, USPTO Assignment, Espacenet|
.Iadd.This is a continuation of U.S. Pat. application Ser. No. 799,760 filed May 23, 1977, which is a Reissue application of U.S. Pat. No. 3,887,638 dated June 3, 1975. .Iaddend.
1. Field of the Invention
This invention relates to regulated DC voltage supplies. More particularly, this invention relates to solid-state (IC) regulators capable of maintaining a substantially constant DC output voltage in the face of temperature variations.
2. Description of the Prior Art
Conventional prior-art regulated voltage supplies commonly have included an internal reference source and an error amplifier arranged to compare the reference voltage with a pre-set fraction of the regulated DC output voltage. The output of the error amplifier is directed to a control element, such as a controllable impedance or the like, arranged to adjust the output DC voltage so as to maintain the two compared voltages equal. Fluctuations in the DC output voltage are thereby reduced.
In transistorized voltage-regulator circuits, the reference source typically has been a Zener diode. However, as is known in the art, Zener diodes have certain inherent characteristics which undesirably restrict the capability of a voltage regulator. An alternative type of solid-state regulator has been developed which does not use a Zener diode reference, relying instead on certain temperature-dependent characteristics of the base-to-emitter voltage (VBE) of a transistor.
U.S. Pat. No. 3,617,859 discloses a circuit of the latter type which includes a diode-connected transistor operated at one current density, and a second transistor operated at a different current density. These two transistors are interconnected with associated circuitry so as to develop a voltage proportional to the difference in the respective base-to-emitter voltages (ΔVBE). This difference voltage has a positive temperature-coefficient (TC), and is connected in series with the VBE voltage of a third transistor, having a negative TC, to produce a composite resultant voltage which serves as the output of the regulator. Since the temperature coefficients of the two individual voltages are of opposite sign, the output voltage can be made relatively insensitive to temperature variations by proper choice of certain parameters.
Although such regulators based on the VBE characteristic of transistors have significant advantages, the circuit arrangements proposed and used heretofore suffer from serious limitations. It is a principal object of the present invention to provide a solid-state voltage regulator which avoids or significantly minimizes such limitations of prior art regulators.
In an exemplary embodiment of the present invention, to be described in detail hereinbelow, there is provided a two-transistor voltage-regulator circuit wherein the ratio of current densities of the two transistors is automatically controlled to a predetermined value (different from unity) by a negative feedback arrangement. A voltage corresponding to the ΔVBE of the two transistors is developed, having a positive TC, and this voltage is connected in series with the VBE voltage of one of the two transistors, having a negative TC. The circuit parameters are selected so that the resultant combined voltage has a very low temperature coefficient. The regulator of this invention provides important advantages over previous regulators, as will be outlined hereinbelow in describing specific embodiments of the invention.
FIG. 1 is a circuit diagram of one embodiment of the present invention;
FIG. 2 is a circuit diagram of a modified arrangement to provide higher regulated output voltages;
FIG. 3 shows a further circuit arrangement modified to reduce base current effects; and
FIG. 4 is a circuit diagram of a voltage reference source including means to establish bias levels and to provide current limiting.
Referring now to FIG. 1, there is shown a circuit diagram representing basic components and interconnections of a regulated voltage source in accordance with the present invention. The circuit includes a pair of transistors Q1 and Q2 which are supplied with operating voltages by positive and negative voltage lines 10 and 12. The emitter of the left-hand transistor Q2 is coupled through two series-connected resistors R2 and R1 to the negative voltage line 12, and the emitter of the other transistor Q1 is connected to the common junction 18 between the two resistors.
The invention proceeds on the concept of (1) developing a first voltage, having a positive temperature coefficient (TC), (2) combining that voltage serially with a second voltage having a negative TC, and (3) relating the two temperature coefficients in a complementary sense such that the resultant composite voltage has a very low TC, approximately zero. To develop the positive TC voltage, the two transistors Q1 and Q2 are controllably operated at markedly different current densities (i.e., referring to density of current flowing through the emitters), and a voltage is produced proportional to the difference in the two transistor base-to-emitter voltages, referred to as ΔVBE.
In the specific embodiment disclosed herein, transistor Q2 is operated at a smaller current density than the other transistor Q1. Such difference in current densities can be produced (1) by using identical transistors operating at unequal currents, (2) by using transistors having unequal emitter areas operating at equal currents, or (3) by some combination of the latter two arrangements. Simply by way of example, in the described embodiment the emitter areas of the transistors Q1 and Q2 are specified as A and nA respectively, with n being greater than one, and the currents through the two transistors are equal.
The transistor currents are forced to be equal by a negative feedback arrangement comprising current-sensing means in the form of equal-resistance load resistors RL1 and RL2 in the collector circuits of transistors Q1 and Q2. These load resistors develop voltages which are proportional to the respective collector currents, and which are directed by leads 24, 26 to the input terminals of a high-gain operational amplifier 28. The output of this amplifier is connected to a common base line of the two transistors Q1 and Q2, and also to an output terminal 30 presenting the DC output voltage of the regulator. The amplifier 28 drives the common base line until the voltage drops across the load resistors RL1 and RL2 are equal, thereby forcing the transistor collector currents to be equal. By using well matched transistors Q1 and Q2, the emitter currents also will be equal.
Since the bases of transistors Q1 and Q2 are connected together, the difference voltage ΔVBE will appear across emitter resistor R2, and the current through that resistor thus will be directly proportional to ΔVBE. The series connected resistor R1 also carries this emitter current, and additionally carries the emitter current of the second transistor Q1. Since the latter emitter current is directly proportional to the first emitter current (actually equal, in this particular example), it will be evident that the total current through resistor R1, and hence the voltage across that resistor, also will be directly proportional to ΔVBE.
It has previously been established that, for two transistors operating at different current densities, the difference in base-to-emitter voltage is given by:
ΔVBE =kT/q ln J1 /J2
where T is absolute temperature, k is Boltzman's constant, q is the charge of an electron, and J1 /J2 is the ratio of the transistor current densities. Accordingly, the voltage developed across resistor R1 is independent of absolute emitter current, and is a linear function of absolute temperature with a positive temperature coefficient.
As is evident from the circuit lay-out, the voltage across resistor R1 is in series with the VBE voltage of transistor Q1 and the resultant composite voltage constitutes the DC output voltage on terminal 30. Since VBE has a negative temperature coefficient, changes in that voltage with temperature tend to complement the positive TC changes in the voltage across resistor R1.
To approach zero TC, the output voltage at the regulator terminal 30, with respect to the negative voltage line 12, should be set approximately to the value of the energy band-gap voltage (VGO), extrapolated to 0°. For silicon, this extrapolated voltage is 1.205V. A slightly higher voltage produces superior results. It can be shown mathematically, based on certain reasonable assumptions, that for zero TC the output voltage should be set at:
VOUT =VGO +(m-1)kTo/q
where m≅1.5 and To is the nominal operating temperature.
This voltage VOUT can be adjusted to the desired value by proper selection of resistor R1, such that the resistive voltage drop complements the VBE of Q1 to optimize the total (sum) voltage for zero TC.
When the DC output voltage (VOUT) at terminal 30 drops below the pre-established optimal level, the ratio of collector currents I2 /I1 is larger than the ratio of load resistors RL1 /RL2 (i.e., larger than one), so that the input to amplifier 28 is positive. This causes the amplifier output to increase, so as to return the voltage VOUT back up to the optimal level. If the DC output rises above optimal, the feedback action of amplifier 28 will have the opposite effect. Thus the voltage-control circuit continuously holds the DC output voltage at the proper level to provide a very low overall temperature coefficient, close to zero.
In some applications, DC output voltages higher than the energy band-gap voltage may be required. FIG. 2 shows an arrangement for that purpose. The basic operation of this circuit is similar to that of FIG. 1, and like reference numerals are used throughout for corresponding elements. However, FIG. 2 differs in that the output of amplifier 28 is connected to a voltage-dividing network comprising two series-connected resistors hR3 and R3. The common junction terminal 32 of these resistors provides a voltage which is a predetermined fraction of VOUT, and this voltage is directed to the commonly connected bases of transistor Q1 and Q2.
As before, the amplifier 28 drives the transistor bases until their collector currents are equal. By proper selection of circuit parameters, the reference voltage (VREF) at this stable point can be set to be optimum for achieving zero TC. The output voltage (VOUT) then will be some predetermined multiple of VREF, specifically (h+1)·VREF.
This arrangement of FIG. 2 provides a quite accurate result. It is degraded only a small amount due to the base current of the transistors. This base current is relatively low, and in any event the positive TC of the transistor beta tends to act with the positive TC of the emitter current to stabilize the base current and reduce any drift.
Where further reduction of such small drift effects may be desirable, a controlled beta PNP may be used to reflect the base current from a pair matched to Q1 and Q2 and connected in cascode with them into the base of Q1 and Q2. Alternatively, the base of transistors Q1 and Q2 can be connected together through a resistor R4, as illustrated in FIG. 3. Here, the voltage across resistor R2 is no longer ΔVBE, since the bases are no longer at the same voltage. However, it can be shown that this arrangement may, with matched betas, produce the basic regulation of the FIG. 2 embodiment, but with reduced drift due to base current, providing R4 is selected such that:
R4 =hR3 /h+1×R2 /R1
To take into account the possible effects of base spreading resistance of the two transistors, further analysis indicates that the base-connecting resistor R4 should be selected such that:
R4 =Rb1 C1 (1 +1/C2 (C1 +1))-Rb2
where Rb1 and Rb2 are the base spreading resistances of transistors Q1 and Q2 ; C1 =Ie1 /Ie2 (emitter currents of Q1 and Q2); and C2 =R1 /R2
The above-derived expression for R4 also indicates that the use of a base-connecting resistor may be helpful in the basic circuit configuration of FIG. 1. For R4 to be zero, Rb2 must be larger than Rb1 ; typically, however, Q2 is the larger transistor with a very low base resistance, and design considerations thus suggest that the base resistance of Q1 should be minimized. It may be possible to correct for the effect with a pinched base resistor in series with the large transistor.
Voltage-regulated supplies in accordance with the present invention have a number of important and beneficial features. Foremost, such voltage supplies provide a highly stable output voltage in the face of changing ambient temperature. Only two matched active elements are required, rather than three as in the above-identified U.S. Pat. No. 3,617,859. Advantageously, the reference voltage in the disclosed circuits appears in the control loop at a point with a high impedance, so that it can readily be driven. Moreover, the reference voltage may be multiplied as desired to produce output voltages higher than the band-gap voltage, by means of a single control loop, and without stacking junctions. In the FIG. 2 configuration, the reference voltage can first be adjusted to minimize temperature coefficient, and then the output voltage can separately be adjusted to a predetermined voltage without affecting the temperature coefficient. The basic circuit is convenient to trim by adjusting a single resistor (R1). Finite beta and beta drift does not result in uncorrectable errors; only beta matching is required.
Referring now to FIG. 4, there is shown a voltage reference source including transistors Q1 and Q2 used to establish the reference voltage in the manner generally as described hereinabove. In this circuit, these transistors are driven so that they will operate at equal collector currents. Neglecting R23, for the moment, the bases of these transistors are driven from the circuit output by the voltage divider consisting of R31 and R24. The output current is provided by Darlington-connected transistors Q4 and Q7, which draw operating current from the input voltage terminal. The base of Q4 is driven by a bias current from Q18.
The circuit output voltage is controlled by adjusting the base voltage of Q4, so that Q4 and Q7 form a voltage follower. A voltage drop provided by Q3 approximately matches the VBE of Q4, with R27 and Q15 providing a voltage drop matching other circuit voltages. The base voltage of Q4 is controlled by the emitter follower Q12 which is driven by Q1 and Q14.
In operation the collector current of Q2 drives the base of Q11 negative. Acting as an emitter follower, Q11 turns on Q13 and drives it until its collector current approximately equals the collector current of Q2. The base of Q13 connects to Q14, a matching transistor. Since R25 and R26 are also matched, the collector current of Q14 will approximately equal that of Q13 and hence of Q2. If the collector current of Q2 exceeds the collector current of Q1, Q14 will drive the base of Q12 positive. Alternatively, if the collector current of Q1 exceeds the collector current of Q2, it will also exceed that of Q14 and will, therefore, drive the base of Q12 negative. The circuit output voltage will follow the base voltage at Q12 as previously explained.
The emitter area of Q2 is eight times larger than that of Q1. When the voltage at the base of Q1 and Q2 is low, the current through R21 and R22 is low. The resulting voltage drop across R22 will be low, and the base-emitter voltages of Q1 and Q2 will be nearly equal. As a result of the area ratio mismatch the emitter current in Q2 will be nearly eight times the current in Q1. This current mismatch will cause Q14 to drive the base of Q12, and, thereby, the output - positive.
If the base voltage applied to Q1 and Q2 is made larger, the current through R21 and R22 will also be larger. At a sufficiently high base voltage the voltage drop across R22 will limit the current in Q2, and it will drop below the current in Q1. The excess collector current in Q1 will drive the base of Q12 negative, and with it the circuit output.
Between these two extremes of base voltage there will be a voltage at which the collector currents of Q1 and Q2 are equal. At this voltage the current in Q14 will balance the current in Q1 and the base of Q12 will be held at a voltage which maintains the circuit output and the Q1 - Q2 base voltage constant. Changes in output loading or other disturbances which tend to change the output voltage will change the voltage on the bases of Q1 and Q2. This will disturb their collector current balance so as to drive Q12 to restore the output voltage. This control loop forcing the collector currents of Q1 and Q2 to be equal satisfies the condition, previously described, to hold constant C1 =1.
With the collector currents of Q1 and Q2 forced to be equal, the voltage drop across R22 will be (kT/q) ln J1 /J2 =(kT/Q) ln 8. The current in R21 will be just twice that in R22 so that the voltage across R21 will be proportional to the drop across R22. Therefore, the voltage at the base of Q1 which results in the balance condition is the sum of the VBE of Q1 and the temperature-dependent voltage on R21. This voltage is set (by selecting the ratio of R21 and R22) so that this voltage is just above the bandgap voltage and satisfies the conditions previously outlined for zero temperature coefficient.
The stabilized base voltage of Q1 is a fraction of the circuit output voltage determined by R31 and R24. The output voltage is, therefore, a temperature stable multiple of the bandgap voltage determined by the resistor ratio. The interbase resistor R23 corrects for the offset and drift due to base current flow in R31. It also corrects for the base spreading resistance of Q1, as previously noted.
The voltage divider R28 and R29 is connected across the circuit output voltage. It is selected to have a Thevenin equivalent output voltage which differs from the circuit output voltage by the bandgap voltage. The equivalent resistance at the divider output is set at twice the resistance of R21. Transistor Q5 is designed to match Q1. As a result of the equivalent voltage and resistance applied across its base and emitter, its emitter and collector currents will be approximately equal to those of Q1. This current drives the common base of Q16 and Q17, a matched transistor pair. The matched emitter resistors, R32 and R33, force the emitter currents of Q16 and Q17 to be equal and raise the output impedance of Q16. This current mirror "reflects" the collector current of Q5 down through Q3, R27 and Q15. A small fraction of this current drives the base of Q4 which in turn drives Q7 and also supplies the current for Q13 and Q14. Since the current in the Q5, Q17, Q16 and Q3 path approximates the current in Q2, it is approximately half the current in Q13 and Q14 combined. This combined current is the majority of the emitter current in Q4. By making the emitter area of Q4 twice that of Q3, the current densities and hence the base-emitter voltages of Q3 and Q4 are made nearly equal. Therefore, the voltage at the top of R27 approximately equals the voltage applied to R25 and R26. The currents in R25, R26 and R27 are approximately equal so that the voltage drops across them are approximately equal. Similarly, Q15 is sized so that its emitter current density approximates that of Q13 and Q14. In this way the base voltage of Q15 is made nearly equal to the base voltage of Q13 and Q14. This equality is translated through the base-emitter voltage of the matched transistors Q11 and Q12 to the collectors of Q2 and Q1 . This keeps the collector voltages of these transistors approximately equal at all temperatures and bias conditions. This minimizes problems resulting from different base width modulation in Q1 and Q2 which might result from unbalanced collector voltage.
The bias voltage stabilization also keeps the free collector voltage of Q15 nearly equal to the base voltage. This helps to insure an equal split of the current in the forced beta transistor Q15 (beta=1). This current split ensures equal emitter currents in Q11 and Q12, thereby minimizing errors due to differences between their base currents.
The circuit as described so far would have a stable "off" state. The epitaxial layer FET portion of Q5 eliminates this possibility. The FET provides a small starting current that turns on the circuit when voltage is applied. Although it diverts some of the current from R28, it has only a small effect on the current delivered to Q17. This total current is determined largely by the voltage drop across the equivalent R28, R29 resistance. The slight change in Q5 VBE which results from the diverted current is a small fraction of the total voltage applied to R28 and R29.
The frequency stability of the output control loop is established by C36. This capacitance rolls off the open-loop gain to unity below the frequency at which excess phase shift in the PNP's might cause instability.
Output overload protection is provided by Q6 and R30. The output current flows through R30 and produces a small voltage drop across it. In the event of overload, this voltage will rise and drive Q6 on. As Q6 comes on it will divert the drive current from the base of Q4 into the load. As a result, the output current is limited to that necessary to drive Q6 on by way of R30.
The overall circuit consists of a current input amplifier which operated the control loop stabilizing the reference voltage. The amplifier input circuit, Q13 and Q14, is bootstrapped to the regulated output. This bootstrap connection minimizes the effects of power supply voltage variation on the amplifier which improves the overall supply voltage rejection of the circuit.
Although several preferred embodiments of the invention have been described hereinabove in detail, it is desired to emphasize that such details have been disclosed for the purpose of illustrating the nature of the invention, and should not be considered as necessarily limiting of the invention which can be expressed in many modified forms to meet particular requirements.
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