|Publication number||USRE36013 E|
|Application number||US 08/870,638|
|Publication date||Dec 29, 1998|
|Filing date||Jun 6, 1997|
|Priority date||Dec 10, 1993|
|Also published as||US5422529, US5736892, WO1995016305A1|
|Publication number||08870638, 870638, US RE36013 E, US RE36013E, US-E-RE36013, USRE36013 E, USRE36013E|
|Inventors||Thomas H. Lee|
|Original Assignee||Rambus, Inc.|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (39), Non-Patent Citations (2), Referenced by (14), Classifications (20), Legal Events (2)|
|External Links: USPTO, USPTO Assignment, Espacenet|
The present invention relates to a differential amplifier having high differential impedance and low common mode impedance.
In general, a differential amplifier, or difference amplifier, receives two signals as inputs, and outputs either the current or voltage difference between the input signals. One problem associated with differential amplifiers is that they exhibit common mode sensitivities. Common mode sensitivities are defined as the dependency of the output common mode voltage on the change in any parameter in the circuit, such as device mismatch. Some differential amplifiers have common mode sensitivities that are of the same order as the differential gain of the amplifier.
FIG. 1a illustrates a typical prior art high gain differential amplifier. The output is coupled from the drains of transistors T1 and T2. As is well known in the art, the incremental differential gain of the circuit illustrated in FIG. 1a is infinitely large if the current source drain loads are perfect, and if transistors T1 and T2 have zero differential output conductance. Although the incremental differential gain is infinite, the differential amplifier also exhibits infinite common mode sensitivity. The infinite common mode sensitivity is a result of the mismatch between the sum of the drain load currents, labeled IDL on FIG. 1a, and the value of the current source at the common source connection. If the drain load currents are not precisely equal, the common mode output voltage approaches the value of either Vdd or Vss.
To mitigate the problem of common mode sensitivity, a feedback bias circuit may be employed as shown in FIG. 1b. In the circuit of FIG. 1b, the common mode output voltage, labeled Va+Vb, is compared with a reference voltage labeled Vref. Any error detected between the desired and actual common mode output levels generates a correction signal that is fed back to control the current source at the common source connection, driving the output voltage to the correct value. Although use of a feedback loop solves the current source biasing problem, the feedback loop generates new problems. For example, the feedback loop introduces complexity to the circuit operation. In addition, a feedback loop inherently exhibits limited bandwidth, thereby placing constraints on the time required to obtain steady state operation when the differential amplifier is powered on and off. The ability to rapidly obtain steady state operation is particularly important in order to extend battery life in portable equipment where it is desirable to activate only those circuits that are in use at a particular moment in time.
Differential amplifiers, when overdriven, also have application for use in charge pump circuits. A charge pump is a circuit, for example, a CMOS circuit, which receives two signals as inputs and generates a voltage across the capacitor indicative of the phase difference between the two input signals. An example of a circuit that uses a charge pump is a phase-locked loop circuit. In such a circuit, a charge pump is used to integrate the phase error between the desired phase and the phase output. A perfectly lossless charge pump provides a theoretically infinite output if there is non-zero phase error.
A perfectly lossless charge pump enables the attainment of theoretically zero phase error (assuming that there are no other error sources), since any finite output can be obtained with a zero value input.
A lossy charge pump, however, requires a non-zero phase error in order to produce a finite output, and hence, a static phase error results. It is therefore desirable to reduce to an absolute minimum any loss in the charge pump.
An idealized prior art technique is shown in the MOS circuit of FIG. 2a. In this circuit, current sources are utilized as high-impedance loads. Transistors T1 and T2 are assumed to be driven by signals that are sufficiently large in amplitude to switch substantially all of the current 2I of the current source alternately through M1 and M2, so that a net current of I or -I flows through the capacitor. This differential circuit requires some means, however, to set the common-mode level of the output voltage (i.e., the average voltage of the drain terminals of T1 and T2).
One method to provide this is through the use of common-mode feedback bias control loop, as shown in the example of FIG. 2b. In the circuit shown in FIG. 2b, the DC value of the output voltage is compared with a reference voltage, and feedback is arranged to adjust the value of the current source to control the common-mode output voltage. However, stability for the common-mode loop is an issue. Additionally, some reference voltage (either explicit or implicit) is needed to set the value of the common-mode output voltage. Furthermore, the biasing circuit consumes substantially additional die area on a relative basis. Finally, in some applications, it is desirable for the common-mode value of the output to lie within fairly narrow bounds, even when the charge pump is turned off. Use of a common-mode loop is impractical in such a case, since, without power, there is no active loop and the value does not lie within narrow bounds.
Therefore, it is an object of the present invention to provide a differential amplifier that possesses low common mode sensitivities.
It is another object of the present invention to provide a s differential amplifier that operates down to low supply voltages.
It is another object of the present invention to provide a differential amplifier possessing a rapid transition from a disabled state to an enabled state.
It is another object of the present invention to provide a differential charge pump circuit that operates with low leakage.
It is another object of the present invention to provide a differential charge pump that operates down to low supply voltages.
Furthermore, it is an object of the present invention to provide a charge pump circuit, which provides well-defined bounds on the common-mode value of the output voltage when powered down.
These and other objects of the present invention are realized in an arrangement including a high gain low voltage differential amplifier exhibiting extremely low common mode sensitivities. The differential amplifier contains a first and second transistor each comprising a first terminal for receiving a supply current, a second terminal for dispensing the supply current, and a third terminal for controlling the amount of supply current flowing from the first terminal to the second terminal. Differential signals are input to the third terminal on the first and second transistors. A power supply, for providing electrical power to the circuit, generates Vdd and Vss voltages. A positive differential load resistance is generated and is connected from the power supply to the first terminal of the first transistor and to the first terminal of the second transistor. To offset the positive differential load resistance, a negative differential load resistance is generated and connects the power supply to the first terminal of the second transistor and to the first terminal of the first transistor. The output signal is generated from the first terminal of the first transistor and from the first terminal of the second transistor. Consequently, the output of the differential amplifier exhibits low common-mode impedance so as to reduce common-mode sensitivities.
For operation as a charge pump circuit, the differential amplifier contains at least one capacitor for generating a capacitance across the first terminal of the first transistor and the first terminal of the second transistor. Control signals are input to the third terminals of the first and second transistors so as to permit switching current, in both directions, from the positive differential load resistance and the negative differential load resistance across the capacitance.
Other objects, features and advantages of the present invention will be apparent from the accompanying drawings, and from the detailed description that follows below.
The present invention is illustrated by way of example and not limitation in the figures of the accompanying drawings in which like references indicate similar elements, and in which:
FIG. 1a shows a prior art idealized differential amplifier;
FIG. 1b shows a prior art differential amplifier with common-mode feedback bias;
FIG. 2a shows a prior-art differential charge pump;
FIG. 2b shows a prior-art differential charge pump with common-mode feedback bias;
FIG. 3 shows a high gain differential amplifier configured in accordance with the present invention;
FIG. 4 shows one embodiment of the differential charge pump of the present invention;
FIG. 5 shows another embodiment of the differential charge pump of the present invention.
A high gain low voltage differential amplifier exhibiting extremely low common mode sensitivities is described. The differential amplifier contains six metal oxide semiconductor field effect transistors (MOSFETs). The high gain and low common mode sensitivity characteristics of the differential amplifier are obtained through a load element. The load element includes a positive differential load resistance and a negative differential load resistance. In one embodiment, the negative and the positive differential load resistances are configured to offset one another. The load element possesses an ideally infinite differential resistance, and low common mode resistance of the order of the reciprocal transconductance of the MOS devices. Because the load element presents a low common mode impedance, the need for common mode feedback is eliminated.
The differential amplifier has application for use in a differential charge pump circuit. The high differential impedance of the differential amplifier allows the attainment of extremely small leakage, while a low common-mode impedance results in simplified biasing.
FIG. 3 shows a low voltage differential amplifier 300. The low voltage differential amplifier 300 contains six metal oxide semiconductor field effect transistors (MOSFETs). Although embodiments of the invention are described in conjunction with a complementary metal oxide semiconductor (CMOS) configuration, other devices, such as bipolar transistors may be used without deviating from the spirit and scope of the invention. Input signals, labeled Input1 and Input2 on FIG. 3, are input to the gates of transistors M1 and M2, respectively. The differential amplifier 300 is connected to a first power source, having a voltage of Vdd, and a second power source having a voltage of Vss. An output for the differential amplifier 300 is taken from the drains of transistors M1 and M2 labeled Va and Vb on FIG. 3. The differential amplifier 300 illustrated in FIG. 3 is a high gain differential amplifier exhibiting extremely low common mode sensitivities, on the order of unity. The high gain and low common mode sensitivity characteristics of S the differential amplifier 300 are obtained by coupling a high differential resistance, but a low common mode resistance, from the first power source to the drain of transistors M1 and M2.
The load element of the present invention comprises p-channel MOSFET transistors M3, M4, M5 and M6. The load element couples the first power source to the drains of transistors M1 and M2. The load element possesses an ideally infinite differential resistance, and low common mode resistance of the order of the reciprocal transconductance of the MOS devices. The transistors M3 and M6 are diode connected such that the gates are coupled to the drains of the respective devices. Consequently, transistors M3 and M6 together comprise a very low differential resistance.
The gate of transistor M3 is coupled to the gate of transistor M4, and the gate of transistor M5 is coupled to the gate of transistor M6. Because the gates of transistors M3 and M4 are connected, ideally transistor M4 mirrors the current of transistor M3. The drain of transistor M4 is coupled to the drain of transistor M2 so that the mirrored current through M4 flows through M2. By mirroring the current through transistor M3, which is responsible for the low differential resistance, no differential current is generated.
In effect, transistors M4 and M5 serve as a negative differential resistance. Similarly, transistor M5 mirrors the current of transistor M6. The current flowing through transistor M6 is provided at the drain of transistor M1. Effectively, transistors M3 and M6 constitute a positive differential resistance. Therefore, the high differential resistance is generated because the negative differential resistance, generated by transistors M4 and M5, cancels the positive differential resistance generated by transistors M3 and M6.
Because transistors M3-M6 present a low common mode impedance, the need for common mode feedback is eliminated. The output common mode level of the differential amplifier 300 is one p-channel source to gate voltage drop below the power supply, Vdd. If the current supplied to the differential amplifier 300 is eliminated, the common mode output voltage cannot drift far from the value retained when the circuit was active. Consequently, power up recovery upon reactivation occurs quickly because the differential amplifier 300 does not utilize common mode feedback bias which requires a feedback loop to settle. In a first embodiment, transistors M3-M6 are constructed substantially identical in size so that the negative differential load resistance cancels the positive differential.
In another embodiment of the differential amplifier of the present invention, transistors M3 and M6 are constructed substantially identical in size, and transistors M4 and M5 are constructed substantially identical in size. Unlike the first embodiment of the present invention, the transistor pair M4 and M5 is mismatched from the transistor pair M3 and M6. By mismatching the transistors comprising both the negative differential resistance and the positive differential resistance, a controlled gain other than infinity is realized. In addition, this embodiment preserves the attribute of low common mode sensitivity. For this embodiment of the differential amplifier, the negative differential resistance of transistor pair M4 and M5 does not precisely cancel the positive differential resistance of transistor M3 and M6, leaving a finite residual differential resistance. The finite residual differential resistance may comprise a positive or a negative resistance, depending on the relative magnitudes of the individual resistances comprising the load elements.
The embodiments of the present invention have use in a variety of applications. For example, the differential amplifier has application for use in a differential charge pump circuit. One embodiment of a low leakage CMOS charge pump circuit of the present invention, is illustrated in FIG. 4. For other embodiments, other types of devices may be used.
In the embodiment shown in FIG. 4, the load is formed of a quad of P-channel MOS devices. As explained above in conjunction with the differential amplifier 300, transistors M3 and M6 are diode-connected, and are located in parallel with cross-connected transistors M4 and M5. For use in a charge pump circuit, transistors M3, M4, M5 and M6 are all made to be substantially of the same size. An integration capacitance C1 is shown as a single device connected to the drains of the drive transistors M1 and M2.
The signals driving the gates of drive transistors M1 and M2 are assumed to be large enough in amplitude to switch substantially . .all of.!. the current . .2I.!. .Iadd.I .Iaddend.of the current source into the integrating capacitance C1.
The diode-connected devices, M3, M6, together present a positive differential resistance. The positive differential resistance by itself provides an undesirable leakage path for the integrating capacitance. To mitigate this leakage, the current through M3, M6 is canceled by the function of transistors M4, M5, which function can be viewed as that of a negative resistance. Thus, transistors M4, M5 cancel out the leakage of M3, M6 in the following manner.
Transistors M3 and M4 comprise a current mirror. Assuming ideal behavior, the two devices carry the same current. Because the drains of devices M3 and M4 are connected to opposite sides of the capacitance, the net contribution by this connection to the differential current is zero. By symmetry, the same reasoning applies to the current mirror formed by transistors M5, M6, so that the net differential resistance presented by transistors M3, M4, M5, M6 is ideally infinite, leaving only finite differential output resistance of the driving pair M1, M2, and the inherent capacitor leakage as differential loss mechanisms.
In practice, however, mismatches in the transistors cause departures from ideal behavior. Hence, it is desirable to choose the effective resistance of the diode-connected devices M3, M6 as high as possible such that any imperfect cancellation by respectively M4, M5, will have a proportionally smaller effect. The effect of finite differential output resistance of M1, M2, is usually negligible, because there is an inherent cascoding effect by M1, M2 on the current source. The charge pump switches . .all.!. the current . .2I.!. .Iadd.I .Iaddend.into the integrating capacitance by the control input signals in either a positive or negative direction. For example, if M1 is turned on and M2 is turned off, transistor M1 acts as a cascoding device for the current source (which source is assumed to be realized with a transistor or collection of transistors), boosting the effective impedance.
This argument applies symmetrically to the case where M2 is on and M1 is off. Hence, the leakage caused by M1, M2 is generally negligible. Further advantage of the invention is that the common-mode output voltage of the charge pump is simply lower than the positive supply voltage by one source-to-gate voltage of a P-channel device.
Therefore, if the charge pump is ever disabled, for example, by shutting off the current source, recovery is relatively quick because common-mode levels cannot be moved far from their equilibrium active values.
FIG. 5 shows a second embodiment of the charge pump circuit of the present invention. For this embodiment, an alternate connection of the integrating capacitance is employed. For some circuits, the most area-efficient capacitance is formed from the gate structure of a MOS transistor. Such capacitors require DC bias that is in excess of approximately one threshold voltage to maximize capacitance and to avoid excessive nonlinearities. In this embodiment, the biasing criteria is satisfied by splitting the capacitor into two equal capacitors and coupling each capacitor to Vss. By connecting each capacitor to Vss, this embodiment provides filtering of power supply (Vdd) noise. Any noise on the positive power supply that couples through the p-channel load devices is bypassed by the capacitors, reducing greatly the amount of the noise passed on to any subsequent stage.
Thus, a differential charge pump that employs a load device that possesses high differential impedance, but a very low common-mode impedance, is described. The high differential impedance allows the attainment of extremely small leakage, while low common-mode impedance results in simplified biasing.
In the foregoing specification, the invention has been described with reference to specific exemplary embodiments thereof. It will, however, be evident that various modifications and changes may be made thereto without departing from the broader spirit and scope of the invention as set forth in the appended claims. The specification and drawings are, accordingly, to be regarded in an illustrative rather than a restrictive sense.
|Cited Patent||Filing date||Publication date||Applicant||Title|
|US3771063 *||Mar 13, 1972||Nov 6, 1973||Calnor El Paso Inc||Bi-polar phase detector|
|US4156851 *||Oct 25, 1977||May 29, 1979||Winters Paul N||Constant-phase delay network|
|US4178554 *||Oct 25, 1977||Dec 11, 1979||Hitachi, Ltd.||Phase difference detector|
|US4285006 *||Jul 26, 1979||Aug 18, 1981||Mitsubishi Denki Kabushiki Kaisha||Ghost cancellation circuit system|
|US4291274 *||Nov 20, 1979||Sep 22, 1981||Tokyo Shibaura Denki Kabushiki Kaisha||Phase detector circuit using logic gates|
|US4364082 *||Feb 18, 1981||Dec 14, 1982||Hitachi, Ltd.||Phase detection circuit and automatic tint control circuit of color television receiver utilizing the same|
|US4479202 *||May 11, 1983||Oct 23, 1984||Tokyo Shibaura Denki Kabushiki Kaisha||CMOS Sense amplifier|
|US4506175 *||Aug 18, 1982||Mar 19, 1985||Rca Corporation||Digital phase comparator circuit producing sign and magnitude outputs|
|US4520321 *||Nov 30, 1982||May 28, 1985||Anritsu Electric Company Limited||Phase difference detector with phase inversion of one input signal when phase difference is small|
|US4623805 *||Aug 29, 1984||Nov 18, 1986||Burroughs Corporation||Automatic signal delay adjustment apparatus|
|US4635097 *||Feb 11, 1985||Jan 6, 1987||Sony Corporation||Method and system for processing digital video signal incorporating phase-correction feature|
|US4721904 *||Dec 17, 1985||Jan 26, 1988||Victor Company Of Japan, Ltd.||Digital phase difference detecting circuit|
|US4751469 *||May 22, 1987||Jun 14, 1988||Hitachi Ltd.||Phase coincidence detector|
|US4868512 *||Jul 31, 1987||Sep 19, 1989||Marconi Instruments Limited||Phase detector|
|US4870303 *||Jun 3, 1988||Sep 26, 1989||Motorola, Inc.||Phase detector|
|US4893094 *||Mar 13, 1989||Jan 9, 1990||Motorola, Inc.||Frequency synthesizer with control of start-up battery saving operations|
|US4904948 *||Mar 14, 1988||Feb 27, 1990||Fujitsu Limited||Phase comparator circuit|
|US4929916 *||Mar 10, 1989||May 29, 1990||Nec Corporation||Circuit for detecting a lock of a phase locked loop|
|US4963817 *||Jun 28, 1989||Oct 16, 1990||Fujitsu Limited||Method and apparatus for detecting a phase difference between two digital signals|
|US5095233 *||Feb 14, 1991||Mar 10, 1992||Motorola, Inc.||Digital delay line with inverter tap resolution|
|US5121010 *||Feb 14, 1991||Jun 9, 1992||Motorola, Inc.||Phase detector with deadzone window|
|US5128554 *||Feb 14, 1991||Jul 7, 1992||Motorola, Inc.||Opposite phase clock generator circuit|
|US5148113 *||Nov 29, 1990||Sep 15, 1992||Northern Telecom Ltd.||Clock phase alignment|
|US5164838 *||Apr 18, 1991||Nov 17, 1992||Pioneer Electronic Corporation||Time base error signal generating apparatus|
|US5179303 *||Jul 16, 1992||Jan 12, 1993||Northern Telecom Limited||Signal delay apparatus employing a phase locked loop|
|US5187448 *||Feb 3, 1992||Feb 16, 1993||Motorola, Inc.||Differential amplifier with common-mode stability enhancement|
|US5220294 *||May 21, 1991||Jun 15, 1993||Nec Corporation||Phase-locked loop circuit|
|US5223755 *||Dec 26, 1990||Jun 29, 1993||Xerox Corporation||Extended frequency range variable delay locked loop for clock synchronization|
|US5248946 *||Dec 11, 1991||Sep 28, 1993||Mitsubishi Denki Kabushiki Kaisha||Symmetrical differential amplifier circuit|
|US5253042 *||Dec 12, 1991||Oct 12, 1993||Sony Corporation||Burst phase detection circuit|
|US5253187 *||Feb 19, 1992||Oct 12, 1993||Canon Kabushiki Kaisha||Coordinate input apparatus|
|US5309162 *||Dec 10, 1992||May 3, 1994||Nippon Steel Corporation||Automatic tracking receiving antenna apparatus for broadcast by satellite|
|US5362995 *||May 3, 1993||Nov 8, 1994||Nec Corporation||Voltage comparing circuit|
|US5394024 *||Dec 17, 1992||Feb 28, 1995||Vlsi Technology, Inc.||Circuit for eliminating off-chip to on-chip clock skew|
|US5422918 *||Dec 9, 1993||Jun 6, 1995||Unisys Corporation||Clock phase detecting system for detecting the phase difference between two clock phases regardless of which of the two clock phases leads the other|
|US5432480 *||Apr 8, 1993||Jul 11, 1995||Northern Telecom Limited||Phase alignment methods and apparatus|
|US5440274 *||Nov 24, 1992||Aug 8, 1995||Texas Instruments Deutschland Gmbh||Phase detector circuit and PLL circuit equipped therewith|
|EP0490690A1 *||Dec 12, 1991||Jun 17, 1992||Mitsubishi Denki Kabushiki Kaisha||Amplifier circuit|
|EP0521215A1 *||Oct 2, 1991||Jan 7, 1993||Exar Corporation||Disk drive read circuit with programmable differentiator delay|
|1||*||Patent Abstracts of Japan, vol. 4, No. 67 (E 11) (549) May 20, 1980 & Japan Application, 55 035 516 (Tokyo Shibaura Denki K.K.) Mar. 12, 1980.|
|2||Patent Abstracts of Japan, vol. 4, No. 67 (E-11) (549) May 20, 1980 & Japan Application, 55 035 516 (Tokyo Shibaura Denki K.K.) Mar. 12, 1980.|
|Citing Patent||Filing date||Publication date||Applicant||Title|
|US6720832 *||Jun 26, 2002||Apr 13, 2004||Infineon Technologies North America Corp.||System and method for converting from single-ended to differential signals|
|US6784735 *||Apr 16, 2003||Aug 31, 2004||Frontend Analog And Digital Technology Corporation||High switching speed differential amplifier|
|US7312643 *||Mar 1, 2004||Dec 25, 2007||Oki Electric Industry Co., Ltd.||Differential current driver and data transmission method|
|US7486117||Nov 13, 2007||Feb 3, 2009||Oki Semiconductor Co., Ltd.||Differential current driver and data transmission method|
|US7521994||Sep 20, 2007||Apr 21, 2009||Micronas Gmbh||Three-stage amplifier|
|US8040168 *||Jul 26, 2005||Oct 18, 2011||Panasonic Corporation||Charge pump circuit|
|US20030227327 *||Apr 16, 2003||Dec 11, 2003||Wei-Cheng Lin||High switching speed differential amplifier|
|US20040189354 *||Mar 1, 2004||Sep 30, 2004||Oki Electric Industry Co., Ltd.||Differential current driver and data transmission method|
|US20060097772 *||Jul 26, 2005||May 11, 2006||Matsushita Electric Industrial Co., Ltd.||Charge pump circuit|
|US20060097778 *||Oct 26, 2005||May 11, 2006||Kwon Hyuk-Joon||Differential amplifier with high voltage gain and stable common mode output voltage|
|US20080079466 *||Nov 13, 2007||Apr 3, 2008||Oki Electric Industry Co., Ltd.||Differential current driver and data transmission method|
|US20080122536 *||Sep 20, 2007||May 29, 2008||Micronas Gmbh||Three-stage amplifier|
|EP1903672A2 *||Aug 30, 2007||Mar 26, 2008||Micronas GmbH||Three-stage amplifier|
|EP1903672A3 *||Aug 30, 2007||Apr 9, 2008||Micronas GmbH||Three-stage amplifier|
|U.S. Classification||327/536, 330/257, 327/563, 330/253|
|International Classification||H03L7/093, H03F3/45|
|Cooperative Classification||H03F2203/45644, H03F2203/45176, H03F2203/45201, H03F3/45183, H03F2203/45466, H03F2203/45208, H03F2203/45674, H03F3/45076, H03F2203/45188, H03F3/45179, H03F2203/45712|
|European Classification||H03F3/45S1B, H03F3/45S1B1, H03F3/45S1|
|Dec 5, 2002||FPAY||Fee payment|
Year of fee payment: 8
|Dec 6, 2006||FPAY||Fee payment|
Year of fee payment: 12