|Publication number||USRE40352 E1|
|Application number||US 10/846,998|
|Publication date||Jun 3, 2008|
|Filing date||May 14, 2004|
|Priority date||Sep 11, 1998|
|Also published as||DE19841754A1, EP1119876A2, US6388287, US20010050549, WO2000016407A2, WO2000016407A3|
|Publication number||10846998, 846998, US RE40352 E1, US RE40352E1, US-E1-RE40352, USRE40352 E1, USRE40352E1|
|Inventors||Gerald Deboy, Franz Hirler, Martin März, Hans Weber|
|Original Assignee||Infineon Technologies Ag|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (13), Non-Patent Citations (2), Referenced by (1), Classifications (21), Legal Events (3)|
|External Links: USPTO, USPTO Assignment, Espacenet|
This application is a continuation of copending International Application No. PCT/DE99/02874, filed Sep. 10, 1999, which designated the United States.
The present invention relates to a switch mode power supply having a switching transistor with reduced switching losses.
The term “switch mode power supply” is intended to cover all types of forward converters, flyback converters, half-bridge and full-bridge converters as well as step-up and step-down controllers, such as those used in power supply units, lamp ballasts, welding converters or RF converters.
For some time now, there has been an increasing trend toward system miniaturization and toward increasing the power density in components. It can be excepted that this tendency in power electronics will also be continued in the future. This trend is associated with a development toward even higher switching frequencies, since this is the only way in which passive components can also be miniaturized in a corresponding manner. Furthermore, particularly in the case of industrial generators, power switching transistors in switch mode power supplies are starting to run into frequency bands that have been reserved for electron tubes for a long time. This is to say, for example, frequency bands up to what is referred to as the “ISM” frequency of 13.56 MHz.
As the switching frequency rises, the switching losses in switching transistors become increasingly important. Roughly speaking, these switching losses can be split into three groups:
The switching losses mentioned in item (a) above and which can scarcely be influenced by the switching transistor itself are reduced or are entirely avoided at the moment at high switching frequencies by using special components, such as Schottky, diodes, or by selecting circuit topologies without any critical communication processes, such as resonant converters. Losses such as these also include, for example, the power loss caused by the recovery charge when the current is actively switched off by pn-diodes.
The other switching losses, as mentioned in items (b) and (c) above, can be influenced significantly by the characteristics of the switching transistor and its actuation. For example, the current-voltage overlap losses are dependent to a major extent on the duration of the switching process itself.
For illustrative purposes,
Because of the high gate charge of present-day power switches, in particular MOSFETs, high switching speeds demand very high driver currents, so that reducing the switching times T frequently runs into limits just for cost reasons.
A capacitor 4 with an external capacitance Cext is therefore provided in parallel with the power switch 1 to relieve the switching-off load. Consideration has also already been given to use the MOSFET's own output capacitance for reducing the switching load (see B. Carsten: “FET selection and driving considerations for zero switching loss and low EMI in HF “Thyristor dual” power converters”, Power Conversion 1996, Conference Proceedings 5/96, pages 91-102).
As can be seen from FIG. 2(b), the capacitor 4 with the capacitance Cext slows down the rise in the voltage Uds. At the same time, two current paths are produced, with a current Ich via the channel of the MOSFET which forms the power switch, and a current via the capacitor 4. Since the current Ich can now be switched off very quickly—without causing any rise in the drain-source voltage Uds (no “Miller” effect, see the gate-drain capacitance Cgd in FIG. 1), the overlapping area of the current Ich and the drain-source voltage Uds, and thus the switching-off energy loss (see the area 6) can be reduced virtually indefinitely.
However, circuitry such as this is suitable only for circuits in which the power switch is switched on at zero voltage (zero voltage switching ZVS), since, otherwise, the losses are just moved from the switching-off to the switching-on process. Specifically, when switching on at a time when voltage is present, the energy stored in the external capacitor 4 and the energy stored in the output capacitance are converted into heat losses in the power switch 1 [see the shaded area 5 in
No satisfactory solution has yet been found to avoid the above switching losses.
It is accordingly an object of the invention to provide a switch mode power supply having a switching transistor which overcomes the above-mentioned disadvantageous of the prior art apparatus of this general type, and which, in particular, is distinguished by drastically reduced switching losses.
With the foregoing and other objects in view there is provided, in accordance with the invention, A switch mode power supply that includes a switching transistor having a load path formed by a first main connection and a second main connection. The first main connection and the second main connection are provided for receiving a voltage applied thereto. The switching transistor includes a semiconductor body with a semiconductor layer of a first conductance type forming a drift area. A load is connected in series with the load path of the switching transistor. A continuous drain region of a second conductance type is incorporated into the drift area and is connected to the first main connection. A continuous source region of the second conductance type is incorporated into the drift area and is connected to the second main connection. A reverse-biased pn-junction is produced by an interaction between the semiconductor body and the continuous drain region and between the semiconductor body and the continuous source region. The reverse-biased pn-junction has a large inner voltage-dependent surface area that is variable as a function of the voltage applied to the first main connection and the second main connection. When the voltage applied is 10 V, the switching transistor is characterized by a first product of a switch-on resistance Ron and a gate charge Qgtot, the first product given by: Ron*Qgtot/10 V≦2.5 ns. When the voltage applied is 400 V, the switching transistor is characterized by a second product of the switch-on resistance Ron and the energy Eds stored in a drain-source capacitance, the second product given by Ron*Eds≦1.6 V2 μs.
In other words, the switch mode power supply has a switching transistor with a first and a second main connection forming a load path. A load is connected in series with the load path of the switching transistor. The switching transistor has a semiconductor body with a semiconductor layer of the first conductance type into which continuous regions of the second conductance type are incorporated which, interacting with the semiconductor body at the reverse-biased pn-junction which is formed, create a large inner, voltage-dependent source area. The surface area is varied as a function of a voltage applied to the main connections.
In accordance with an added feature of the invention, the switching transistor is in the form of a MOSFET. The output capacitance values for low drain-source voltages (for example less than 45 V for high-voltage MOSFETs) are very high, with this capacitance decreasing, as the drain-source voltage rises, to values which are so low that the energy stored in the transistor assumes very low values.
In accordance with an additional feature of the invention, the voltage-dependent surface area of the pn-junction is reduced as the voltage applied is increased.
In accordance with another feature of the invention, an amount of charge in the switching transistor, which is calculated via a line integral along a line at right angles to the pn-junction, remains below a material-specific breakdown charge.
In accordance with a further feature of the invention, the continuous drain region of the second conductance type and the continuous source region of the second conductance type are configured in a structure selected from the group consisting of a vertical structure and a lateral structure.
In accordance with a concomitant feature of the invention, a charge storage device is connected in parallel with the load path of the switching transistor.
Other features which are considered as characteristic for the invention are set forth in the appended claims.
Although the invention is illustrated and described herein as embodied in a switch mode power supply with reduced switching losses, it is nevertheless not intended to be limited to the details shown, since various modifications and structural changes may be made therein without departing from the spirit of the invention and within the scope and range of equivalents of the claims.
The construction and method of operation of the invention, however, together with additional objects and advantages thereof will be best understood from the following description of specific embodiments when read in connection with the accompanying drawings.
Referring now to the figures of the drawing in detail and first, particularly, to
In order to reduce the switching losses in comparison to the prior art, the MOSFET of the power switch 1 specifically has the following characteristics:
None of the currently available power switches can satisfy the characteristics (b) and (c) at the same time. However, this can actually be achieve by the present invention since, as is shown in
In the switching transistor according to the invention, the drain-source capacitance Cds falls very rapidly to very low values as the drain-source value Uds rises, so that the energy stored in the drain-source capacitance Cds:
Eds(Umax)=∫o UmaxCds(U)·U·dU (equation 1)
for the envisaged operation voltage remains less than with conventional MOSFETs. In this way, not only are the switching-off losses reduced, but also the switching-on losses when switching with voltage applied. In the case of tuned circuits, the small amount of energy in the drain-source capacitance Cds of the power switch in the switching transistor according to the invention advantageously reduces the communication energy required for zero voltage switching.
In comparison with present-day MOSFETs, a transistor with optimized power loses has particularly low products of the switch-on resistance Rdx(on) and the gate charge Qgtot, and also of the switch-on resistance and stored energy Eds, and in consequence, achieves extraordinarily low switching losses. For example, for 600 V MOSFETs, the product Ron*Eds (400 V) is preferably ≦1.6 V2 μs and the product Ron*Qgtot (10 V)/10 V is preferably ≦2.5 ns. The bracketed expressions (400 V) etc. indicate the applied voltages.
The extremely non-linear capacitance profile of the power switch 1 furthermore leads to a soft switching edge, which has few harmonics an is advantageous from the EMC point of view. As illustrated in
The stated non-linear capacitance profile can be noticed particularly advantageously in bridge circuits since in this case—because of the very high capacitance values for low drain-source voltages—both the lower and the upper “corner” of the voltage edge are rounded.
To a first approximation, the output capacitance Coss corresponds to the capacitance Cds and Cgd connected in parallel. In this case, Cgd should remain as low as possible in order to minimize the reaction (“Miller” effect).
The desired high value of the drain-source capacitance Cds can be achieved in a power switch by deliberately increasing the surface area of the reverse-biased pn-junction. Technically speaking, structures such as this can be produced by inserting, for example, the p-conductive regions in the n-conductive drift path of the semiconductor power switch. The reverse-biased pn-junction must in this case have a continuous inner surface area, that is to say all of the p-regions must be conducively connected to one another. The majority of the output capacitance in such a configuration results from the drain-source capacitance. In order to keep the amount of energy stored in the power switch as low as possible, the output capacitance must assume very low values at high voltages, as shown in Equation 1.
In MOSFETs, there is admittedly a capacitance reduction, which is inversely proportional to the square root of the drain-source voltage Uds and reflects the increases (when considered in one dimension) in the width of the space-charge zone.
However, it is necessary to reduce the surface area of the reverse-biased pn-junction as the voltage rises if a considerably more rapid reduction in the drain-source capacitance Cds is to be achieved. If the doping and dimensions of the p-regions incorporated in the drift zone are dimensioned such that their charge is depleted by the opposite charge of the surrounding semiconductor material by transverse electrical fields at low voltages, then this results in a rapid reduction in the surface area as the voltage rises. The dimensioning must in this case be such that the line integral of the doping of the p-regions at right angles to its surface remains below the material-specific breakdown charge. The depletion voltage in this case becomes less, the shorter the distances between the semiconductor regions of opposite conductance type.
The above condition does not demand complete compensation for the basic material, and in fact, the principle also operates with incomplete compensation. In the event of overcompensation, on the other hand, an additional vertical electrical field is required in order to deplete the p-regions. The steep drop in the drawin-source capacitance is thus moved to higher voltages.
The abovementioned principle can likewise be used in the field of low blocking-capability transactors.
A silicon semiconductor layer 10, which is, for example, between 4 and 50 μm thick is located on an n+-doped silicon semiconductor substrate 9. A p−-conductive source zone 11 and a p−-conductive drain zone 12, which each have a penetration depth of about 2 μm, are embedded in the silicon semiconductor layer 10. Underneath the source zone 11 and drain zone 12, a p-conductive region 13 or 14, respectively, with a penetration depth of 4 to 40 mm is formed, so that it ends at a distance of about 1 to 10 mm from the semiconductor substrate 9. The higher values above apply to high-voltage components, while the lower values are appropriate for low-voltage components.
Another gate electrode 15 is provided between the source and drain zones 11 and 12, respectively.
The indicated conductance types can, of course, also in each case be reversed. It is also possible to use lateral structures instead of vertical structures.
The configuration of n-conductive continuous regions 14 in a p-conductive semiconductor layer 10 is shown in a plan view in
In the exemplary embodiments, the regions 13, 14 are in the form of strips. The MOSFET is shown laterally in the illustration. The reference numeral 12 denotes the drain zone, and the reference number 11 denotes the source zone. The gate electrode 15 is indicated by dashed lines. The n-conductive regions 14 are incorporated in the p-conductive semiconductor layer 10. The profile of the space-charge zone for various drain-source voltages is also shown in
The invention makes it possible to produce a switch mode power supply with reduced switching losses using a MOSFET power switch 1. The output capacitance Cds of the MOSFET power switch 1 falls quickly to low values as a function of the drain-source voltage Uds, such that a load current Id applied to the MOSFET power switch 1 changes from initially being the channel current of the MOSFET power stitch 1 to being the charge current of the output capacitance Coss.
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|International Classification||H01L29/10, H02M3/00, H03K17/695, H01L29/78, H01L27/04, H01L21/8234, H03K17/0814, H01L27/06, H01L21/822, H01L29/76|
|Cooperative Classification||H01L29/7816, H01L29/7802, H03K17/08142, H01L29/1095, H01L29/0634|
|European Classification||H01L29/78B, H01L29/10G, H03K17/0814B, H01L29/78B2, H01L29/06B2B3R2|
|Aug 21, 2007||AS||Assignment|
Owner name: INFINEON TECHNOLOGIES AG, GERMANY
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Effective date: 20070820
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