|Publication number||USRE42799 E1|
|Application number||US 12/147,975|
|Publication date||Oct 4, 2011|
|Filing date||Jun 27, 2008|
|Priority date||Oct 2, 2000|
|Also published as||CN1504015A, DE10197148T0, DE10197148T1, US7068987, US20020042256, WO2002054606A2, WO2002054606A3|
|Publication number||12147975, 147975, US RE42799 E1, US RE42799E1, US-E1-RE42799, USRE42799 E1, USRE42799E1|
|Inventors||Keith R. Baldwin, Mark A. Webster|
|Original Assignee||Intellectual Ventures I Llc|
|Export Citation||BiBTeX, EndNote, RefMan|
|Patent Citations (85), Non-Patent Citations (17), Classifications (21), Legal Events (4)|
|External Links: USPTO, USPTO Assignment, Espacenet|
The present application is based on U.S. Provisional Patent Application entitled “Packet Acquisition and Channel Tracking For A Wireless Communication Device Configured In A Zero Intermediate Frequency Architecture”, Ser. No. 60/259,731, filed Jan. 4, 2001, which is hereby incorporated by reference in its entirety. The present application is also a continuation-in-part (CIP) of U.S. Patent Application entitled “A Calibrated DC Compensation System For A Wireless Communication Device Configured In A Zero Intermediate Frequency Architecture”, Ser. No. 09/677,975, filed Oct. 2, 2000, which is hereby incorporated by reference in its entirety.
The present invention relates to wireless communications, and more particularly to packet acquisition and channel tracking for a wireless communication device configured in a zero intermediate frequency (ZIF) architecture that utilizes a DC control loop to enable direct conversion of radio frequency signals to baseband frequency and a calibration procedure that more accurately determines and controls DC voltage levels.
Network communication is a growing area of technology both for business and home applications. A network system enhances communication and provides a suitable environment for enhanced productivity and capabilities both at home and in the workplace. The Internet for example, is a global, mostly wired, communication network that couples devices together on a world-wide basis that enables world-wide communication between any devices coupled to the Internet. The Internet enables access to a plurality of services, such as file sharing, faxing, chat, email and information access via websites to files, libraries, databases, computer programs, etc.
Many businesses and commercial entities include a relatively established and sophisticated network environment for enhanced productivity and communication. For example, Extranets or Intranets provide enhanced yet protected or secure communication to a selected group of people on the Internet. Many small businesses and homes are coupled to the Internet via some variation of local area network (LAN) or the like. It is becoming more advantageous and common for small businesses and home environments to include LAN capabilities to connect to the Internet or to access other services, such as file sharing, printing, faxing, etc. and to further enable communication such as via chat and email services, and the like and to provide access to common databases and libraries, etc. Many such small networks are connected through a set of wires. For example, a network may be established in a small office or home through standard phone wires. Phone wires are already available in each office of a business and in several rooms of a typical home. Technology also exists to establish network communications via power lines which are typically available in every room of a house. Many small offices and homes may alternatively be wired with network wires, such as a twisted-pair telephone wires with corresponding RJ-45 connectors utilized by various Ethernet embodiments.
Wired networks provide a certain level of convenience but have many limitations. Each device coupled to the network must be attached to a corresponding wire through which the network is established. The location of each device, therefore, is limited to enable access to the network wires. Cable management is also a significant issue, since devices must be placed to enable proper routing of wires. It is desired that the wires be conveniently placed and for aesthetic reasons, out of sight. Wires should be located in such a manner as to reduce or eliminate any chance of accidental interference or disconnect or hazards such as tripping. Once wired devices are properly placed, movement of the devices is very limited or otherwise not practical without substantial re-configuration or re-routing of the wires. Maintenance of wired network devices can be inconvenient and often requires that the wires be removed during service and then reconnected properly.
Certain wireless technologies are known, such as infrared technology. Infrared technology works well for certain applications, such as remote control systems or the like. For network applications, infrared technology is a relatively inexpensive option but has certain limitations, including limited bandwidth, range limitations, and line-of-sight issues. Infrared technology has been utilized in certain applications, such as access points (APs) and point to point relay nodes to extend a network down hallways and the like. For example, infrared devices are known for use in hospitals, hotels and other relatively large structures. The APs or nodes, however, are usually fixed and located in such a manner, such as on the ceiling, to avoid potential interference with physical objects. Due to line of sight issues, infrared technology is not particularly convenient for network communications at the end points of the network where human interaction is necessary.
Radio frequency (RF) technology appears to be the technology of choice for establishing a viable wireless local area network (WLAN). RF technology for LAN systems, however, is not particularly optimized for small office or home use. Wireless technology is established for industrial and commercial uses and applications such as courier services, vehicle rentals, warehouse operations and inventories, etc. The wireless embodiments for commercial and industrial applications are too expensive or otherwise specialized and thus are not suited for direct use in the small office or home environment.
The Bluetooth technology is being developed for application in the home or office. Bluetooth technology offers relatively limited bandwidth at very low cost to enable connectivity and network communications between certain communication devices, such as cellular phones, computer systems including notebook, laptop and desktop computers and further including other hand-held devices such as personal digital assistants (PDAs) or the like. The Bluetooth technology, however, has limited bandwidth and therefore relatively low data throughput capability. The consumer market demands higher data throughput and reliability such as is necessary for DVD and other multimedia applications.
The typical environment for a WLAN is very noisy and not optimal for wireless communications. For example, most homes include many electronic devices resulting in an electronically noisy environment that may interfere with WLAN communications, such as microwave ovens, garage door openers, radios, television sets, computer systems, etc. Further, the communication medium between wireless devices constantly changes. For example, most environments or rooms include multiple reflective surfaces creating multipath noise in the wireless environment. Furthermore, movement of items or devices or the like such as hands, bodies, jewelry, mouse pointers, etc. or activation of electronic devices, such as cooling fans or the like, affects the overall wireless communication path and potentially degrades wireless communication performance.
Low cost and low power wireless communication devices for enabling a WLAN system or the like for use at home or in the small business is desirable. It is further desired to provide low cost and low power wireless communication devices for any type of wireless system for any type of application. The system must be relatively robust with significant performance and be capable of significant data throughput.
A method of controlling operation of a wireless communication device configured in a zero intermediate frequency (ZIF) architecture including a DC feedback control loop and a gain feedback control loop according to embodiments of the present invention includes processing energy in a wireless medium to generate a corresponding receive signal, monitoring the receive signal via a predetermined measurement window, detecting a changed condition in the wireless medium, holding the gain feedback control loop at a constant gain level after detecting the changed condition, and operating the DC feedback control loop in an attempt to search a stable DC value for the receive signal while the gain loop is held constant. There are several changed conditions contemplated herein, including onset of a new packet causing DC saturation, completion of transmission of a packet for preparation of clear channel assessment (CCA), and completion of transmission of a packet and preparation for an expected acknowledgement (ACK) packet.
The processing of energy may involve processing the noise energy in the wireless medium while no packets are being transmitted. In this manner, the wireless communication device tracks the noise floor of the channel prior to the changed condition. The changed condition may include detecting DC saturation of the measurement window, which typically indicates or is otherwise caused by operation during onset of a new packet. In one embodiment, DC saturation is detected when the sampled receive signal “rails” to one side of a digital measurement window. In normal operation, the DC feedback control loop is operated in an attempt control the DC level of the receive signal to within a predetermined maximum DC threshold level so that the gain loop may lock on the signal. However, DC saturation calls for a different procedure to quickly change the DC level of the receive signal so that the incoming signal is not obscured and is observable in the limited measurement window. Upon DC saturation, the gain feedback control loop is held and the DC feedback control loop is operated to add opposite polarity DC to the receive signal to remove the DC saturated condition. When the DC saturation condition is removed, the gain feedback control loop is released to control the power level of the receive signal to a predetermined target power level.
The method may further include detecting gain saturation in which the measurement window is clipped at both of predetermined minimum and maximum values above a predetermined clip rate and in which clipping is relatively balanced between the predetermined minimum and maximum values. In the event of gain saturation, the method may further include operating the gain feedback control loop to perform a clip-step procedure to reduce the gain of the receive signal. In one embodiment, the clip-step procedure includes using a graduated clip gain adjustment in which gain is adjusted based on an amount of clipping of the receive signal.
The method may further include processing noise energy in the wireless medium while no packets are being transmitted, operating the gain feedback control loop until a noise floor gain value is determined, and storing the noise floor gain value. For expected ACK packet acquisition, the method may further include detecting an end of transmission of a packet in the wireless medium (such as by indication of a local MAC), retrieving and applying the stored noise floor gain value in the gain feedback control loop, holding the gain at the retrieved noise floor gain value during a predetermined quiet period, and releasing the gain feedback control loop to operate in normal mode after expiration of the quiet period. In this case, it is expected that the wireless medium has not changed significantly after transmission of only one packet, so that the gain is retrieved as soon as possible after the device transmits a packet in order to facilitate re-establishing control the gain loop as soon as possible. The method may further include determining a DC noise value of the DC feedback control loop prior to transmission of the packet (while tracking the noise floor of the wireless medium), storing the DC noise value, and retrieving and applying the DC noise value to the DC feedback control loop after detecting the end of transmission of the packet. In this case, the gain of the DC loop is also restored to the same level it was prior to packet transmission in order to facilitate re-establishing control the DC loop as soon as possible after packet transmission.
In order to perform CCA, the method may further include detecting an end of transmission of a packet being transmitted in the wireless medium, retrieving and applying the stored noise floor gain value in the gain feedback control loop for a predetermined period of time, and determining whether the DC feedback control loop converges to a stable DC level within the predetermined period of time. In this case, it may be determined that the wireless medium is busy if the DC feedback control loop does not converge to a stable DC level within the predetermined period of time. This may occur if another device is transmitting a packet, such as an ACK packet in response to the original packet detected in the channel, or if some other channel phenomenon is preventing convergence of the DC loop. In either case, the wireless device backs off and does not attempt packet transmission until after CCA is determined.
A method of operating a wireless communication device (including gain and DC loops) to perform initial acquisition of a packet being transmitted in a wireless medium includes processing radio frequency (RF) energy in the wireless medium to generate a corresponding receive signal, determining if a DC threshold condition of the receive signal is exceeded, and if so, holding the gain feedback control loop at a constant gain level and operating the DC feedback control loop to reduce DC of the receive signal until the DC threshold condition of the receive signal is met, and, when the DC threshold condition of the receive signal is met, operating the DC feedback control loop to control the DC level of the receive signal and operating the gain feedback control loop to control a power level of the receive signal to a predetermined target power level.
This latter method is more particular to onset of a new packet in the wireless medium. In one embodiment in which the receive signal is an analog signal, the method may include sampling the receive signal with an analog to digital converter (ADC) to generate corresponding digital samples, where the ADC has a sufficient range to maintain signal integrity without covering a total potential signal range of the receive signal. In this case, DC saturation is determined when a DC threshold condition is exceeded. The digital samples may range between a minimum value and a maximum value, where the DC threshold condition is exceeded when a percentage of digital samples at the minimum value or a percentage of digital samples at the maximum value exceeds a predetermined threshold percentage of a total number of digital samples. For example, this may occur when the number of clipped signals at one rail exceeds a predetermined clipping rate. The predetermined clipping rate is arbitrarily determined depending upon the particular configuration, but is typically a relatively high percentage, such as 90% or the like.
When the DC threshold condition is exceeded, the method includes operating the DC feedback control loop by adding DC to the receive signal. Several methods are contemplated, including conducting a step search procedure by repeatedly adding a predetermined DC amount until the DC threshold condition is met. The predetermined DC amount may be any arbitrary value, but should be selected to avoid overshooting the gain window and thus may be related to the relative size of the measurement window. In an alternative embodiment, the method includes performing a successive approximation DC search procedure until the DC threshold condition is met. In this case, a binary search or the like is performed to converge on the applicable DC level. The step search procedure may take longer than the successive approximation technique in many cases, but is sufficient, simple, and relatively easy to implement.
The method may further include detecting a gain saturation state in which clipped digital samples occur at both of the minimum and maximum levels at a rate greater than a predetermined clip ratio threshold and in which a number of digital samples at the minimum level is relatively balanced with a number of digital samples at the maximum level. The balance may be as rough or accurate as desired, such as 3:1, 2:1 or as close to 1:1 as desired. When the gain saturation state is detected, the method includes operating the gain feedback control loop in a clipping mode while in the gain saturation state. The clipping mode may involve adjusting the gain level of the gain feedback control loop based on an amount of clipping using a graduated clip gain adjustment. The graduated clip gain adjustment may include a corresponding one of a predetermined plurality of gain level adjustments for each of a plurality of predetermined clip ratio ranges. The graduated clip gain adjustment may be graduated between a high gain adjustment for a high clip ratio and a low gain adjustment for a low clip ratio.
A method of operating a wireless transceiver device in preparation for an expected ACK packet includes holding a gain level of the gain feedback control loop constant after a packet is transmitted and during a predetermined quiet period, operating the DC feedback control loop during the predetermined quiet period in an attempt to resolve DC level while the gain feedback control loop is held, and releasing the gain feedback control loop to operate in a normal packet acquisition mode after the quiet period. In this case, the method may include storing a gain level value of the gain feedback loop before the packet was transmitted and retrieving the stored gain level value and holding the gain feedback control loop at the retrieved gain level value during the predetermined quiet period after transmission of the packet. Also, the method may include storing a DC value of the DC feedback control loop before transmission of the packet and applying the stored DC value after packet transmission. In this method, the wireless transceiver device may be the device that transmits the packet and expects to receive the ACK packet, or the wireless transceiver device may be a different device operating in ACK priority mode, such as network monitor or analyzer operating in a “sniffer” mode or the like to analyze protocol.
A method of operating a wireless communication device to determine CCA of a wireless medium includes storing a gain level value of the gain feedback control loop, detecting a packet being transmitted, retrieving the stored gain level value and holding the gain feedback control loop at the retrieved gain level value after transmission of the packet is completed, operating the DC feedback control loop to search a stable DC level, and determining whether the DC loop converges to a stable DC level. Again, the gain level value may be determined and stored while tracking the wireless medium with no packets being transmitted. Also, the method may include storing a DC value of the DC feedback control loop before detecting a packet being transmitted, and applying the stored DC gain setting after packet transmission. Further, the method may include holding the gain level of the gain feedback control loop constant during a predetermined quiet period, and releasing the gain feedback control loop after the predetermined quiet period to operate in a normal mode. In this case, the method may include determining if the DC feedback control loop converges to a stable DC level within a predetermined period, and if so, operating the DC feedback control loop and the gain feedback control loop in a normal mode. The wireless medium is determined to be busy if the DC feedback control loop does not converge to a stable DC level within a predetermined period of time.
A better understanding of the present invention can be obtained when the following detailed description of one or more embodiment(s) of the invention is considered in conjunction with the following drawings, in which:
Within the area 101, wireless devices 111 and 113 are able to communicate with each other via the AP 121, and within the area 103, wireless devices 115 and 117 are able to communicate with each other via the AP 123. The AP 121 enables the devices 111 and 113 to communicate to the server 127 via the LAN 125 and the AP 123 also enables the devices 115 and 117 to communicate to the server 127 via the LAN 125. It is further noted that the LAN 125 enables the devices 111, 113, 115 and 117 to communicate with each other and with any other device coupled thereto.
APs, if used, are usually connected to a wired LAN, such as the LAN 125, although they need not be. The number and location of APs usually depends on the particular configuration and needs, such as the number of users served, range of coverage, and/or adjacent channel interference. It is noted that a single AP can serve a whole floor of rooms subject to user density and interference. Multiple APs are used much like cells in cellular communication. Adjacent APs may operate on adjacent channels to prevent interference. Roaming capability may be enabled to allow devices to move from one area to another, such as from area 101 to 103 and vice versa. Generally, each AP has additional software and has access to house or building main power.
An alternative embodiment is shown by area 105 including devices 131, 133 and 135. The devices 131, 133 and 135 communicate directly with one another rather than via any APs. Networks without APs are ad hoc in nature and typically contain fewer clients. The primary difference between the devices 111, 113, 115 and 117 from the devices 131, 133 and 135 is the mode of operation. Each of the wireless devices 111, 113, 115, 117 and 131, 133, 135 includes a wireless transceiver that may be implemented according to an embodiment of the present invention for enabling wireless communications.
The devices 111, 113, 115, 117, 131, 133, and 135 maybe anytype of device that includes wireless communication capabilities. In an office or home environment for example, the devices may each comprise any one or more personal computers, laptop computers, desktop computers, etc., printing devices including any type of printer technology, personal digital assistants (PDAs) or the like, scanners, fax machines, etc. The use of wireless technology enables any of the devices to be located anywhere within a given area to communicate with, and transfer data and information between, other devices within the same communication area. For example, the device 111 may be a computer that wirelessly communicates to the device 113 which may be a printer. Further, the device 111, as a computer, may send and receive files to the server 127. The device 111 may be moved to the area 103 from area 101 and still maintain communication with the LAN 125 and the server 127 via the AP 121. The LAN 125 may comprise any type of wire technology, such as telephone wires, power wires, twisted-pair, coaxial cables, etc., and may be implemented according to any type of suitable architecture, such as any type of Ethernet configuration or the like. It is further noted that a wireless LAN 125 is contemplated, where the APs 121, 123 is include corresponding wireless transceivers and are mobile or portable devices. Within the area 105, the devices 131-135 themselves establish a wireless LAN (WLAN) for communicating to each other in a similar fashion.
A problematic characteristic of traditional ZIF architectures is the introduction of substantial DC offset voltages in the receiver that must be compensated for or otherwise eliminated to capture incoming signals and allow communications. The wireless transceiver 200 is configured with an automatic gain control (AGC) loop 345 (
The wireless transceiver 200 may utilize any desired carrier frequency and modulation technique to achieve any of several corresponding data throughputs. For example, the wireless transceiver 200 may be configured to operate according to the Institute of Electrical and Electronics Engineers (IEEE) 802.11b with a carrier frequency of approximately 2.4 gigahertz (GHz) and with data throughputs of 1, 2, 5.5 or 11 Megabits per second (Mbps). Alternatively, the wireless transceiver 200 may be configured according to IEEE 802.11a with a carrier frequency of approximately 5 GHz for data throughputs of 6, 12, 18, 24, 36 or 54 Mbps. In the embodiment shown, the wireless transceiver 200 operates in accordance with IEEE 802.11b at a carrier frequency of 2.4 GHz with data throughput rates of 1, 2, 5.5 or 11 Mbps. The direct sequence spread spectrum (DSSS) modulation technique is used in the embodiment shown, although it is understood that the present invention is not limited to any particular communication or modulation technique or standard.
Any of the devices 111-117 or 131-133 or the APs 121, 123 may be configured according to the wireless transceiver 200. The particular configuration depends upon the type of device and communication interface. The wireless transceiver 200 may be configured as a plug-in peripheral or expansion card that plugs into an appropriate slot or interface of a computer system. For example, for portables and laptop computers and the like, the wireless transceiver 200 may be implemented as a Personal Computer Memory Card International Association (PCMCIA) card or PC Card that plugs into a PC Card slot of the computer. The antenna may be incorporated on the PC Card itself, or provided externally or on the computer and interfaced to the PC Card in any appropriate manner. For desktop computers and the like, the wireless transceiver 200 may be implemented according to any type of expansion or peripheral standard, such as according to the peripheral component interconnect (PCI), the Industry Standard Architecture (ISA), the Extended-ISA (EISA) standard, etc. For example, the wireless transceiver 200 may be implemented on a PCI card that plugs into a PCI slot. Again, the antenna may be integrated or coupled externally. Mini PCI cards with antennas embedded in displays are also contemplated. Self-contained or standalone packaging with appropriate communication interface(s) is also contemplated, which is particularly advantageous for APs. For example, the wireless transceiver 200 may be implemented as a separate unit with serial or parallel connections, such as a Universal Serial Bus (USB) connection or an Ethernet interface (twisted-pair, coaxial cable, etc.), or any other suitable interface to the device.
Digital data sent from or received by the wireless transceiver 200 is processed through a medium access control (MAC) device 205. For transmission, the MAC device 205 asserts digital data signals to a packet encoder 207, which formulates the data into packets for transmission. In packet encoder 207, a packet is formed by the concatenation of three serial bit streams. The first serial bit stream, referred to as the preamble, is a 1 Mbps stream of ones or zeroes followed by a synchronization (sync) pattern. The second serial bit stream, referred to as the header, is a 1 or 2 Mbps stream of packet information, such as packet length and the data rate of the data portion of the packet. The third serial bit stream, referred to as the data portion or payload, is any selected one of 1, 2, 5.5, or 11 Mbps streams of data.
The packet encoder 207 provides baseband encoded packets to a spreading encoder (SPREAD) 209. In the embodiment shown, the 1 and 2 Mbps bit rates are encoded according to Barker word encoding spreading, whereas the 5.5 and 11 Mbps bit rates are encoded according to Complementary Code Keying (CCK) encoding. The spreading encoder 209 uses a quadrature generation technique and provides in phase (I) and quadrature (Q) signals on respective I and Q channels. For 1 Mbps, every bit in 11 chips of a Barker word are output identically on the I and Q channels. If the input bit is zero, then the sense of the 11 chips is inverted, and if the input bit is one, the sense of the 11 chips is not inverted. The data is thus spread by the Barker code. For 2 Mbps, for every two bits input, 11 chips are output on each of the I and Q channels. One bit is used to invert or not the I channel and the other bit is used for the Q channel. For 5.5 Mbps, 4 bits are mapped to 8 complex (I and Q channel) chips. 8 chip complex code words are chosen and then rotated according to specific rules defined in the standard. 11 Mbps is similar to 5.5 Mbps, except that 8 bits are mapped to 8 complex chips.
The I channel signal is provided to an I digital-to-analog converter (I DAC) 211 and the Q channel signal is provided to a Q DAC 213, where the I DAC 211 and the Q DAC 213 also receive a clock signal CREF from a clock source 257. In one embodiment, the CREF signal is 44 MHz based on the 22 MHz fundamental for IEEE 802.11b. The I DAC 211 and the Q DAC 213 up-sample and digitally filter the I and Q channel signals, respectively. For 11 Mbps, the I and Q DACs convert each chip to four samples. The I DAC 211 asserts an I channel analog output signal to an anti-aliasing low-pass filter (AA LPF) 215, which provides an I channel transmit signal TX1 to an LPF 219 within the ZIF transceiver 201. The Q DAC 213 asserts a Q channel analog output signal to another AA LPF 217, which provides a Q channel transmit signal TXQ to the input of another LPF 221 within the ZIF transceiver 201.
An external voltage controlled oscillator (VCO) 229 provides an output local oscillator (LO) signal at approximately 4.8 GHz to a phase lock loop (PLL) 231 and an input of an I/Q LO generator 227. In one embodiment, a reference crystal is used to tune the VCO 229 by a voltage control to generate the 4.8 GHz LO signal. The PLL 231 receives the CREF clock signal which is divided down to an appropriate low frequency signal. The PLL 231 also divides down the VCO 4.8 GHz signal to the same appropriate low frequency signal. A phase detector circuit (not shown) within the PLL 231 compares these two low frequency signals—one being the CREF signal and the other from the VCO 229, and generates a correction (C) signal which after filtering is input to the VCO 229. This phase locked loop system reduces jitter on the VCO output LO signal. The I/Q LO 227 divides the 4.8 GHz carrier signal from the PLL 231 by two and generates two separate 2.4 GHz carrier signals, including an in-phase (I) carrier signal, and a quadrature (Q) carrier signal. The I and Q carrier signals are 90 degrees out of phase with respect to each other. The I carrier signal is provided to one input of an I channel mixer 223, and the Q carrier signal is provided to one input of a Q channel mixer 225. The other input of the I channel mixer 223 receives the output of the LPF 219 and the other input of the Q channel mixer 225 receives the output of the LPF 221. The I channel mixer 223 combines the I carrier signal with the I channel transmit signal to up-convery the I channel from baseband to 2.4 GHz radio frequency (RF). The Q channel mixer 225 combines the Q carrier signal with the Q channel transmit signal to up-convert the Q channel from baseband to 2.4 GHz RF. The outputs of the F/Q channel mixers 223, 225 are combined at the input of a variable gain amplifier (VGA) 233. The VGA 233 asserts an amplified transmit signal to the input of a fixed-gain amplifier 235, which provides its output to a radio frequency power amplifier (RFPA) 237. The RFPA 237 asserts an amplified RF signal to an RF LPF 239, which provides a filtered RF output signal to one input of a transmit/receive (T/R) switch 241. For transmission, the T/R switch 241 provides the RF output of the LPF 239 to a bi-directional band pass filter (BPF) 243, which provides a filtered RF output to a diversity switch 245. The diversity switch 245 asserts the RF transmit signal on a selected one of two antennas 247, 249.
The RFPA 237 asserts a transmit detect (TX DET) signal back to a transmit analog-to-digital converter (ADC) 251 within the baseband processor 203, which provides a corresponding digital feedback signal to the MAC 205. The MAC 205 includes a power control algorithm that detects the transmitted output signal via the ADC 251 and provides a feed forward signal to the input of a transmit (TX) DAC 253, which asserts a transmit automatic gain control (TX AGC) signal to control the gain of the VGA 233. Further, the ZIF transceiver 201 includes a voltage reference source 255 to provide a voltage reference (VREF) signal for the ZIF transceiver 201 and for the baseband processor 203. The VRFF signal may be any convenient voltage, such as 1.2 volts or the like. The clock source 257 provides the CREF signal to the PLL 231 within the ZIF transceiver 201.
The T/R switch 241 is selected to receive signals from either of the antennas 247, 249 and provides a filtered signal (through BPF 243) to the input of a balanced/unbalanced (BALUN) impedance matching network 259. The BALUN 259 provides a received RF input signal RRF to the input of a variable LNA 261. The LNA 261 asserts its output to the input of a fixed-gain LNA 263. The LNA 263 asserts the amplified RF receive signal to respective inputs of an I channel mixer 265 and a Q channel mixer 267. The I/Q LO 227 asserts the I carrier signal to another input of the I channel mixer 265 and the Q carrier signal to another input of the Q channel mixer 267. The I channel mixer 265 splits the I carrier frequency from the I channel output signal in the RF receive signal and provides the I channel output signal to an LPF 269. In a similar manner, the Q channel mixer 267 splits the Q carrier frequency from the Q channel output signal in the RF receive signal and provides the Q channel output signal to an LPF 271.
The output of the LPF 269 is provided to the input of a variable baseband automatic gain control (BB AGC) amplifier 273. In a similar manner, the LPF 271 asserts its output to the input of another BB AGC amplifier 275. The BB AGC amplifier 273 asserts an amplified I channel analog receive signal (RXI) to the input of an I channel receiver AA LPF 277, which asserts its output to the input of an I channel ADC (I ADC) 281 within the baseband processor 203. The BB AGC amplifier 275 asserts an amplified Q channel analog receive signal (RXQ) to the input of a Q channel AA LPF 279, which provides its output to an input of a Q channel ADC (Q ADC) 283 within the baseband processor 203. The I ADC 281 and the Q ADC 283 assert respective I and Q channel digital receive signals RDI, RDQ to a spreading decoder (DE-SPREAD) 285, which performs the opposite process as the spreading encoder 209 to retrieve received baseband encoded packets. The spreading decoder 285 provides the received baseband packets to a packet decoder 287, which, among other functions, retrieves the packet payloads and generates a stream of received data signals. The RDI and RDQ signals are also provided to compensation logic 284, further described below. The data signals from the packet decoder 287 are provided to a receive input of the MAC device 205, which provides received signals to the corresponding device across the implemented interface. The MAC interface may be incorporated internally within a device, such as on a PC CARD or the like, or may be external with appropriate external connectors, such as according to USB or the like.
An overload detector (OD) 289 within the ZIF transceiver 201 has first and second inputs coupled to the respective outputs of the LPF 269 and LPF 271 to detect an overload in the received input signal. The overload detector 289 asserts a baseband overload (BB OVLD) signal to an OVLD ADC 291 within the baseband processor 203, which provides a corresponding digital overload signal OVLD to the compensation logic 284. In this manner, the compensation logic 284 detects an overload of the received signal and asserts a HI/LO signal to control the variable LNA 261 of the ZIF transceiver 201. In the embodiment shown, the variable LNA 261 has an approximate 33 decibel (dB) step differential between a high gain (HI) and a low gain (LO). The gain is initially set high to detect weak signals and is switched to low gain upon certain conditions if the receive signal causes an overload condition as detected by the overload detector 289.
As more fully described below, the compensation logic 284 controls the gain of the received signal to a target power level by asserting a gain adjust signal GADJ to a gain DAC referred to as the AGC DAC 297. The AGC DAC 297 asserts a corresponding analog feedback gain control signal GAIN ADJUST to control the gain of both of the BB AGC amplifiers 273, 275. A single gain control signal is provided to both of the BB AGC amplifiers 273, 275 so that the gain of the I and Q channels of the receive signal appropriately track one another. The compensation logic 284 further asserts respective digital I and Q channel DC offset signals IDCOFF, QDCOFF to an I channel offset (I OFF) DAC 293 and a Q channel offset (Q OFF) DAC 295, respectively. The I OFF DAC 293 asserts an I channel DC offset analog signal (I OFFSET) to the LPF 269 and the Q OFF DAC 295 asserts a Q channel DC offset analog signal (Q OFFSET) to the LPF 271. In this manner, the compensation logic 284 attempts to measure and reduce or otherwise eliminate DC offsets in both the I and Q channels of the receive signal within the ZIF transceiver 201.
A ZIF architecture is utilized to obtain sufficient performance and higher data throughput at lower cost and power consumption. This is particularly advantageous to the consumer market which demands multimedia and DVD applications requiring relatively good performance. The consumer market also demands a lower cost. The ZIF architecture is one way to achieve lower cost with sufficient performance by eliminating IF components and external filters. The target carrier frequencies are in the GHz range such as 2-5 GHz ranges and higher, although the present invention is not limited to any particular frequency range. The 2-5 GHz bands are relatively noisy with significant amounts of interference. The ZIF architecture is utilized to maintain a level of performance in a noisy environment.
A particular problem with the ZIF architecture is the development of DC offsets at baseband that degrade signal-to-noise (SNR) ratio, which is directly related to performance of the system. The IF stage, which may be utilized to combat a significant level of DC offset, is not provided in the ZIF architecture. A number of sources of DC offset are due to temperature change, such as die self-heating of integrated circuits. The receive mixers 265, 267 internally inject DC offset from the carrier frequencies generated by the I/Q LO 227. This LO DC offset varies with RF frequency or channel selection as well as a selected level of gain of the variable LNA 261. This change occurs very quickly with changes of channel or gain. A slow DC drift with temperature also occurs. Externally, the LO carrier frequencies from the I/Q LO 227 leaks to the antenna input port causing DC offset. This external DC offset varies with changes in the external environment, such as movement of hands, bodies, jewelry, mouse pointers, activation of electronic devices such as cooling fans or the like and re-radiation from nearby reflectors around the antenna. Movement of the wireless devices or changes in antenna direction also causes significant changes of the propagation characteristics of the channel or area. Such environmental changes induce impedance changes that dynamically change LO leakage magnitude and phase.
The receive mixers 265, 267 also exhibit DC offsets at their outputs. This DC offset is primarily a function of circuit matching, such as bipolar and Metal Oxide Semiconductor (MOS) device matching and resistor matching. This DC offset also drifts due to temperature variations and self-heating. The I and Q channel LPFs 269, 271 also exhibit offset at their outputs. This DC offset is also primarily a function of circuit matching. The I and Q channel baseband AGC amplifiers 273, 275 also exhibit DC offset at their inputs. This DC offset is also primarily a function of circuit matching. Due to the analog control circuitry associated with this function, the impact of matching variation is more severe than the DC offset experienced by the receive mixers 265, 267 and the LPFs 269, 271. This DC offset also changes non-linearly with gain setting, and drifts due to temperature variation and self-heating. The control voltage versus gain characteristic may also vary slowly with time. The main reason for drift is slower package temperature variations. A certain level of DC offset due to internal radio jamming from harmonics of digitally generated signals also occurs. As radio jamming is injected into the antenna and/or RF front end, this DC offset is a function of channel and LNA gain as well as a strong function of the particular digital circuit that is creating the spurious signal. Below a certain frequency (f) such as 1 kilohertz (kHz), 1/f noise appears as an impairment similar in effect to DC offsets.
It is noted that one possible solution of removing DC offset is through AC coupling, such as using coupling capacitors or the like. However, AC coupling results in filtering that tends to filter out low frequency content. The amount of data thrown away becomes significant above 1 kHz, so that AC coupling must only filter below 1 kHz. Within this range, however, the settling time is too long, such as on the order of approximately 100 microseconds (μs), which is not practical in a bursty environment such as packet-based communications. The settling time is limited due to the bursty nature of communication. Short preamble time lines of EEE 802.11a and IEEE 802.11b standards have little room for accurate DC estimation.
All of the sources of DC offsets are referenced to the inputs of the BB AGC amplifiers 273, 275. The gain range of the BB AGC amplifiers 273, 275 must be sufficient to guarantee acceptable performance in a variety of environments. In the embodiment shown, the gain range of each of the BB AGC amplifiers 273, 275 is approximately −6 dB to 60 dB, or 0.5 V/V to 1000 V/V to obtain the desired operation range of approximately 0.7 V/V to 800 V/V. It has been determined that the DC offset range can be on the order of ±50-100 millivolts (mV). It is desired that the DC correction occur before the baseband amplifiers 273, 275 because of its significant gain range.
The received RF input signal RRF from the antennas 247, 249 through the BALUN 259 is provided to an RF mixer circuit referred to as the LNA/mixer 301, which represents the LNAs 261, 263 and the receive mixers 265, 267. The LNA/mixer 301 provides a baseband receive input signal RBB to one input of a combiner 303, which provides a DC-adjusted receive input signal RADJ to the input of a baseband BB LPF 305 representing the LPFs 269, 271. The other input of the combiner 303 receives a DC OFFSET signal. The combiner 303 operates as a combiner that combines the DC OFFSET signal with the RBB receive signal to provide the DC-adjusted RADJ signal. In one embodiment, the combiner 303 is a summing junction that subtracts the DC OFFSET signal from the RBB receive signal to provide the DC-adjusted RADJ signal. It is noted that the combiner 303 may alternatively operate as a summing junction that adds an inverted DC OFFSET signal. The invention is not limited to any particular implementation or design configuration.
The output of the BB LPF 305 is provided to the input of a baseband gain amplifier BB AGC amplifier 307, representing operation of the BB AGC amplifiers 273, 275. The output of the BB LPF 305 is also provided to the overload detector 289, which asserts the baseband overload signal BB OVLD to the OVLD ADC 291 within the baseband processor 203, which provides the OVLD signal to an overload (OV) detect block 309 of the compensation logic 284. The OV detect block 309 asserts the HI/LO signal to the LNA/mixer 301 to switch the variable LNA 261 between high and low gains. The BB AGC amplifier 307 asserts the amplified input signal R to the input of an AA LPF 311 representing the AA LPFs 277, 279. The output of the AA LPF 311 is provided to an ADC 313 representing the ADCs 281, 283. The ADC 313 provides a digital version RD of the amplified receive input signal R at its output, where the RD signal represents the I and Q digital receive signals RDI and RDQ. The RD signal from the ADC 313 is provided to the spreading decoder 285.
The RD signal is provided to a signal power estimate block 315 and a DC estimate block 319 of the compensation logic 284. The signal power estimate block 315 provides a input signal power estimate signal REST to one input of a combiner 321. The combiner 321 receives a target power (TP) signal from a target power block 323 and combines the TP signal with the REST signal to provide a receive error signal RE, which is provided to an input of another combiner 325. In one embodiment, the combiner 321 operates as a summing junction that subtracts the TP signal from the REST signal to provide a receive error signal RE to an input of the combiner 325. The combiner 325 combines the RE signal with a gain accumulation signal GACC from an accumulator 329 to provide the gain adjust signal GADJ at its output. In one embodiment, the combiner 325 also operates as a summing junction that subtracts the RE signal from the gain accumulation signal GACC from an accumulator 329 to provide the gain adjust signal GADJ at its output. The GADJ signal is provided to the respective inputs of the AGC DAC 297, the accumulator 329 and a gain converter 331. The AGC DAC 297 converts the digital GADJ signal to the analog GAIN ADJUST signal, which controls the gain of the BB AGC amplifier 307. The accumulator 329 continuously or periodically adjusts the GACC signal to track changes of the GADJ signal.
The DC estimate block 319 provides an output DC estimate digital signal ODCEST to the input of a DC amplifier 333, which is adjusted by a gain conversion signal GCON from an output of the converter 331. The DC amplifier 333 amplifies or attenuates (1/G) the ODCEST signal by an amount determined by the GCON signal from the converter 331, and provides an input DC estimate signal IDCEST to one input of a combiner 335, which receives a DC offset accumulation signal DACC from an accumulator 339 at its other input. The combiner 335 combines the IDCEST signal with the DACC signal to provide a DC offset signal DCADJ, which is provided to the input of a DC DAC 337 and to the input of the DC accumulator 339. In one embodiment, the combiner 335 operates as a summing junction that adds the IDCEST signal to the DACC signal to provide the DC offset signal DCADJ. The accumulator 339 continuously adjusts the DACC signal to the level of the DCADJ signal to maintain an accumulated DC value. The DC DAC 337 converts the digital DCADJ signal to the analog DC OFFSET signal provided to the combiner 303. The DCADJ signal represents operation of the IDCOFF and QDCOFF signals and the DC OFFSET signal represents operation of the I OFFSET and Q OFFSET signals.
It is noted that many alternatives and variations are contemplated. For example, the combiner 303 may be implemented as a simple summing junction, where the DC amplifier 333 or the DC DAC 337 performs negation or inversion to generate a negative DC OFFSET signal, which is then added to the RBB signal by the combiner 303.
Control logic 343 is coupled to the OV detect block 309, the target power block 323 and the DC estimate block 319 for controlling operations, setting parameters, etc. For example, the control block 343 may be used to enable a manufacturer or user to set a target power level used by the target power block 323 to control the TP signal.
In operation, the RRF signal from the antennas 247, 249 is converted to the RBB signal at baseband frequency by the LNA/mixer 301. The OV detect block 309 initially sets the gain of the LNA portion of the LNA/mixer 301 high by asserting the HI/LO signal to HI in order to ensure detection of a valid but weak RF signal received by either of the antennas 247, 249. The RBB signal is DC adjusted by the combiner 303, which combines or otherwise subtracts the DC OFFSET signal and provides the RADJ signal to the BB LPF 305. The BB LPF 305 provides a filtered version of the RADJ signal to the BB AGC amplifier 307. The DC OFFSET signal is based on an estimate of the amount of undesired DC at the input of the BB AGC amplifier 307. The overload detector 289 detects whether an overload condition of the received input signal at the output of the BB LPF 305 exists, and if so, asserts the BB OVLD signal. If an overload condition is detected, the OV detect block 309 asserts the HI/LO signal to LO to reduce the input gain of the LNA/mixer 301 to bring the received input signal RBB closer to a target power level.
The BB AGC amplifier 307 amplifies the RADJ signal to provide the R signal to the AA LPF 311 and then to the ADC 313 of the baseband processor 203. The ADC 313 converts the analog R signal to the digital receive signal RD, which is provided to the spreading decoder 285 (as the RDI and RDQ signals). The signal power estimate block 315, the combiners 321 and 325, the target power block 323, the accumulator 329 and the AGC DAC 297, in combination with the signal path devices including the BB AGC amplifier 307, the AA LPF 311 and the ADC 313, form the AGC loop 345, which is a gain feedback circuit centered around the BB AGC amplifier 307. The AGC loop 345 includes gain control logic that receives the amplified input signal, that estimates input signal power and that asserts the gain adjust signal in an attempt to keep the input signal power at a target power level. In particular, the signal power estimate block 315 estimates the power of the baseband input signal within the RD signal and asserts the REST signal indicative thereof. In one embodiment, this power estimate excludes any DC level of the RD signal. The combiner 321 compares the REST signal with the TP signal to generate the RE signal, which identifies an amount of power error of the RD signal relative to the target power level represented by the TP signal. The combiner 325 adjusts the present value of the GADJ signal in an attempt to compensate for any power error that exists. The AGC DAC 297 converts the GADJ signal to the analog GAIN ADJUST signal, which is provided to the gain control input of the BB AGC amplifier 307 to control the power level of baseband input signal within the RD signal. Thus, the AGC loop 345 attempts to maintain or otherwise regulate the power level of the input baseband signal within the RD signal to the target power level.
In the embodiment shown, the BB AGC amplifier 307, representing either of the BB AGC amplifiers 273, 275, uses a logarithmic gain scale, such as measured in dB, that ranges between −6 and 60 dB. The RD signal is in digital format, so the signal power estimate block 315, the target power block 323, the accumulator 329 and the combiners 321, 325 are digital devices and the REST, TP, RE, GACC and GADJ signals are digital signals. The AGC DAC 297 is a 7-bit DAC with 128 steps that outputs an analog current signal with a control current range of approximately ±1 mA to achieve the desired gain range for the BB AGC amplifier 307. The current output of the AGC DAC 297 is converted to a voltage signal, such as by a resistive network or the like (not shown), to obtain the GAIN ADJUST signal.
The DC estimate block 319, the gain converter 331, the DC amplifier 333, the combiner 335, the accumulator 339, the DC DAC 337 and the combiner 303, in combination with the receive signal path between the combiner 303 and the ADC 313, form the DC loop 347 that operates to attempt to reduce or otherwise remove DC from the RD signal in the input signal path. The DC loop 347 includes DC control logic that estimates an amount of DC in the amplified input signal and provides a DC offset in an attempt to reduce DC in the amplified input signal. In particular, the DC estimate block 319 estimates the DC offset of the RD signal, and provides the ODCEST signal indicative of the amount of DC at the output of the BB AGC amplifier 307. Since the DC OFFSET signal is applied at the combiner 303 at the input of the BB AGC amplifier 307, and since any remaining DC of the RADJ signal is effectively amplified by the BB AGC amplifier 307, the DC amplifier 333 operates to compensate for the gain of the BB AGC amplifier 307. The gain converter 331, receiving the GADJ signal and generating the GCON signal that controls the gain of the DC amplifier 333, operates as a gain interface that converts gain levels between the gain control logic and the DC control logic. In one embodiment, the gain converter 331 inverts the gain (1/G). The gain converter 331 may further compensate for different gain ranges and/or different gain scales between the BB AGC amplifier 307 and the DC amplifier 333. In this manner, the IDCEST signal, asserted by the DC amplifier 333, represents an error of the DC OFFSET signal. The combiner 335 adjusts the DCADJ signal, maintained by the accumulator 339 as the DACC signal, by the IDCEST signal to compensate for errors in the DC OFFSET signal.
In the embodiment shown, the RD signal is in digital format, so the DC estimate block 319, the DC amplifier 333, the accumulator 339 and the combiner 335 are digital devices and the ODCEST, IDCEST, DACC and DCADJ signals are digital. Also, the DC DAC 337, representing either of the I, Q OFF DACs 293, 295, is a 12-bit, current-based DAC operated in 2's-complement format that outputs an analog current with an approximate range of ±5 milliamps (mA). The DAC output current is converted to the DC OFFSET voltage signal having a range of approximately ±75 mV to compensate for an expected DC offset range of approximately ±64 mV. The gain converter 331 may operate entirely as a digital device to convert the digital GADJ signal to a digital GCON signal. Alternatively, it is contemplated that the GCON signal may be an analog signal depending upon the configuration of the DC amplifier 333. In the embodiment shown, the DC amplifier 333 is a linear gain amplifier. In general, the BB AGC amplifier 307 amplifies the input signal including any DC, so that the DC amplifier 333 operates as an amplifier that amplifies the ODCEST signal to maintain control of the DC loop 347. The amount of amplification by the DC amplifier 333 is controlled by the gain converter 331 via the GCON signal. The gain converter 331 receives the GADJ signal and adjusts the GCON signal accordingly.
As described previously, the gain converter 331 operates as a gain interface that converts gain levels between the gain control logic and the DC control logic. In one embodiment, the gain converter 331 inverts the gain between the BB AGC amplifier 307 and the DC amplifier 333. The gain converter 331 may further convert between the corresponding ranges or units along with any bit weighting of the GADJ and GCON signals so that the amplification of the ODCEST signal corresponds to the gain of the RADJ signal. In the alternative or in addition, the gain converter 331 further converts between gain scales, such as between logarithmic and linear scales. In particular for the embodiment shown, since the BB AGC amplifier 307 is a logarithmic gain amplifier measured in decibels (dBs) and since the DC amplifier 333 is a linear gain amplifier, the gain converter 331 converts from the GADJ signal in dBs to the linear gain GCON signal. For embodiments in which the BB AGC amplifier 307 and the DC amplifier 333 are both linear amplifiers, the gain converter 331 converts between the corresponding ranges or units along with any bit weighting of the GADJ and GCON signals without need of logarithmic conversion.
As previously described, there are many sources of DC, and the DC level changes over time. The BB AGC amplifier 307 itself introduces DC offset that may change significantly from one gain step to another. In the embodiment shown, the calibration block 401 is a digital device that operates as a calibrator, which monitors the digital RD signal, makes appropriate measurements and calculations, and programs the adjust memory 405 accordingly. The converter 331 receives the GADJ signal, determines a corresponding initial value for the GCON signal, and makes any necessary adjustments utilizing adjust values in the adjust memory 405 if provided. The adjust value may be incorporated in any one of several manners. In one configuration, the adjust value is a multiplier value which is multiplied by the initial conversion value determined by the gain converter 331 and the result is used as the GCON signal to control the DC amplifier 333. Alternatively, the adjust value is an offset or additive value that is added to or subtracted from the initial conversion value determined by the gain converter 331 and the sum is used as the GCON signal to control the DC amplifier 333. In alternative embodiments, the calibration block 401 may simply assert a calibration signal that is used for adjusting the GCON signal. The calibration signal is maintained at a current adjust level as controlled by the calibration block 401, and utilized by the gain converter 331 in a similar manner as a single adjust value. It is noted, however, that due to the unpredictable and often non-linear nature of DC voltage in ZIF architecture embodiments in many environments, a single-valued calibration signal may not be sufficient to compensate for DC.
Any suitable number of adjust values are contemplated to achieve any desired level of accuracy. In one configuration, a low gain adjust value and a high gain adjust value are used. The particular GADJ value that is used as the threshold to switch between the low and high adjust values may be determined arbitrarily or experimentally. A half-way point is contemplated, although any suitable threshold point is contemplated depending upon the particular configuration and design. Alternatively, any number of adjust values could be used for each of a corresponding number of gain steps or levels of the GADJ signal. If the AGC DAC 297 is a 7-bit DAC with 128 corresponding discrete gain steps, for example, then the number gain adjust values programmed into the adjust memory 405 may range from 2 to 128 different adjust values. The embodiment of a different adjust value for each discrete gain step of the GADJ signal provides relatively high accuracy.
Upon timeout of the timing logic 403, the control logic 343 determines whether the wireless transceiver 200 is performing any communication functions, such as sending or receiving any packets of information. If not busy, or after the wireless transceiver 200 has performed its functions and becomes idle, the control logic 343 indicates to the calibration block 401 to conduct a calibration procedure. The calibration block 401 is coupled to control certain functions of the wireless transceiver 200, as indicated by dotted line 351, to conduct calibration. During calibration, the calibration block 401 may control the LNA/mixer 301 and/or the combiner 303 to temporarily remove the input signal RBB from the compensation system 300. For example, the input of the combiner 303 that receives the RBB signal is temporarily disconnected from the LNA/mixer 301 and grounded or otherwise left floating. It is noted, however, that control of the LNA/mixer 301 and/or the combiner 303 may require a more complicated interface between the ZIF transceiver 201 and the baseband amplifier 203. Alternatively, the calibration procedure and functions may be wholly contained within the baseband processor 203. In either case, the calibration block 401 controls the DC DAC 337 and the AGC DAC 297 to temporarily disable the AGC loop 345 and the DC loop 347 to conduct the calibration procedure.
During the calibration procedure, the calibration block 401 controls the DC DAC 337 to inject a predetermined or arbitrary DC level or setting into the compensation system 300, such as a DC value of zero volts or any other predetermined or suitable voltage. The calibration block 401 controls the AGC DAC 297 to a predetermined gain step or level, or otherwise to consecutively step through one or more gain steps. The calibration block 401 then samples the RD signal one or more times for each gain step. From this data, the calibration block determines one or more gain values or otherwise determines one or more DC offset values corresponding DC levels of the RD signal. In some embodiments, the calibration block 401 makes any necessary calculations based on measured parameters, such as DC levels of the RD signal and/or gain of the BB AGC amplifier 307, and programs the adjust memory 405 with one or more adjust values accordingly.
In one embodiment, the calibration block 401 controls the DC DAC 337 to assert an arbitrary and known DC voltage to the combiner 303 and controls the AGC DAC 297 to a particular gain level. The calibration block 401 then samples the RD signal. The calibration block 401 may use an assumed or theoretical gain of the BB AGC amplifier 307, calculate the theoretical value of the RD signal based on the gain level, and compare with the actual value of measured RD signal. The difference between the theoretical gain value and the measured value is a DC differential value that is utilized to determine the necessary DC offset at that gain level. Alternatively, the calibration block 401 determines one or more gain values representing measured gain of the BB AGC amplifier 307 in any appropriate manner, and uses the determined gain values rather than theoretical gain in the calculation. For example, the calibration block 401 may determine first and second DC offset values programmed into the DC DAC 337 that result in the RD signal reaching corresponding first and second range values, such as 75% upper and lower range values, respectively. As described further below, the first and second DC offset values are utilized to calculate a gain value of the BB AGC amplifier 307 at the particular gain step of the AGC DAC 297.
The calibration block 401 repeats the process for one or more gain steps of the AGC DAC 297 resulting in a plurality of gain values and DC differential values that are used to determine the adjust values programmed into the adjust memory 405. The calibration block 401 may incorporate any differences in gain ranges or gain scales between the BB AGC amplifier 307 and the DC amplifier 333 to convert each DC differential value to the appropriate adjust value. For example, conversion from logarithmic to linear scales is contemplated. Alternatively, the adjust memory 405 is programmed with the DC differential values, where the gain converter 331 performs any necessary conversions during operation. Although the calibration block 401 and the adjust memory 405 are additional components that further complicate the circuitry and logic, the DC loop 347 converges and eliminates the DC offset more quickly. The calibration procedure is repeated on a periodic basis since the operating environment is dynamic and unpredictable.
The calibration block 401 performs any necessary calculations on the measured gain values and/or DC offset values to convert between the AGC loop 345 and the DC loop 347 to program the gain conversion values within the LUT 501. In particular, the calibration block 401 converts between corresponding ranges or units along with any bit weighting of the GADJ and GCON signals as previously described. In the alternative or in addition, the calibration block 401 further converts between different scales, if any, between the BB AGC amplifier 307 and the DC amplifier 333, such as between logarithmic and linear scales as previously described. The calibration block 401 programs a separate gain conversion value within the LUT 501 that corresponds to at least one gain level or step of the AGC DAC 297. A separate gain conversion value for each gain step is contemplated, which provides an advantage during operation in that no conversion is necessary so that the LUT 501 supplies an appropriate GCON value for every GADJ value.
Instead of directly applying the RE signal, the limit block 601 provides a way of limiting the amount of change of the GADJ signal to prevent loss of loop control, as further described below. Many different embodiments are contemplated. In one embodiment, the limit block 601 is preprogrammed with a predetermined gain change limit that always applies, such as a 90% factor applied to the RE signal or the like, or a constant maximum gain change limit that is not exceeded during operation of the wireless transceiver 200. Alternatively, the limit block 601 is programmed by the calibration block 401 via one or more GLIM signals or the like, in a similar manner described above for the LUT 501.
It is possible that the ADC 313 include as many bits as necessary to capture most, substantially all, or all possible ranges of the receive signal R asserted by the BB AGC amplifier 307. In a particular embodiment, the BB AGC amplifier 307 has a gain range of approximately 66 dB (−6 dB to 60 dB) to provide a sufficient dynamic gain range to enable appropriate operation in the anticipated environments. The AGC DAC 297 includes a sufficient number of bits to achieve the desired accuracy over the entire gain range of the BB AGC amplifier 307. For example, in one embodiment, the AGC DAC 297 is 7 bits. Such configuration would require that the ADC 313 has a significantly large number of bits to track the receive signal R regardless of the gain level of the BB AGC amplifier 307. It is further noted that the ADG 313 could further include additional bits to further incorporate the entire range of DC offsets incorporated within the receive signal that might not yet be completely compensated for by the DC loop 347. Again, this is a substantially large number of bits that would require a large ADC 313. Further, the baseband processor 203 could be designed to appropriately handle the significantly large number of bits of the ADC 313.
In a more practical embodiment, the ADC 313 includes only as many bits as necessary to maintain signal integrity with sufficient head and foot room (collectively, head room). In one embodiment, for example, the ADC 313 is a 6-bit converter with a 36 dB of range resulting in approximately 6 dB per bit. In this manner, the ADC 313 does not attempt to handle the entire operating range of the BB AGC amplifier 307, but instead is sufficient to maintain signal integrity with sufficient head room. The target power block 323 asserts the TP signal at the appropriate target power in an attempt to maintain the receive signal R with reduced or eliminated DC offset to within the operating range of the ADC 313. It is noted that the target power is less than the full scale of the ADC 313 and in one embodiment is set to approximately ±75 percent of the full scale power observable with the ADC 313.
The ZIF transceiver 201, therefore, may provide a received R signal that is beyond the range of the ADC 313 of the baseband processor 203. This may be particularly true during initial packet acquisition. For example, an excessive DC level may cause the ADC 313 to rail to either its maximum or minimum limits. An overpowered input receive signal may cause clipping at either or both rails depending upon corresponding DC level. The AGC loop 297 and the DC loop 347 are provided to compensate for signal power and DC level to control the level of the RD signal to within the desired target power and operating range of the baseband processor 203. Furthermore, even after the RD signal converges to the desired operating range, the environmental conditions may suddenly change causing a change of signal power and/or DC level. Since the BB AGC amplifier 307 has unpredictable DC at its input that may change significantly with gain level changes, significant gain changes requested by the signal power estimate block 315 may otherwise cause a substantial change of the DC level of the RD signal causing a rail condition of the ADC 313 and/or loss of DC loop 347 control. This is particularly problematic at high gain levels, since any new DC level is substantially amplified at high gain by the BB AGC amplifier 307 potentially causing quick loss of loop control.
The limit block 601 is provided in the compensation system 600 to reduce or eliminate over-correction by the signal power estimate block 315. At higher level gain steps of the AGC DAC 297 corresponding to higher gain levels of the BB AGC amplifier 307, a certain level of the REST signal may correspond to a significant amount of differential of the GADJ signal and the AGC DAC 297, which could introduce too much DC offset for control by the DC loop 347. Alternatively, a significant change in the GADJ signal may substantially reduce DC offset otherwise supplied by the BB AGC amplifier 307, such that the accumulated DC offset signal itself over-compensates and thus introduces too much DC voltage that may not be controllable. Thus, the limit block 601 reduces the amount of correction of the RE signal at higher gain steps. The limit block 601 compares the RE signal with the GACC signal and determines the level of the RLE signal to maintain a controllable modification of the GADJ signal to maintain proper control loop operation.
Many different embodiments and variations of the limit block 601 are contemplated. In one embodiment, the limit block 601 is pre-programmed with a single maximum gain differential value that is applicable for all operating gain levels. Alternatively, the limit block 601 is programmed by the calibration block 401 via the GLIM signals during calibration with one or more gain limit values. A single maximum gain change limit GM is contemplated as determined and programmed by the calibration block 401. One or more gain change limits may also be determined and programmed depending upon particular gain steps. The limit block 601 may be programmed with a different gain change limit for each gain step of the GADJ signal, operable in a similar manner as the LUT 501. Alternatively, the limit block 601 may be programmed with gain change limit values only at higher gain levels, such as the higher 30 dB gain levels. Alternatively, the limit block 601 may be implemented with any combination of gain limitations, including a maximum gain change limit operative across the entire gain range and particular gain change limits at particular gain levels, such as the higher gain levels.
As mentioned previously and as further described below, in one embodiment, the calibration block 401 measures or otherwise determines a gain value representing the gain of the BB AGC amplifier 307 for each gain step of the GADJ signal (or gain step of the AGC DAC 297) during each calibration procedure. The calibration block 401 may then use an upper bound method that further utilizes an assumed DC offset model or characteristic of the BB AGC amplifier 307. In the assumed model, the change of DC offset (dV) is linear with linear gain change (dG), so that it is assumed that the rate of change of the DC offset is constant with respect to voltage gain, or dV/dG=k, where prefix “d” denotes a derivative or delta value. Using the model, it is assumed that the rate of change of DC offset with respect to voltage gain is the constant value “k”. Although this may not be true for particular configurations of the BB AGC amplifier 307, this model is sufficiently accurate from an upper bounding perspective. The linear DC offset change model, when converted to the decibel range (dV/dGdB), is no longer linear and changes more rapidly with increasing gain dB. However, over small ranges of gain, the linear approximation is a relatively accurate estimation of the change of DC offset at any given gain point. The rate of change is relatively small at lower gain points but begins to increase rapidly at the mid range point of approximately 30 dB. Thus, correction may be applied over the entire gain range or at the higher gain levels.
To achieve a relatively accurate slope for dV/dGdB, it is desired to find the differences in DC and gain at two relatively close points. For convenience, the two points are the current operating gain G1 dB and the gain at one (1) dB greater, or G2 dB=G1 dB+1. In this manner, dGdb=1 dB, which is simpler for implementation purposes and sufficiently accurate. To find dV (or delta DC), the linear curve with constant “k” is utilized at two gain points by converting between the linear model and dB scales. First, the linear gain G1=invlog G1 dB=10^(G1 dB/20), where the “^” symbol denotes a power factor or “raised to the power of”. The first DC offset DC1=k*G1, where an asterisk “*” represents multiplication. If G2 dB=G1 dB+1, then the second DC offset DC2=k(G2)=k10^[(G1 dB+1)/20]. Thus, dV=DC2−DC1=dV/dGdB.
The gain change limit, referred to as MaxGainStep, is the step in gain from an initial operating point that results in extra DC that is the largest observable at the ADC 313. If DC is initially fairly well compensated, then the largest tolerable extra DC is half the range of the ADC 313, referred to as MaxDCStep. In this manner, MaxGainStep for the current gain step is MaxGainStep=MaxDCStep/(dV/dGdB). The ADC 313 has a range of approximately ±500 mV, so that MaxDCStep=500 mV is an acceptable limit value for the embodiment shown. In one embodiment, a separate dVi, denoted with index “i”, is measured for each gain level or gain step (i) of the BB AGC amplifier 307, or each gain step of the GADJ signal. A corresponding MaxGainStepi value is then determined for each gain step (i). In one embodiment, all of the MaxGainStepi values are stored within the limit block 601 by the calibration block 401. For an ADC DAC 297 of 7 bits, 128 different MaxGainStepi values are calculated and stored. Alternatively, a subset of MaxGainStepi values are calculated and stored, such as the gain steps corresponding to the upper 30 dB gain range of the BB AGC amplifier 307. In operation, the current gain as determined by the GACC signal is used to access the corresponding MaxGainStepi value, if any, stored within the limit block 601 in a similar manner as a lookup table embodiment. The RLE signal is asserted by the limit block 601 as the lesser of the current RE signal and the corresponding MaxGainStepi value. In another alternative embodiment, each dVi is calculated, but only the maximum value is stored, or dVM. Then, a maximum gain change limit, or GM, is calculated using dVM, and stored in the limit block 601 and used over the entire gain range.
In another alternative embodiment, the linear model is not used in favor of actual DC offset measurements. In the embodiment shown, the AGC DAC 297 is a current-based, 7-bit DAC with GD_lsb=approximately 2/128 mA per step or click for a full-scale range of 2 mA, where GD_lsb represents the least significant bit (lsb) conversion value for the AGC DAC 297. If “i” is an index representing the gain steps of the GADJ signal corresponding to the AGC DAC 297, if Vo(i) represents a measured DC offset at a particular gain level at operating point (i), and if dGmA is the gain change in milliamps, then dV/dGmA(i) for each gain step (i) is determined according to the following equation (1):
where the quantity Vo(i+1)−Vo(i−1) is a local DC offset change over two gain steps as would be observed after the BB AGC amplifier 307 at the input to the ADC 313. The conversion from mA to dB is nearly linear, where dV/dGdB(i) for each gain step is determined by conversion of dV/dGmA(i) according to the following equation (2):
where maxGdb(i) and minGdb(i) are the maximum and minimum measured gain values in decibels, respectively. Then dV/dGdB(i) is determined according to the following equation (3):
Prior to discussing specific operation, a brief discussion of the overall operation is provided. As described before, the calibration block 401 controls the AGC DAC 297 and the DC DAC 337 to perform measurements while the wireless transceiver 200 is idle and not communicating. A successive approximation algorithm is utilized to establish a voltage difference at the BB AGC amplifier 307 input that corresponds to a desired voltage difference of the RD signal. During the calibration procedure, the AGC DAC 297 is stepped through each gain step (i) beginning from lowest to highest. Then, for each gain step, the DC DAC 337 is adjusted to obtain a target level of the RD signal output by the ADC 313. In the particular embodiment described, high and low values of the RD signal are determined by successive approximation and stored for each gain step of the AGC DAC 297. The use of two separate target levels of the RD signal enables calculation of the gain using differences in which the DC offsets are common mode and cancel out. Also, utilizing summation in conjunction with the previously calculated gain, the actual DC offset voltage may be calculated.
For the successive approximation, the value of the DC DAC 337 is initially started low and successively increased until the RD signal is sampled at a predetermined high value. Then, the DC DAC 337 is initially started high and successively decreased until the low value of the RD signal is sampled at a predetermined low value. A successive approximation or binary search method is used in order to converge on the desired value very quickly. In particular, each bit beginning with the most significant bit (MSB) to the least significant bit (LSB) is flipped and examined independently until the RD signal is less than or equal to the high value or greater than or equal to the low value.
At block 801, DC_DAC is initially set equal to ALL_ZERO which means that all 12 bits of DC_DAC are set equal to binary zero. In signed 2's complement form, this also represents a binary and a decimal value of zero (0). In the first iteration of the calibration procedure, the DC DAC 337 is increased from the lowest value of −2048 and successively increased until the RD signal becomes equal to a ADC_HI_RANGE constant, which in the embodiment shown is +24. If the DC DAC 337 is to increase from −2048 towards the maximum value of +2047 in the successive approximation, then it is first set approximately equal to the mid-range value of zero. At next block 803, the GainStep variable is written to the AGC DAC 297 to set the gain of the BB AGC amplifier 307 at its lowest level. At next block 805, the DC_DAC variable is written to the DC DAC 337. In this manner, in the first iteration, the AGC DAC 297 is at the lowest gain step and the DC DAC 337 is set equal to zero or at its mid-range level.
At next block 807, the RD signal is sampled one or more times by the calibration block 401. At next block 809, the arithmetic mean value of valid samples taken at block 807 is stored in a variable ADC_Mean. In the embodiment shown, 48 samples are taken at block 807 and the first 16 are discarded, so that the last 32 samples are considered valid. Thus, the mean value of the last 32 samples is determined and ADC_Mean is set equal to the determined mean value.
At next block 811, it is queried whether ADC_HiLo is equal to 1. Since ADC_HiLo was set equal to one at block 801 during the first iteration, operation proceeds to next block 813, in which it is queried whether the ADC_Mean variable is greater than the ADC_HI_RANGE constant. As described previously, each bit of the DC DAC 337 using the DC_DAC variable is successively tested for each gain step, which in the first iteration is the minimum gain step of GS_MIN, until the RD signal reaches the ADC_HI_RANGE constant. If the ADC_Mean value has increased beyond ADC_HI_RANGE as determined at block 813, then operation proceeds to next block 817 in which the Next_Bit of the DC_DAC variable is discarded. To discard a bit, its value is changed or “flipped”, where if it is 0, it is changed to 1 and vice-versa. In effect, the bit is discarded if the mean value has overshot the target value of ADC_HI_RANGE constant. For example, in the first iteration where Next_Bit is equal to 12 in the embodiment shown and initially set to 0, if the mean value ADC_Mean is greater than the ADC_HI_RANGE constant (+24), then the 12th or most significant bit of the DC_DAC variable is flipped “back” to 1 to discard the bit. If this occurs, then DC_DAC becomes equal to 100000000000b or the minimum value of −2048. Otherwise, if the ADC_Mean value is not greater than the ADC_HI_RANGE constant, then block 817 is skipped so that the Next_Bit or bit 12 remains at 0.
After the Next_Bit of DC_DAC is flipped at block 823, operation proceeds back to block 805 to write the new value of DC_DAC to the DC DAC 337. Operation loops in this manner between blocks 805-823 via block 813 for each bit of the DC_DAC variable until Next_Bit is decremented to zero as determined at block 819. When this occurs, the DC_DAC variable, when programmed into the DC DAC 337 at the lowest gain step of the AGC DAC 297 provides the RD signal as close as possible to, and less than or equal to, the predetermined ADC_HI_RANGE constant. Operation then proceeds to block 825 where the ADC_HiLo variable is queried. Since ADC_HiLo is still 1 in the first iteration, operation proceeds to next block 827 in which an indexed variable DC_DAC_HIi is set equal to the DC_DAC variable for storage and/or calculation. Also at block 827, a mean value variable ADC_HIi is set equal to the current value of ADC_Mean. It is noted that although ADC_Mean should be equal to the ADC_HI_RANGE constant, it may vary somewhat and any variation is stored for use in calculations.
Operation proceeds to block 829 from block 827 to reset variables for the second half of the first iteration of the successive approximation for the current gain step (i). In particular, Next_Bit is set back equal to MSD_BIT, ADC_HiLo is set equal to 0 and the DC_DAC variable is set equal to a binary variable ALL_ONE meaning that all bits of DC_DAC are set equal to one. This initializes the variables for the second half of the calculation for the current gain step in which the DC DAC 337 is started at the maximum value or +2047 and successively decreased according to the successive approximation algorithm until a predetermined ADC_LO_RANGE constant is achieved as determined at block 815. It is noted that DC_DAC is effectively set equal to −1 since 111111111111b in sign 2's complement form represents −1which is approximately half way between the maximum DAC value of +2047 and the minimum DAC value of −2048.
From block 829, operation proceeds back to block 805 where the new DC_DAC value is written into the DC DAC 337. Again, one or more samples of the RD signal are taken at block 807 and the mean value of valid samples are determined and stored into the ADC_Mean variable at block 809. At next block 811, since ADC_HiLo is equal to 0, operation proceeds to block 815 in which it is determined whether ADC_Mean is less than the ADC_LO_RANGE constant. If not, operation proceeds directly to 819 and if so, operation proceeds to next block 817 where the Next_Bit is discarded in a similar manner as previously described. If the Next_Bit is not discarded during the first iteration, then operation proceeds with a DC_DAC value of −1 or 111111111111b. If the Next_Bit is discarded at block 817, then the DC_DAC value is set equal to +2047 or 0111111111111b, which is the maximum DAC value.
At next block 819, it is queried whether Next_Bit is equal to 0 and if not, operation proceeds to blocks 821 and 823 where, as before, Next_Bit is decremented and the corresponding bit of the DC_DAC variable is flipped for evaluation. In a similar manner as before, if DC_DAC is equal to −1 or 111111111111b, then at block 823 DC_DAC is next set equal to −1025 or 0111111111111b. Alternatively, if DC_DAC was +2047 or 011111111111b, then at block 823 DC_DAC is set equal to +1023 or 001111111111b.
Operation loops in a similar manner for the low to high successive approximation between blocks 805-823 via block 815 until Next_Bit is decremented to zero as determined at block 819. At this point, the ADC_Mean should be close to, and greater than or equal to, the ADC_LO_RANGE constant as determined at block 815. After all bits have been examined, operation proceeds to next block 825 in which the value of ADC_HiLo is queried. Since this is the second half of the first iteration at the current gain step, where ADC_HiLo is 0, operation proceeds to next block 831 where the DC_DAC variable is stored into an index variable DC_DAC_LOi for the current gain step. Also, the current value of ADC_Mean is stored into an index value ADC_LOi at block 831. At next block 833, the variables are initialized for the first half of the next iteration of the successive approximation procedure, as tracked by the GainStep variable. In particular, Next_Bit is set equal to 12, ADC_HiLo is set back equal to 1, and DC_DAC is set back equal to ALL_ZERO. Furthermore, GainStep and the index “i” are both incremented. As described previously, GainStep is initially set to GS_MIN or −64 in the embodiment shown. Thus, at the first occurrence of block 833, GainStep is set equal to −63 and “i” is incremented to 2. At next block 835, it is queried whether GainStep has become greater than the GS_MAX constant to determine if calculations have been completed for all gain steps. In the first iteration of block 835, operation proceeds back to block 803 where the new GainStep value is written into AGC DAC 297. The entire first and second halves of the successive approximation utilizing the DC_DAC variable is repeated for the next GainStep. In this manner, for each GainStep, the indexed variables DC_DAC_HIi, ADC_HIi, DC_DAC_LOi and ADC_LOi are determined and locally stored if necessary.
After all calculations for each GainStep have been determined in which GainStep is incremented at block 833 to be greater than GS_MAX as determined at block 835, then operation proceeds to block 837 as shown in
where ADC_lsb is the step value conversion of the ADC 313, DC_DAC_lsb is the step value conversion of the DC DAC 337 and where ADC_HIi, ADC_LOi, DC_DAC_HIi and DC_DAC_LOi are the indexed variables previously determined at block 827 and 831. In the embodiment shown, ADC_lsb is approximately equal to 1 Vpp/64 or 15.625×10−03 volts per step and DC_DAC_lsb is approximately equal to 150 mV/4096 or 36.62×10−03 mV per step. 128 gain data values G(i) are determined in the embodiment shown. At next block 839, a filter is applied to smooth the gain data G(i). In one embodiment, a five-wide moving average filter is applied across the gain data G(i) to smooth out the curve. A three-wide filter is used at the end points for the second and second to last values and the end point values are not changed. In particular, the first three values are averaged and the second value is set equal to the average. Then, the first five values are averaged and the third value is set equal to the average. Then, the second through fifth values are averaged and the fourth value is set equal to the average. Operation of the five-wide moving average continues in this manner until the last five values are averaged and the third to the last value is set equal to the average. Then, the last three values are averaged and the second to last value is set equal to the average, and the filter operation is completed. A filter is utilized since the measured gain values may be calculated in a very noisy environment where substantial variations and inaccuracies would otherwise result.
Operation proceeds from block 839 to block 841 in which a calculated DC offset value Voffset(i) is determined for each gain step (i), where Voffset refers to the calculated DC offset at the input to the BB AGC amplifier 307. A simplified version of the DC offset value, denoted Voffset(i)S, is determined according to the following equation (5):
The simplified version assumes that the ADC_HIi and ADC_LOi variables are set to the ADC_HI_RANGE and ADC_LO_RANGE constants, respectively. In operation, however, the ADC_HIi and ADC_LOi variables may vary by a step or two, so that an adjust value ADJi is calculated for each gain step (i) according to the following equation (6):
The adjust value is utilized to calculate more accurate DC offset values Vo(i) according to the following equation (7):
It is noted that for each case in which ADC_HIi is equal to ADC_HI_RANGE and where ADC_LOi is equal to ADC_LO_RANGE, that the adjust value ADJi cancels and becomes zero. Otherwise, the adjust value is utilized to improve accuracy. Operation proceeds to block 843 to smooth out the DC offset data Voffset(i) in a similar manner as described above for block 839, such as the five-wide moving average filter with the three-wide filter at both ends.
At next block 845, DC offset derivative data dV/dGdB(i) is determined for each gain step (i) by first calculating the DC offset value at the output of the BB AGC amplifier 307 according to the equation Vo(i)=Voffset(i)*G(i), and then plugging each Vo(i) value into equation (3) previously described. At next block 847, the DC offset derivative data dV/dGdB(i) is smoothed in a similar manner as previously described, such as with the five-wide moving average filter with a three-wide filter used at either end points. At next block 849, the measured and/or calculated data is converted and/or otherwise stored depending upon the particular configuration. For example, for the compensation systems 500-700, the gain data G(i) is stored in the LUT 501, for the compensation system 700, the DC offset data Vo(i) is stored in the LUT 701, and for the compensation system 600, the derivative data dV/dGdB(i) is used to determine the maximum gain change limit GM or one or more gain change limits for one or more gain steps. The determined gain change limit value(s) are stored in the limit block 601.
It is appreciated that removal of the IF portion of a high performance wireless transceiver, with proper DC compensation and DC calibration, results in a relatively high performance, low cost wireless ZIF transceiver with reduced power requirements. The use of a calibrated DC compensation loop interfaced to the gain loop achieves these goals. Estimation of DC at the output of the baseband gain amplifier and amplification of the DC estimate based on a gain conversion linked with the gain loop provides a DC offset at the input to effectively control DC provided to the baseband processor decoder devices. The calibration procedure provides additional accuracy by periodically measuring actual gain and DC offset and storing one or more gain adjust values or DC offset adjust values. The adjust values are utilized during operation to modify the gain conversion signal or to limit gain change to maintain loop control. Such capability enables a WLAN system to be designed for use at home or in the small business that is relatively robust and that has significant performance with relatively high data throughput operation. A ZIF design in accordance with embodiments of the present invention provides high sensitivity and allows for fast settling of the gain and DC offset loops. Fast settling is desirable because of the bursty and packetized nature of communications. The design also contends well with large amounts of inherent noise which is averaged out or otherwise eliminated in order to compensate for DC offsets.
As described above, a calibration procedure is initially conducted upon power up to measure and store parameters to prepare the wireless transceiver 200 for packet acquisition. Changes in the wireless medium may result in the need for gain changes. Although DC is not transmitted over the wireless medium, changes in the receive radio front end may result in the need for both gain and DC changes. Because the wireless medium is dynamic and unpredictable and because changes may occur in the receive radio front end, the calibration procedure is repeated periodically.
Even though the calibration procedure, if employed, facilitates stability, the conditions of the wireless medium are constantly changing and the wireless transceiver 200 must detect the changes and operate the AGC loop 345 (gain feedback control loop) and the DC loop 347 (DC feedback control loop) to track the changing conditions of the wireless medium. The wireless transceiver 200 detects several contemplated condition changes and performs procedures, such as by the control logic 343, the signal estimate power block 315, the DC estimate block 319 and other control logic, to identify the changed condition and take the appropriate response. The contemplated changed conditions include onset of a new packet in the wireless medium causing DC saturation of the ADC measurement window, completion of transmission of a packet in which the wireless device 200 must perform clear channel assessment (CCA), and completion of transmission of a packet by the wireless device 200 itself, in which case the wireless transceiver 200 must prepare to acquire an acknowledgement (ACK) packet if transmitted by the receiving device. As described further below, the wireless transceiver 200 has a limited amount of time to detect each condition change, to determine whether an appropriate response is necessary, and to take the appropriate response.
For each changed condition situation described herein, the wireless transceiver 200 holds the gain feedback control loop at a constant gain level and operates the DC feedback control loop in an attempt to search a stable DC value for the receive signal. The ZIF transceiver 201 continuously processes energy in the wireless medium to generate and provide a receive signal to the baseband processor 203. Such processing is performed even when there are no packets being transmitted on the wireless medium, so that the gain and DC feedback control loops effectively track the noise floor of the wireless medium. Onset of a new packet usually cases the gain loop to make gain adjustments in an attempt to achieve the target power level for packet acquisition, which could lead to gain saturation. Several gain iterations may be required since the range of the ADC 313 is limited as compared to the total possible signal range. Such changes in gain may cause DC saturation in which the sampled signal rails to maximum or minimum levels of the ADC measurement window. Thus, a very limited amount of information is initially available to the control logic 343, the signal power estimate block 315 and the DC estimate block 319.
An attempt to continue operating both the gain and DC loops to acquire the new packet may result in instability since gain change also causes a DC level change. Instead, upon DC saturation, the AGC loop 345 is held constant and the DC level is searched during initial packet onset. Since the noise floor level is not expected to change appreciably after transmission of the packet, the noise floor gain setting, and optionally the DC noise setting, are stored prior to onset of the packet. As soon as packet transmission is completed, the noise gain setting is retrieved and applied to the gain loop and the gain loop is once again held while DC is searched. Optionally, the DC noise setting, if stored, may also be retrieved and applied to the DC loop as a starting point.
The gain of the variable LNA 261 is initially set high to detect weak signals and is switched to low gain upon certain conditions if the received signal causes an overload condition as detected by the overload detector 289, for example. If an overload condition is detected and the LNA 261 is “tripped” and switched to low gain, this low gain condition is initially held until it is determined whether a packet is being transmitted. If a packet is not detected, a noise spike or the like may have caused the overload condition and the LNA 261 is allowed to switch back to high gain mode if and when the noise condition disappears. If a packet is detected, the state of the LNA 261 may be allowed to change during the preamble of the packet. The LNA 261 may not trip initially upon detection of a packet, but is allowed to trip during the preamble in the event of a strong packet signal. After the preamble portion of the packet is completed, the state of the LNA 261 is held and not allowed to change during the payload portion of the incoming packet.
As described above, in one embodiment the ADC 313 includes only as many bits as necessary to maintain signal integrity with sufficient head room. For example, the ADC 313 may be only a 6-bit converter with a 36 dB of range resulting in approximately 6 dB per bit. The BB AGC amplifier 307, on the other hand, has a significantly greater gain range (e.g., −6 dB to 60 dB) to provide a sufficient dynamic gain range to enable appropriate operation in anticipated environments. In this manner, regardless of the actual signal strength at the receiving device, the existence of DC may cause the signal to initially appear outside the limited 36 dB ADC measurement window range. It is desired to control the AGC loop 345 and the DC loop 347 to resolve the incoming signal as quickly as possible to within the target power range (e.g., 75 percent of the full scale power) of the ADC 313. The signal power estimate block 315 and the DC estimate block 319 monitor the RD signal output from the ADC 313 to determine whether conditions have changed and/or a packet is being received.
Assuming initial DC conditions are correct and that DC conditions do not substantially change, the signal power estimate block 315 controls the LNA 261 and the AGC loop 345 to bring the signal within target power level. The DC estimate block 319 remains operative to control any DC level changes. Regardless of signal strength, a substantial change in DC level, for whatever reason, may saturate the ADC 313. DC saturation, by itself, does not necessarily cause on overload condition to trip the LNA 261 to low gain since the AC power of the signal itself may be weak. The DC saturation causes a rail condition of the ADC 313, which completely obscures the incoming signal. A DC saturation or rail condition means that all or substantially all of the digital samples are at one “rail” or the other, although not at both rails (which may indicate a gain saturation condition, described below). Two rails are defined including a maximum digital value and minimum digital value at either end of the ADC measurement window. It is desired to reduce or otherwise eliminate DC and set gain so that the signal is at a target power level in which all or substantially all of the digital samples occur within the rails with sufficient headroom to enable correction for signal fluctuations. A DC threshold condition is defined in which a percentage of digital samples that occur at one rail is at or below a predetermined threshold percentage. In one embodiment, the threshold percentage is 90%, so that the DC threshold condition is exceeded when more than 90% of the samples occur at one rail and met when 90% or less of the samples occur at one rail. Since the signal strength is initially unknown and since gain changes affect DC, the AGC loop 345 is initially held at the existing gain setting while the DC loop 347 compensates for DC during the rail condition. The DC saturation or rail condition is corrected by adding a sufficient level of DC offset of the opposite polarity. For example, if the ADC 313 rails positive (or negative), then a sufficient level of negative (or positive) DC offset is applied at the combiner 303 to substantially center the RD signal within the range of the ADC 313.
There are several methods that may be employed to eliminate the DC saturation condition. A first method is a step search procedure in which a predetermined DC step amount is repeatedly added until the DC threshold condition is met, or when the percentage of clipped samples drops below the threshold percentage, such as 90% or any other arbitrarily determined threshold percentage. The predetermined DC step amount may also be arbitrarily determined, but should be selected in such a manner to avoid the possibility of overstepping or completely bypassing the ADC measurement window, causing an opposite rail condition. In one embodiment, the predetermined DC step amount is related to the size of the ADC measurement window, so that the added DC does not move the signal by more than the relative size of the ADC measurement window. For example, if the DC step amount is chosen to move the signal by an amount approximately equal to the size of the ADC measurement window, then if the signal is just above an upper rail of the ADC measurement window, the added DC step moves the signal to just above the lower rail of the ADC measurement window. In alternative embodiments, the DC step amount corresponds to any other portion of the ADC measurement window size, such as half size or quarter size, etc.
The step search procedure is advantageous in that it is relatively simple to implement, is relatively well-behaved and converges on the incoming signal in a reasonably sufficient amount of time. Another technique is successive approximation or a binary search method in which the DC range between the existing ADC measurement window and the maximum level range is searched in a binary fashion. For example, the rail condition indicates a positive or negative direction, and the successive approximation approach determines the maximum possible signal range in that direction and then tries full range, half range, quarter range, etc. The direction of each successive binary step is indicated by the rail condition until the signal is detected within the ADC measurement window. The successive approximation approach is bounded in that no more than a predetermined maximum amount of time is consumed no matter how much error exists upon packet onset.
The step search method may be faster than successive approximation in certain instances, such as for small signal onset cases with a modest DC change resulting from a small gain change. The step search method, however, may also take significantly longer for large signal onsets with large gain changes that may produce larger DC offsets. The binary search algorithm of the successive approximation approach, on the other hand, is bounded and will converge to a specified limit within a known uncertainty. The use of one method over the other depends on the expected nature of the input conditions and the consequences when the input is not as expected. For initial packet onset, it may be preferable to use the step search because the critical small packets are acquired faster, which leaves more time for post AGC/DC processing. Since there is adequate time to ensure convergence, the consequence for large packet onsets is acceptable. For the end of packet scenarios, less time is available and bounded convergence time is most important. Trading longer acquisition times to gain rapid acquisition of smaller packets may not be desired.
Eventually, the rail condition of the RD signal is removed by DC compensation and the signal is either within range or changes to a clipping condition that meets the DC threshold condition previously described, such as clipping at one end or both ends but still less than or equal to the threshold percentage (e.g., 90%). If an unbalanced clip condition still remains in which clipping at one rail is still present at or below the specified DC threshold condition, then the gain feedback control loop is released and both the DC and gain loops are operated in a normal mode using the unbalanced clipped signal to eventually lock onto the incoming signal. The DC estimate block 319 controlling the DC loop 347 remains operative to control DC as before. In this case, even though the clipped signal may not be an accurate representation of the actual signal, the estimated gain and DC are close enough so that the actual signal should be acquired sufficiently fast enough with normal loop operations. If the power level of the incoming signal is high or otherwise if the peak-to-peak levels are beyond the range of the ADC 313, then the clipping condition may be reduced but may not be resolved solely by operation of the DC loop 347. As long as the percentage of clipped digital signals (at both ends of the gain range) is at or below a predetermined clip rate threshold, then the gain feedback control loop is released and both loops operate in a normal mode to acquire the incoming signal. Again, even though the clipped signal may not be an accurate representation of the actual signal, the estimated gain and DC are close enough so that the actual signal should be acquired sufficiently fast enough with normal loop operations. The clip rate threshold is any suitable arbitrary value, such as 50% or the like. In various embodiments, the AGC loop 345 is unlocked when the clipping condition is substantially balanced when the DC condition has been corrected or when the ratio of clipping between the positive and negative rails of the ADC 313 is below or within the predetermined clip rate threshold.
If, on the other hand, the percentage of clipped digital signals (at both ends of the ADC measurement window range) is above the predetermined clip rate threshold and if clipped samples at both rails is relatively balanced (e.g., if the number of digital samples at the maximum level is roughly equivalent with the number of digital samples at the minimum level), then a gain saturation state or condition is detected and the signal power estimate block 315 enters a clipping mode employing a clip-step procedure to reduce the power level of the received signal to eliminate clipping. The gain saturation condition indicates a significantly high amount of clipping occurring at both rails. The balance condition is any suitable and arbitrary ratio between the minimum and maximum clipped samples, such as 3:1 or 2:1 or the like. If the balance condition is defined as a 3:1 clip ratio, for example, 75% of the clip samples occur at one rail while the remaining 25% at the opposite rail, which would still qualify as “balanced” even though the 3:1 ratio is a relatively rough balance criterion. In the clipping mode, the signal power estimate block 315 samples the RD signal and determines the amount of clipping present. In one embodiment, the signal power estimate block 315 determines the number of values or samples at maximum range (positive or negative rail) compared to the number of samples less than maximum range for a given total number of samples to estimate the power level based on the clipping information. The signal power estimate block 315 then adjusts gain to reduce the power level. Thus, the signal power estimate block 315 determines the number of rail (or clip) samples for a total number of samples to estimate power level.
A graduated set of thresholds may be employed during the gain saturation condition. If the relative number of clip samples is high, then a higher gain adjustment is used whereas if the relative number of clip samples is low, then a lower gain adjustment is used to resolve the power level of the received signal. For example, a graduated clip gain adjustment may be employed based on clip sample ratios, such as 7/8 (high clipping, high gain adjust), 3/4, 5/8, 1/2, 3/8, 1/8 (low clipping, low gain adjust), etc. Any number of thresholds or break points may be defined or utilized. The amount of clip gain adjust values may depend upon the particular configuration. For example, larger gain adjustments may be used for the calibrated compensation system 700 as compared to other calibrated compensation embodiments since the compensation system 700 more accurately compensates DC in response to gain changes. During the clipping mode, the DC estimate block 319 and the DC loop 347 remain operative to control the DC level. When the clipping condition is eliminated, the AGC loop 345 and the DC loop 347 are both operated as normal to obtain and maintain the RD signal at the target power level.
It is noted that once the wireless transceiver 200 enters the clipping mode to eliminate the gain saturation condition, there are several options as to when to exit the clipping mode in favor of the normal mode to complete gain and DC lock on the signal. In one embodiment, as soon as the clip rate is below the clip rate threshold, such as, for example, when the total clipping rate drops below 50%, then the gain saturation conditions are no longer met and the normal mode is entered from the clipping mode. In an alternative embodiment, the wireless transceiver 200 may remain in the clipping mode until the clipping condition is substantially removed even after one or more of the gain saturation conditions are no longer true.
A next mode of operation is end of packet re-acquisition (of the wireless medium) which includes two sub-modes including clear channel assessment (CCA) and receive (RX) priority for reception of an expected acknowledge (ACK) packet. An ACK packet is often transmitted by a “receiving device” that has just received a packet from a “transmitting device”. ACK packets are highly recommended for most types of information transfer in wireless systems but are not always necessary or even desired. If an ACK packet is expected to be sent, the receiving device, after successfully receiving a packet, transmits the ACK packet. If an ACK packet is expected by the original transmitting device, it attempts to acquire the ACK packet if sent. The transmitting device may resend the packet if an ACK packet is expected but not received from the receiving device. Thus, a transmitting device expecting an ACK packet must detect and acquire the ACK packet if sent. The other devices in the network, if any, need only detect the presence of, but do not need to acquire, the ACK packet in the wireless medium.
ACK packets are given preferential treatment in that there is a predetermined reserved period of time for ACK transmission immediately after a packet has been sent. In particular, after a packet is received successfully, the receiving device is provided a certain amount of time to send the ACK packet. The device transmitting the ACK packet need not perform CCA but simply transmits the ACK packet as soon as possible and within the reserved period of time for ACK transmission. A device expecting the ACK packet, such as the wireless transceiver 200, enters the ACK receive priority mode as soon as possible after transmitting a packet. As shown in
During the ACK receive priority mode, the expecting device holds its noise floor gain setting during the quiet period between indication of the end of the original packet 1003, occurring at time T5, and the detected onset of the ACK packet 1007 at the baseband processor, occurring at time T5. For wireless Ethernet configurations, this period is approximately 7-9 μs at a minimum. In one embodiment, the gain and DC settings prior to transmission of the original packet are stored by the transmitting device. As described previously, in one embodiment, prior to transmission of the packet 1001, the wireless transceiver 200 tracks the noise floor of the wireless medium while no packets are being transmitted and determines and stores a noise floor gain setting, and optionally a DC noise setting.
To enter the ACK receive priority mode after transmission of the packet, the transmitting device immediately retrieves and resets the stored gain and DC settings, such as the noise floor gain and DC noise settings. While the gain setting is frozen and held constant, the ACK-expecting device searches the DC level until the quiet period is over. Much of the DC uncertainty can be determined during the quiet period. In other words, the stored gain is retrieved and applied to the AGC loop 345, which is held at the retrieved gain value during interim period while DC is searched. Then, when the quiet period is over, the AGC loop 345 is released and both loops are operated in the normal packet acquisition mode, described previously, to acquire the ACK packet. It is noted that the DC loop 347 may or may not converge during the quiet period. Regardless of whether it has converged or not, both loops are operated together after the quiet period in normal mode so that the ACK-expecting device can receive and acquire the expected ACK packet if it has been sent. In the Ethernet embodiments, the ACK packet should be initiated within a 20 μs period. If the ACK packet is not received, the device may or may not re-send the original packet depending upon higher-level operations. It is noted that a receiver of a device other than the device that transmitted the packet may also desire to operate in ACK priority mode, such as a network monitor the operates in “sniffer” mode or the like used to analyze protocol.
After a packet has been transmitted from the transmitting device to the receiving device, any remaining devices (third party devices) in the same wireless medium enter the CCA mode to determine when the channel is clear to transmit if they have a packet to send. If any packet, including an ACK packet, is transmitted in the wireless medium during CCA, then the remaining devices must back off until the end of the packet to conduct CCA once again.
The gain and DC loops of the wireless transceiver 200 must lock onto the energy in the wireless medium at a time T3 in order to have sufficient time to complete the CCA determination by the time T4. In the embodiment shown, the time T3 is placed at approximately 10 μs before the time T4 in order to complete the CCA assessment, so that the wireless transceiver 200 has approximately 9 μs between times T2 and T3 to lock. As described previously, the wireless transceiver 200 determines and stores a noise floor gain setting prior to transmission of the packet 1101. In the CCA mode, the noise floor gain setting is retrieved and applied in the gain feedback control loop as soon as possible following transmission of the packet 1101, such as at time T2 after the receiver delay. The noise power level of the medium may change somewhat, but is not expected to change appreciably after only one packet. The retrieved noise floor gain setting is then held, and the DC level is searched by the DC feedback control loop. In effect, the AGC loop 345 is jammed and held, while a step procedure or binary search or successive approximation or the like is conducted by the DC loop 347 during the CCA mode.
If there is no packet signal in the channel, then the channel is clear and the DC loop should converge to a stable value by the time T3. If, however, the DC loop does not converge by the time T3, then there is either a packet and channel is not clear or the wireless transceiver 200 is otherwise unable to make the CCA assessment and declaration in time. If a signal is not present, the DC loop 347 typically converges relatively quickly. In one embodiment, the AGC loop 345 is unlocked after approximately 9 μs and packet acquisition is conducted normally. Although an ACK packet, if present, may not be acquired by a third party device, it is of no consequence since third party devices need not acquire the ACK packet. If the DC loop 347 does not converge at the time T3, even after the AGC loop 345 is released, then the channel is deemed to be busy.
A tracking mode is also contemplated, which is a slower, higher level and longer term mode as compared to the modes described above. The tracking mode is a steady state operation and is employed to track the channel over time. A significant amount of data is reviewed over a substantial amount of time and finer, more precise adjustments are made in the operation. In general, steady state conditions are monitored and longer term trends are tracked and adjustments made to follow the channel conditions. Updates to the steady state operating conditions are made more slowly at longer time intervals, such as 10 μs or the like. Short-term trends may be tracked, but adjustments are made more slowly to avoid over-compensation.
It is noted that data is stored over several successive short cycles and then evaluated. If the error between current settings, such as gain and DC settings or the like, is relatively small, then any such error is stored, but an adjustment is not necessarily made at the next update interval. In this manner, a predetermined “null zone” or threshold error level is employed, so that updates and changes are not made unless and until the measured error exceeds the predetermined null mode values. In other words, updates are made only when a significant amount of error exists to avoid overcompensation and chattering during operation. The null zone operation is even more significant at higher gain settings, because even small gain changes at high gain levels may have a significant impact on operation. In one embodiment, gain adjustments are not made at all if the gain is above a certain threshold level.
Although a system and method according to the present invention has been described in connection with one or more embodiments, it is not intended to be limited to the specific form set forth herein, but on the contrary, it is intended to cover such alternatives, modifications, and equivalents, as can be reasonably included within the spirit and scope of the invention.
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|U.S. Classification||455/232.1, 455/239.1, 455/343.2, 455/234.1|
|International Classification||H04B1/06, H04B7/00, H04B1/30, H04B1/16, H03G3/30, H04L25/06, H03D3/00|
|Cooperative Classification||H03L7/0812, H04L25/061, H03D3/008, H04B1/30, H03L7/085|
|European Classification||H03L7/081A, H03L7/085, H03D3/00C1, H04B1/30, H04L25/06A|
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Owner name: INTELLECTUAL VENTURES I LLC, DELAWARE
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