WO1996003919A1 - Method and apparatus for a baseband processor of a receive beamformer system - Google Patents

Abstract

This invention presents a multi-beam base band processor (R-125) for making post-beam formation adjustments to the complex (imphase/quadrature) pre-detection scan line samples acquired from a receive beam former (R-100) of an ultrasound imaging system (R-20). Specifically, the invention includes a programmable finite impulse response (FIR) filter and a programmable complex multiplier. The programmable filter performs signal shaping and sample rate conversion. The signal shaping capability may be used to compensate for transducer and analog signal path responses, to increase signal to noise ratio (SNR) by rejecting out of band noise frequencies, to operate as a matched filter to the transmitted waveform shape, and to act as both a smoothing and anti-alias filter during rate conversion operation. The programmable multiplier permits both amplitude and phase adjustments of the complex data that assures amplitude and phase coherency between acquired scan lines, compensating for the effects of a selected scan format or the differences in ultrasound frequencies used to produce each scan line.

Description

METHOD AND APPARATUS FOR A BASEBAND

PROCESSOR OF A RECEIVE BEAHFORHER SYSTEM

BACKGROUND OF THE INVENTION

This invention relates to coherent imaging systems including, for example, radar, sonar, seismic, and ultrasound systems, using vibratory energy, and in particular, but not limited to, phased array ultrasound imaging systems for scan formats such as, by way of example only, linear, steered linear, sector, circular, Vector^®, steered Vector^® in imaging modes such as, by way of example only, B-mode (gray-scale) imaging mode and color Doppler imaging mode. In particular, this

invention describes post-beamformation signal processing adjustments. Although the invention will be discussed with respect to an ultrasound system, the invention can be implemented with other types of coherent imaging systems.

Beam-to-beam coherency is not required in most prior art ultrasonic imaging systems, although the need for channel-to-channel coherency in phased array imaging systems is well understood. In prior art systems, a requirement for image uniformity is that the amplitude response of the system to a point target at any range on any scan line be substantially identical to the amplitude response to the same target at the same range on an adjacent scan line. The additional requirement for beam- to-beam coherency further implies the phase response (jointly represented with the amplitude response by, for example, an in-phase and quadrature (I/Q) response) of the system to a point target at any range on any scan line also to be substantially identical to the phase response to the same target at the same range on an adjacent scan line. Systematic phase variations can arise in some scan formats. For example, if the

apertures associated with successive transmit and receive beams change relative to each other, systematic scan- line-to-scan-line phase variations can be introduced. Likewise, systematic scan-line-to-scan-line phase

variations can be introduced if the center frequencies of successive transmit and receive beams change relative to each other. This method requires range-dependent and scan-line-dependent phase correction or adjustments of such systematic variations. Such phase corrections or adjustments may be predetermined and stored in memory to be applied by a complex multiplier to the acquired coherent samples prior to further coherent operations, such as synthesizing new coherent samples.

It is also desirable to adjust for systematic phase variations to establish coherent phase alignment among pre-detected beams in a scan caused by differences in beam-to-beam transmit/ receive frequencies by

remodulating prior to detection. This is most

efficiently performed on the beamformed baseband I/Q signals .

Ultrasound imaging is accomplished by firing

(transmitting) into body tissue or other objects to be imaged a scan sequence of focused ultrasonic beams centered along straight lines in space called transmit scan lines. The transmit scan lines are generated by a transmit beamformer and an ultrasound transducer array. The transmit scan lines are spaced to produce a planar linear, planar sector or other display of the tissue via a pre-defined firing or scanning pattern. Focused to some defined depth in the tissue, the ultrasonic transmit continuous-wave (CW) or pulse-wave (PW) signal,

propagating at an assumed constant propagation velocity of nominally c=1540 m/sec through the tissue, interacts with the tissue and reflects a small portion of the signal back to the ultrasound transducer array that initiated the ultrasound signal. The round trip delay time is shortest for those targets closest to the ultrasound transducer array, and longest for those targets farthest from the transducer array. With the application of appropriate time delays, the receive beamformer can dynamically focus receive beams along straight lines in space called receive scan lines

commencing, for example, with the shallowest range

(depth) of interest and evolving toward the deepest range of interest.

Analog and hybrid (analog-digital) phased array beamformer systems are available in the known art. For example, phase array beamformer systems can be found in the following patents.

Digital receive beamformer systems have also been proposed in the art. The basic feature of a digital receive beamformer system can include: (1) amplification of the ultrasound signal received at each element of an array such as, for example, a linear array; (2) direct per channel analog-to-digital conversion of the

ultrasound signal with an analog-to-digital sampling rate at least twice the highest frequency in the signal; (3) a digital memory to provide delays for focusing; and (4) digital summation of the focused signals from all the channels. Other processing features of a receive

beamformer system can include phase rotation of a receive signal on a channel-by-channel basis to provide fine focusing, amplitude scaling (apodization) to control the beam sidelobes, and digital filtering to control the bandwidth of the signal. This art points out the ever present desire to achieve, in an efficient manner, a reconstructed image of high quality. SUMMARY OF THE INVENTION

The present invention relates to a method and apparatus which provides for a baseband processor for making post-beamformation adjustments to the complex (in- phase/ quadrature) pre-detection scan line samples acquired from a receive beamformer of an ultrasound imaging system.

In a preferred embodiment, tne invention includes a programmable finite impulse response (FIR) filter and a programmable complex multiplier. In one aspect of the invention, the programmable filter performs signal shaping. The signal shaping capability may be used to compensate for transducer and analog signal path

responses that distort the transmitted and received ultrasound waveforms. It may also be programmed to increase signal-to-noise ratio (SNR) by rejecting out-of- band noise frequencies. It may also be programmed to operate as a matched filter to maximize the output SNR relative to a selected transmit waveform shape. The invention is capable of operating with time-interleaved data samples associated with multiple beams.

In another aspect of the method and apparatus of the invention, the baseband processor may operate as a sample rate converter which converts the sample rate of the receive beamformer signal to a rate that is advantageous for an image display processor. In another aspect of the method and apparatus of the invention, the baseband processor includes a programmable complex multiplier which follows the programmable filter. The complex multiplier corrects systematic processing amplitude and phase differences between scan lines so that there is scan-line-to-scan-line (beam-to-beam) amplitude coherence (also called amplitude matching) and phase coherence (also called phase alignment). Such an aspect supports the novel feature of synthesizing samples and synthetic scan lines. The method and apparatus further supports adjustable frequency scanning in order to mitigate grating lobes. The complex multiplier in another aspect can provide frequency remodulation of scan line data to maintain scan-line-to-scan-line phase coherence as the frequency changes while scanning.

In still another aspect of this invention, the baseband processor can track receive signal frequency downshifting with tissue depth due to attenuation.

Additional advantages, objects, and novel features can be obtained from a review of the specification and the figures. BRIEF DESCRIPTION OF THE FIGURES

Figs, la and lb conceptually depict the transmission and reception of ultrasound beams to and from body tissue. Fig. 2a depicts a block diagram schematic of a novel ultrasound beamformer system of an ultrasound medical imaging system including an embodiment of a baseband processor system of the invention. Figs. 2b and 2c taken together depict a block diagram of the ultrasound beamformer system of Fig. 2a.

Fig. 3 depicts a block diagram of an embodiment of a digital baseband multi-beam processor for the receive beamformer of Fig. 2. Fig. 4a and 4b depict alternative embodiments of a baseband processor system. The first alternative in Fig. 4a switches the order of the baseband filter and complex multiplier. The second alternative in Fig. 4b moves the amplitude and phase adjustment into the receive

beamformer.

Fig. 5 is a block diagram of an embodiment of the baseband processor control of the invention which

provides for filtering and amplitude/phase adjustment among receive scan lines.

Figs. 6a, 6b and 6c depict graphs of typical signal frequency downshifting profiles that can be applied for signal remodulation in the baseband processor.

Fig. 7 illustrates the waveform response at the same range point for three ultrasound scan lines. Fig. 8 illustrates the method of phase accumulation at range r_t for a Vector scan geometry in which the phase differences for actual acquired scan lines (solid) at nominal line spacing are indicated. The dashed lines indicate target scan lines which were not acquired.

Fig. 9 demonstrates a plot (left) of the

differential phase for a sequence of acquired ultrasonic scan lines (usl) and a plot (right) of the accumulated (integrated) phase.

DETAILED DESCRIPTION OF THE PRESENTLY PREFERRED

EMBODIMENTS

The present invention comprises a component of a medical ultrasound imaging system for which additional patent applications, listed above, have been simultaneously filed in the United States Patent and Trademark Office.

A. Overview of Preferred Beamformer System Architecture

1. Ultrasound Signal Description

Referring now to the drawings, Figs, 1a and 1b depict representations of transmit and receive scan lines (solid) and straight-line signal propagation paths from individual elements (dashed), respectively. In Fig. 1a, the transmit beamformer is generally identified by T-50 with the transducer array T-52 containing a multiplicity of individual transducer elements T-54 organized as a linear phased array in this particular embodiment. As is known in the art, there are a great variety of transducer array configurations available for use with ultrasound transmit and receive beamformer systems. As can be seen in Fig. 1a, the transmit beamformer T-50 sends

appropriately time-delayed electrical signals to the individual transducer elements T-54. These transducer elements T-54 then in turn convert electrical signals into acoustic waves that propagate into the body tissue T-56. By applying different time delays to the

excitation signals sent to the individual transducer elements T-54, transmit scan lines T-60 and T-62, having respective foci r₁ and r₂, can be established. It is to be understood that each of these transmit scan lines is representative of a center line of a different transmit beam which is steered and focused into the body to be imaged.

The transmit beamformer T-50 can generate

simultaneous multiple beams along different scan lines, or different focal depths along the same scan line

(compound focus). Further, the multiple transmit beams can each scan the entire image format or be transmitted such that each of the multiple beams only scans a specified section of the image format.

Fig. lb depicts a digital receive beamformer R-58 which is also connected to the transducer array T-52. Also depicted in Fig. lb are receive scan lines R-64, R- 66 corresponding to a dynamically focused first receive beam and a dynamically focused second receive beam, respectively. The beams are sampled in range at a plurality of focal depths (r₁, r₂, r₃) along each scan line. In the digital receive signal path of the present invention, transducer array signals can be selectively separated into data representative of multiple individual beams.

Each scan line of a transmit or receive scan pattern can be parameterized by the origin on the transducer array, the scan line orientation (angle θ) and the focus depth or range (r). The ultrasound imaging system of the present invention stores a pre-computed sparse data set of focusing time delay and aperture apodization values indexed by these parameters (based on geometric

considerations as is known in the art) and expands the values by real-time computational means to control the transmit and receive beamformation systems that produce the desired scan lines.

2. Beamformer System

Figs. 2a, 2b, 2c depict an overall block diagram of a medical ultrasound imaging system R-20. Ultrasound system R-20 includes a beamformer system R-22, one or more transducers T-112, a display processing system R-26 with a display R-28 and an ultrasound imaging system control R-40. As used herein, the term ultrasonic refers to frequencies above the range of human hearing. However, the transducer arrays T-112 are optimized for frequencies typically within the range of 2-10 MHz.

In Figs. 2a, 2b, or 2c, the beamformer system R-22 includes inventive and novel (1) digital transmit beamformer system T-102, (2) digital receive beamformer system R-100, (3) beamformer central control system C- 104, (4) adaptive focusing control system G-100,

(5) Doppler receive beamformer system A-400, (6) baseband multi-beam processor R-125, and (7) coherent sample synthesizer S-100. These systems are depicted as

functional block diagrams. The blocks are abstracted from the actual implementation of a preferred embodiment in order to better illustrate the signal processing functions performed.

In Figs. 2a, 2b, 2c, the control signals are

communicated over the light lead lines while the signal paths are depicted with heavy lead lines.

As indicated in Figure 2a, beamformer system R-22 provides two sources of digital beam data to the display processing system R-26: (1) Doppler receive beamformer single-beam complex in-phase/quadrature data representing coherent temporal sampling of the beam (CW case) or coherent temporal sampling at one range location along the beam (PW case), and (2) digital receive beamformer multi-beam complex in-phase/quadrature data representing coherent sampling in range along each receive scan line. Beamformer system R-22 can be operated to provide a sequence of scan lines and associated samples as above to provide data for a variety of display modes. By way of example, possible display modes and their associated processors include (1) brightness image and motion processor R-30 for B-mode (gray-scale imaging) and M-mode (motion display), (2) color Doppler image processor R-32 for flow imaging, and (3) spectral Doppler processor R-34 for wide dynamic nonimaging Doppler velocity vs. time displays. Additional display modes can be created from the two complex data sources of R-22, as will be obvious to those skilled in the art.

Ultrasound system R-20 also includes a transmit demultiplexer T-106 for routing the output waveforms from the transmitters T-103 to the transducer elements T-114, a receive multiplexer R-108 for routing the input

waveforms from the transducer elements T-114 to the receivers R-101, one or more transducer connectors T-110 and transducer arrays T-112. Many types of transducer arrays can be used with the present system.

Ultrasound system R-20 also includes an ultrasound imaging system control R-40, archival memory R-38 for storing scan parameters and scan data, and operator interface R-36.

The transducer array T-112 is interchangeable with a variety of different kinds of transducer arrays,

including but not limited to linear, curved, curvi-linear and annular transducer arrays. A variety of transducer array shapes and frequencies are desirable in order to satisfy the requirements of a variety of different clinical settings. However, the transducer arrays T-112 are typically optimized for frequencies within the range of 2-10 MHz. The medical ultrasound system R-20 performs the three major functions of driving the ultrasonic transducer array of elements T-114 to transmit focused ultrasound energy, receiving and focusing back-scattered ultrasound energy impinging on the transducer array T- 114, and controlling the transmit and receive functions to scan a field of view in scan formats including (but not limited to) linear, sector or Vector^® format. 3. Digital Transmit Beamformer System

The digital transmit beamformer T-102 (Fig. 2c) is comprised of a plurality of digital multi-channel

transmitters T-103, one digital multi-channel

transmitters for one or more of the individual transducer elements T-114. The transmitters are multi-channel in that each transmitter can process, in a preferred

embodiment, up to four independent beams. Thus, for example, 128 multi-channel transmitters have 512

channels. In other preferred embodiments, more than four independent beams can be achieved. Processing more than four beams per processor is within the scope of the invention. In a preferred embodiment, each of the digital multi-channel transmitters T-103 produces as its output in response to an excitation event the superposition of up to four pulses, each pulse corresponding to a beam. Each pulse has a precisely programmed waveform, whose amplitude is apodized appropriately relative to the other transmitters and/or channels, and delayed by a precisely defined time delay relative to a common start-of-transmit (SOT) signal. Transmitters T-103 are also capable of producing CW.

Each digital multi-channel transmitter T-103

conceptually comprises a multiple beam transmit filter T- 115 which provides an output to a complex modulator T- 117. The output from complex modulator T-117 is

communicated to a delay/filter block T-119, and therefrom is provided to a digital-to-analog converter (DAC) T-121. The output of the DAC T-121 is amplified by an amplifier T-123. The multiple beam transmit filter T-115, the complex modulator T-117 and the delay/filter block T-119 comprise a digital multi-channel transmit processor T- 104. The transmit filter T-115 can be programmed to provide any arbitrary real or complex waveform responsive to a start-of-transmit (SOT) signal. The transmit filter T-115 is implemented with a memory which stores real or complex samples of any desired and arbitrary pulse waveform, and a means of reading the samples out

sequentially in response to the start-of-transmit (SOT) signal delayed by a component of the focusing delay. In a preferred embodiment, the memory of T-115 is programmed to store baseband representations of real or complex pulse envelopes.

Block T-115, although primarily a memory, is

referred to herein as a transmit filter, as the output of block T-115 can be thought of as the time response of a filter to an impulse. The complex modulator T-117 upconverts the envelope to the transmit frequency and provides appropriate focusing phase and aperture

apodization.

Delay/filter block T-119 conceptually provides any remaining focusing delay component and a final shaping filter. The digital-to-analog converter (DAC) T-121 converts the transmit waveform samples to an analog signal. The transmit amplifier T-123 sets the transmit power level and generates the high-voltage signal which is routed by the transmit demultiplexer T-106 to a selected transducer element T-114. Associated with each multi-channel transmit

processor T-104 is a local or secondary processor control C-125 which provides control values and parameters, such as apodization and delay values, to the functional blocks of multi-channel transmit processor T-104. Each local or secondary channel control C-125 is in turn controlled by the central or primary control system C-104. 4. Digital Receive Beamformer System

The digital receive beamformer R-100 is illustrated in Fig. 2b. The signals from the individual transducer elements T-114 represent return echoes or return signals which are reflected from the object being imaged. These signals are communicated through the transducer

connectors T-110 to the receive multiplexer R-108.

Through multiplexer R-108, each transducer element T-114 is connected separately to one of the plurality of digital multi-channel receivers R-101 which taken together with summer R-126 comprise the digital receive beamformer R-100 of the invention. The receivers are multi-channel in that each receiver can process, in a preferred embodiment, up to four independent beams.

Processing more than four beams per processor is within the scope of the invention.

Each digital multi-channel receiver R-101 can, in a preferred embodiment, comprise the following elements which are represented by the function block diagram in Fig. 2b. These elements include a dynamic low-noise and variable time-gain amplifier R-116, an analog-to-digital converter (ADC) R-118, and a digital multi-channel receive processor R-120. The digital multi-channel receive processor R-120 conceptually includes a

filter/delay unit R-122 and a complex demodulator R-124. The filter/delay unit R-122 provides for filtering and coarse focusing time delay. The complex demodulator R- 124 provides for fine focusing delay in the form of a phase rotation and apodization (scaling or weighting), as well as signal demodulation to or near baseband. The digital multi-channel receivers R-101 communicate with summer R-126 where the signal samples associated with each beam from each receive processor are summed to form final receive scan line samples, and the resulting complex samples provided to baseband processor R-125. The exact functioning and composition of each of these blocks will be more fully described below. A local or secondary control C-210 is associated with each digital multi-channel receiver R-101. Local processor control C-210 is controlled by central or primary control C-104 and provides timing, control and parameter values to each said receiver R-101. The parameter values include focusing time delay profiles and apodization profiles.

5. Doppler Receive Beamformer System The Doppler receive beamformer system A-400 for wide dynamic range, nonimaging Doppler acquisition includes analog receivers A-402, each of which receives echo signals from a respective one or more transducers T-114. Each of the Doppler receivers A-402 includes a

demodulator/range gate A-404 which demodulates the received signal and gates it (PW mode only) to select the echo from a narrow range. The analog outputs of the Doppler receivers A-402 are communicated to a Doppler preprocessor A-406. In preprocessor A-406, the analog signals are summed by summer A-408 and then integrated, filtered, and sampled by analog processor A-410.

Preprocessor A-406 then digitizes the sampled analog signal in an analog-to-digital converter (ADC) A-412.

The digitized signal is communicated to the display processing system R-26.

Associated with all Doppler receivers A-402 is a single local or secondary Doppler beamformer control C- 127. Doppler beamformer control C-127 is controlled by central or primary control system C-104 and provides control and focusing parameter values to the Doppler receive beamformer system A-400. The present beamformer system R-22 advantageously combines an imaging digital receive beamformation system R-100 and the nonimaging Doppler receive beamformation system A-400 in a manner which uses the same digital transmit beamformation system T-102 and the same

transducer array. This arrangement allows the digital receive beamformation system R-100 to be optimized for imaging modes such as B-mode and color Doppler imaging, and therefore has high spatial resolution, while the accompanying Doppler receive beamformation system has a wide dynamic range and may be optimized for use in acquiring signals for nonimaging Doppler processing.

6. Beamformer Central Control System The beamformer central control system C-104 of the present invention controls the operation of the digital transmit beamformer system T-102, the digital receive beamformer system R-100, the Doppler receive beamformer system A-400, the adaptive focusing control system G-100, and the baseband processor R-125.

The main control functions of the central control system C-104 are depicted in Fig. 2c. The control functions are implemented with four components. The acquisition control C-130 communicates with the rest of the system including the ultrasound system control R-40 and provides high level control and downloading of scanning parameters. The focusing control C-132 computes in real time the dynamic delay and apodization digital values required for transmit and receive beamformation, which includes pre-computed and expanded ideal values plus any estimated correction values provided by adaptive focusing control system G-100. The front end control C- 134 sets the switches for the demultiplexer T-106 and the multiplexer R-108, interfaces with the transducer

connectors T-110, and sets the gain and bias levels of all transmitter amplifiers T-123 and all receive

amplifiers R-116. The timing control C-136 provides all the digital clocks required by the digital circuits.

This includes the sampling clocks for all the transmitter DACs T-121 and receiver ADCs R-118.

In a preferred embodiment central control C-104 expands sparse tables of focusing time delay and aperture apodization values based on pre-computed and stored data, through such techniques as interpolation and

extrapolation. The expanded delay and apodization values are communicated to the local processor controls as a profile of values across the transducer aperture where the delay and apodization data expansion in range is completed to per-transducer-element, per-sample, per-beam values.

7. Adaptive Focusing Control System Adaptive focusing control system G-100 provides for real time concurrent adaptive focusing. Adaptive

focusing control system G-100 is comprised of an adaptive focus processor G-505 which provides focus correction delay values to the focus control C-132 of the central control C-104. Adaptive focus processor G-505 operates on output produced by aberration value estimators G-502 from data gathered from the subarray summers R-126 of the digital receive beamformer system R-100. Accordingly, aberration correction values, preferably aberration delay and amplitude values, are adaptively measured for each receive scan line or for a subset of receive scan lines in range regions corresponding to transmit focal depths by the adaptive focusing control subsystem G-100 shown in Figure 2c. It is to be understood that in addition to the adaptive focusing control system which adjusts focus delays, that a number of adaptive control systems are contemplated. These systems, by way of example, include (1) adaptive contrast enhancement control system for adjusting focus delays and aperture apodizations, (2) adaptive interference cancellation control for adjusting focus delays and phases, aperture apodizations, and (3) adaptive target enhancement control for adjusting focus delays and phase, aperture apodizations, imaging transmit and receive frequencies and baseband waveform shaping.

Another aspect of adaptive focusing which can be included in the preferred embodiment of the adaptive focusing control system G-100 is a geometric aberration transform device G-508/509 which can provide aberration correction delay values to the adaptive focus processor G-505 for scan lines and scan line depth locations for which measured aberration values were not collected by aberration value estimators G-502. More specifically, measured aberration correction values are written to a delay table in G-508/509. G-508/509 retrieves values from the delay table according to look-up rules of the geometric aberration transform to form focusing delay correction profiles across the aperture valid for depths, scan geometries, and acquisition modes other than the depth, scan geometry, and mode for which aberration correction values were measured.

8. Baseband Processor System

The baseband processor R-125 provides for filtering, and receive-scan-line-to-receive-scan-line (beam-to-beam) amplitude and phase adjustments as discussed herein. The baseband processor R-125 additionally includes a baseband filter, a complex multiplier, and a baseband processor control which controls the operation of the baseband filter and complex multiplier. The baseband processor control is controlled by central control C-104. 9. Coherent Sample Synthesizer System

The coherent sample synthesizer system S-100 is illustrated in Figure 2a. This system exploits the multi-beam transmit and multi-beam receive capability of the invention to acquire and store coherent (predetection) samples of receive beam data along actual scan lines and to perform interpolation of the stored coherent samples to synthesize new coherent samples at new range locations along existing scan lines or along

synthetically-created scan lines. Both acquired and synthesized samples are passed to the display processing system R-26.

10. Transmit and Receive Multiplexers The connectivity between the transducer array elements T-114 and the processors T-103, R-101, A-402 of the digital transmit, digital receive, and Doppler receive beamformer systems is established through a transmit demultiplexer T-106 and a separate receive multiplexer R-108, as shown in Figure 2a. The multiple- transducer multiplexer configuration shown in Figure 2a permits selection of transmit and receive apertures lying entirely within a single transducer array or straddling across two transducer arrays. The two multiplexers are independently controlled by the beamformer central control system C-104 and may be programmed to support a number of acquisition modes, including sliding aperture and synthetic aperture modes. B. Baseband Processor System Preferred Embodiments The digital baseband multi-beam processor R-125 (Fig. 3) comprises a baseband filter R-127, a complex multiplier B-254, and a baseband processor control C-270. 1. Baseband Filter and Complex Multiplier

The complex baseband beam signal (or time- interleaved beam signals in the multiple beam case) from the digital receive beamformer system R-100 is

communicated to the baseband multi-beam processor R-125. The baseband filter R-127 performs filtering, signal shaping, and/or sample rate conversion by a factor L/M, where L is the integer interpolation factor and M is the integer decimation factor. Complex multiplier B-254 provides for (1) scan-line-dependent and range-dependent amplitude and phase adjustments of the filtered I/Q signal B-260 required to correct for amplitude and phase differences resulting from scan-line-to-scan-line apodization changes, scan geometry, and non-aligned effective transmit and receive origins, and (2)

remodulation (frequency adjustment) of the filtered signal to correct for phase variations in range resulting from different transmit frequencies per scan line. The advantage of the use of a scan-line-to-scan-line variable frequency mode on transmit and receive beamformation is the reduction of grating lobes.

Such amplitude, phase, and frequency adjustments between scan lines, particularly two or more adjacent scan lines, is, for example, for purposes of creating coherent samples needed to implement coherent image processing techniques. Other post-beamformation

operations requiring phase-coherent samples between scan lines will also benefit from the adjustments provided by the complex multiplier.

Baseband filter R-127 preferably includes an

upsampler B-255 (Fig. 3), a downsampler B-257, and a multi-tap FIR filter B-256, designated by the impulse response sequence h4[n], which is programmable using either real or complex coefficients. The upsampler, working in conjunction with the filter, acts as an interpolator which increases the sample rate by an upsampling integer factor L. The downs.ampler, working in conjunction with the filter, acts as a decimator which decreases the sample rate by a downsampling integer factor M. When both the upsampler and downsampler are operated, the net sample rate conversion factor is L/M. The sample rate conversion is advantageous for adjusting sample rates to downstream processors, such as the display processing system R-26. In the preferred

embodiment, a 16-tap complex coefficient FIR filter will satisfy the various filtering operations for which the baseband filter may be programmed. During sample rate conversion operation, fractionally-shifted versions of the desired filter response in the coefficient registers of the filter are appropriately sequenced during filter operation.

Baseband filter R-127 can additionally accomplish three other tasks. First, baseband filter R-127 can be programmed to increase the signal-to-noise ratio by rejecting out-of-band noise frequencies.

Second, the baseband filter may alternatively be programmed to maximize the signal-to-noise ratio by programming as a matched filter or quasi-matched filter design, preferably for matching to substantially

Gaussian-shaped transmit pulses as well as pulses of other shapes. Gaussian-shaped pulses are especially useful as they represent waveforms that do not distort during transmission through attenuative media such as the body.

Third, baseband filter R-127 enables pulse

equalization and shaping by compensating for both the transducer frequency response and the analog signal path prior to the ADC R-118.

2. Baseband Processor Control

The filter coefficients h4[n], decimation factor M, and interpolation factor L for baseband filter. R-127 are programmed into baseband processor R-125 by being downloaded from the central control C-104 to coefficient and rate memory C-278 (Fig. 5) in the

baseband processor control C-270 (Fig. 3). The

downloaded filter coefficients and sample rate conversion factors can be changed at any time by introducing new coefficients and factors into the central control C-104. The coefficients and factors stored in the coefficient and rate memory C-278 are selectable by the central control C-104 for programming the filter and sample rate conversion ratio L/M of the baseband filter R-127.

Following complex multiplier B-254 is a register C- 296 which stores scan line sample data so that it can be reported to the DMA processor C-202 of the central

control C-104 for providing scan-line-to-scan-line

calibration. The complex multiplier B-254 includes a control function which is contained in baseband processor control C-270 (Fig. 5). In baseband processor control C-270, a scan-line-to-scan-line (beam-to-beam) amplitude

adjustment value A and a phase adjustment value Φ per sample of each beam is generated in a time-interleaved manner corresponding to the time-interleaved beam samples B-260. As discussed above, the phase correction value Φ is the sum of phase terms that include: (1) a phase adjustment term required to correct for phase differences due to scan-line-to-scan-line apodization changes, (2) a phase adjustment term for scan geometry which results in non-aligned effective transmit and receive origins (the scan-line-dependent and range-dependent phase adjustment term), and (3) a phase term required to remodulate the signal as though each scan line had been transmitted and received at a common carrier frequency. Using a

frequency scaling factor or frequency vernier factor, each beam can be produced with a different carrier frequency. The complex multiplier accordingly provides for remodulation of the received beam signals between scan lines so that all beams are adjusted to account for any differences in carrier frequencies.

In operation a source data set including scan format geometry parameters, sparse scan line gain and delay values, interpolation coefficients for expanding the sparse values, interpolation factor L, and decimation factor M are downloaded from the central control C-104 to the baseband processor control C-270. Additionally, frequency parameters used in the frequency profile generator of the central control C-104 in accordance with Figs. 6a, 6b and 6c are downloaded to the baseband processor control C-270. These profiles account for frequency attenuation with depth. The baseband processor control C-270 of Fig. 5 includes a gain and phase RAM C-280, a scan-line-indexed interpolator C-282 which is supplied with pre-calculated and pre-stored line interpolation coefficients (α _line) by the central control C-104, and a range-indexed

interpolator C-284 with a range accumulator C-286, which is supplied with a rational sample rate conversion factor L/M and a phase zone width, both of which values are precalculated and pre-stored in the central control C-104. The rational sample rate conversion factor L/M is the same L and M values supplied to the baseband filter R- 127. Additionally as is known in the art, a sample rate conversion in accordance with the rational sample rate conversion factor L/M is accomplished in order to match the input sample data rate to the baseband filter R-127 with the output sample data rate to the display

processing system R-26.

Alternatively the range-indexed interpolator/ extrapolator C-284 can be supplied with programmable interpolation/extrapolation coefficients which are, by way of example, either (1) pre-calculated and pre-stored in or calculated by the central control or (2) calculated locally in baseband processor control C-270 by a

coefficient generator. The baseband processor control C-270 also includes a remodulation frequency processor C-292 which is

preferably implemented as a double phase accumulator. The double phase accumulator calculates phase adjustment values to correct for scan-line-to-scan-line frequency differences and thus remodulates the signal as though a common carrier frequency had been used across all scan lines.

To understand the basis of the remodulation

operation, consider an idealized representation of a signal at the output of a beamformer which has been coherently summed across multiple elements, and has undergone modulation on transmit, demodulation on

receive, and coherent summation:

x( t-2r/c) = e ( t-2r/c) ._ej [ω_n · ( t-2r/c) ] · _e -j [ω_d · t] ( 1 ) where ,

e (t ) = a baseband I/Q signal envelope ,

ω_m = 27πf_m = a modulation frequency [MHz] ,

ω_d = 2π f_d = a demodulation frequency [MHz]

r = some imaging depth (range) [cm] . Note that the actual center frequency of the imaging pulse, x(t-2r/c), depends additionally on other things, such as tissue attenuation, filtering in the transmit and receive processing chains, and other effects not

explicitly considered in the above equation (1). Also not explicitly included in above equation (1) are the detailed representations of focusing delay and phase adjustments necessary for beamformation, though these could be surmised by those skilled in the art. This detail is not required to motivate the particular results presented here.

The transmit modulation frequency, the receive demodulation frequency, or both, may in general be range dependent. In particular,

ω_m = ω_m (R_t) , and ω_d = ω_d (R_r) ,

where

R_t = the distance from the active array center to the transmit focus,

R_r = the distance from the active array center to the receive focus.

For a system with dynamic focus, this means that ω_d is continuously updated. We now consider a scan line 1 corresponding to a modulation frequency ω_m ¹, a demodulation frequency ω_d ¹, and a post-beamformer remodulation frequency ω_r.¹; and an adjacent scan line 2, with respective modulation,

demodulation, and remodulation frequencies ω_m ², ω_d ², ω_r ². It can be shown that the post-beamformed phase difference between these two scan lines as a result of the different modulation, demodulation, and remodulation frequencies can be bounded by an amount Δv, where

Δv < (ω_m ²-ω_m ¹) ·T_ρ- [ (ω_d ²+ω_r ²) - (ω_d ¹+ω_r ¹ ] ·2R_r./C (2) where, T_p = the imaging pulse duration at any depth of the receive beamformer signal output .

This expression is valid at the receive focal depth, R_r, at the point of post-beamformer remodulation. It is again noted that there may be other terms apart from Δv which are needed to ensure phase coherence at the

beamformer output apart from the above equation (2) .

Examples of such other terms include, but are not limited to, terms which account for the offset in the beam origin, such as naturally arise in Vector^®, linear, and curved linear formats, particularly with end alignment. As expected, Δv = 0 when ω_m ² = ω_m ¹, ω_d ² = ω_d ¹, and ω_r ² = ω_r ^1. We now make the observation, from the above equation (2), that providing for remodulation at the post- beamformer, pre-detected output with a frequency ω_r permits scan-line-to-scan-line phase coherence by its proper selection. In particular, by selecting ω_r ¹ and ω_r ² such that

ω_d ¹ + ω_r ¹ = ω_d ² + ω_r ² (3) then the second term of equation (2) may be substantially ignored. Note that if ω_d is range dependent, such as would be the case for a range tracking system, then ω_r must also be range dependent.

The first term of equation (2), given by (ω_m ²-ω_m ¹) ·T_ρ may be readily managed by keeping (ω_m ²-ω_m ¹) sufficiently small. As an example, consider the requirement that Δv<π/4, and suppose that, as might be typical, the imaging pulse measured at the point of remodulation for a tracking focused system has a duration that is four cycles of the nominal modulation frequency. Then the required limit on scan-line-to-scan-line frequency

variation becomes approximately, from equations (2) and (3), f_m ² - f_m ¹ < f_m ¹/32. If the nominal modulation frequency is 5 Mhz, then the scan-line-to-scan-line modulation frequency difference is constrained to be less than 0.156 MHz, in this example.

Thus, if post-beamformation, pre-detection receive processing requires beam-to-beam phase coherence for all . beams in a scan, then the maximum transmit carrier frequency differential between any two beams in the scan should be chosen to meet the above criteria.

The above relationship (3) defining the remodulation frequencies is independent of the modulation frequencies on transmit. Such independence assumes that both the modulation signal and the demodulation signal for all transmit and receive channels are phase-locked to a common timing clock reference. That is, the phases of all such modulation and demodulation signals are defined relative to a common time reference. The above relationship (3) also assumes that the modulation frequencies on successive transmit scan lines and the demodulation frequencies on successive receive scan lines are each slowly varying to avoid 2ττ phase ambiguities. That is, f_d ¹ ≈ f_d ² and f_m ¹ ≈ f_m ². This

constraint is consistent with the problem being solved.

The above relationship (3) also assumes a "well- focused" system, wherein any observation made concerning a point in the field of view occurs at a time when the receive focus is at that point (i.e. tracking, or dynamic focus), regardless of whether a target is also at that point.

From the central control C-104, pre-calculated and pre-stored values representing the frequency differences between scan lines (delta frequency values) are sent to the remodulation frequency processor C-292. These frequency difference values are based on frequencies and frequency slopes such as specified in Figs. 6a, 6b and 6c. By way of example only, let it be assumed that the frequency profiles for two scan lines look like Fig. 6b but with different start frequency, F_start, values and different downshift slope, Δ F_downsloρe, values.

Accordingly, downloaded to baseband processor control C- 270 from the central control for the two scan lines are the difference in frequencies between the scan lines and the difference in the rate of change of the frequency profiles over time. These values are calculated by the acquisition processor C-130 based on stored parameters and dependent upon the particular rational conversion factor L/M currently being used. The first accumulator of processor C-292 accumulates the difference in the rates of change of the frequency profiles over time between scan lines while the second accumulator

accumulates the difference in the frequencies between the scan lines over time. If there is no difference in the rate of change of the frequency profile over time, (i.e. the profile are the same exact for initially different F_start values, or after T_break in figure 6b when the slope goes to zero) the first accumulator performs no function. With no difference in the rate changes of the frequencies between the scan lines, only the second accumulator accumulates the frequency differences over time resulting in a corrective remodulation phase value per sample. The phase adjustment due to scan-line-to-scan-line apodization changes, the phase adjustment due to scan geometry which results in non-aligned transmit and receive origins, and the phase adjustment due to

remodulating the signal to an effective common carrier frequency are added in a summer C-288 and the summed phase value is then converted in a look-up table C-290 to sine and cosine representations. As part of the look-up table 3 C-290 function, the gain is multiplied by the sine and cosine representations. This complex adjustment value Ae^jΦ is provided to complex multiplier B-254 for application to the beam signals B-260.

It is to be understood that other embodiments of the baseband processor control are possible within the scope of this invention.

As indicated above, the complex multiplier B-254 ensures that coherent signal relationships are maintained between scan lines. The transmit samples and the echo or receive samples of the signals from beams are defined as being coherent when sufficient information is stored, preserved, or maintained to enable the samples of the return signals to be phase and amplitude corrected from scan-line-to-scan-line. The process of actually making the phase and amplitude corrections need not have yet taken place, as long as sufficient information with respect to a reference is maintained.

When a signal sample is processed coherently, the processing continues to maintain sufficient information to perform phase and amplitude correction at a later time. When two or more samples are processed coherently (e.g., coherently summed), the phase and amplitude corrections necessary to achieve phase and amplitude coherence must have previously been performed.

Coherent processing of two or more signal samples yields significant benefits, such as being able to calculate synthetic samples, as described in the above co-pending application. Due to beamformer central control C-104 specifying and accounting for all aspects of the transmit and receive signal, the entire beamformer system maintains all signal samples as coherent samples throughout the transmit and receive signal path, until the signal is finally detected in an operation which is external to beamformation.

C. Alternative Embodiments

In the preferred embodiment, the amplitude and phase adjustments are accomplished by a complex multiplier following the baseband filter. If x[n] represents a data sample value to be adjusted, then x[n] -Ae^jΦ is the adjusted value, where A is the amplitude adjustment value and Φ is the phase adjustment value. The same adjustment can be accomplished by moving the complex multiplier in the signal processing path to a position prior to the baseband filter (Fig. 4a). Equivalently, the amplitude and phase adjustment can be provided by moving the

operation into each of the receive processors R-101 of a the receive beamformer R-100 (Fig. 2b). In particular, the demodulating value applied by each of the complex demodulators R-124 to each receive processor channel sample can be augmented by the amplitude and phase adjustment factor, as illustrated in Figure 4b. Other embodiments will be obvious to those skilled in the art.

D. Basis for Amplitude and Phase Adjustments

The necessity for amplitude and phase correction, applied after receive beamformation and baseband

filtering, can be understood by considering Fig. 7 which shows the scan geometry of two acquired collinear B-mode transmit/receive scan lines with a single target located between these scan lines, at range r along a scan line that passes through the target. The origins and angles of the scan lines are shown, with the dashed line shown on each acquired transmit/receive scan line pair

approximating the wavefront of the transmitted wave at range r_t, assuming the apodization used is symmetrical about the scan line origins. The target scan lines, although defined on the same grid as the acoustic

acquisition grid, need not coincide with one of the acquired transmit/receive scan lines and, therefore, must be synthetically created by, for example, interpolating between two neighboring acquired transmit/receive scan lines pairs.

The interpolated scan line should appear as if it were an actual scan line responding to the target, using samples at range r_t from the corresponding neighboring scan lines. However, the phase of the return signal at range r_t on each acquired transmit/receive scan line is not the same as the phase that would be measured at range r_t had an actual scan line been acquired coinciding with the target scan line. This is due to the slightly different paths traveled by the waves emanating from each acquired scan line, with origins essentially centered at their apodization centroids, to the target and back.

Fig. 7 shows that the two-way path length for scan line 1 is greater than 2r₁, while the path length for scan line 2 is less than 2r₁. An additional component contributing to a path length difference arises from the displacement of the apodization centroid from the scan line origin, and leads to additional phase error between the acquired signals along scan line 1 and scan line 2. Therefore, this phase difference between the scan line signals at range r_t from each acquired scan line exists and must be compensated before coherent processing such as

interpolation between scan lines is performed. This is illustrated by the return signal waveforms of Fig. 7.

Suppose that the phase measured at range r_t along scan line 2 differs by 180° from the phase measured at range r_t along scan line 1, given a target centered between scan line 1 and scan line 2 at range r_t. If the amplitudes of the return signals were approximately equal, then interpolation between scan line 1 and scan line 2 at range r_t would yield essentially a zero result. However, if an actual collinear transmit/receive scan line pair were fired along the target scan line, the non-zero response shown by the dashed waveform in Fig. 7 would be the actual result. In general, the phase difference to be compensated by the baseband processor is a function of range and scan line origin and angle.

In general, there are both amplitude and phase differences between acquired scan line signals at a range r, given a target range r along a target scan line, which depend in a complicated relationship on the scan format, array shape, target geometry with respect to the array, transmit/receive apodizations, transmit/receive

apodization end-treatment, single firing or multiple firing synthetic scan line formation, frequency, and waveform shape. An approximate closed form expression for the scan line relative response to a reference point amplitude and phase is based on the assumptions that (1) target scan line lies within the main beam of each of the respective scan line pairs, i.e., the transmit and receive scan lines are spaced less than or equal to the Nyquist spacing, and (2) the apodization function is a Gaussian weighting across the array aperture. The relative response at range r_t to an absolute reference phasor (Ae^jΦ) is another complex phasor that is a function of three major terms. The relative response phasor ρ(r_t) is given by the set of equations in Table 1. Note the list of dependent parameters at the end of Table 1, which include all typical scanning format parameters

encountered in ultrasonic imaging. To compensate for the effect of this difference, complex beam samples could, for example, simply be multiplied by the inverse 1/ρ (r_t) before coherent

operations on samples at a common range, but different scan lines, is performed. Alternately, only the relative phasor difference between two spatial points in the acquired scan line data needs to be corrected before coherent operations are performed with the data.

Assuming a uniform target has been intersected by the scan lines, let ρ₁ - A₁·e^{jψ 1} be the relative (to a

reference point) complex phasor response at a point 1 along some scan line, and let ρ₂ = A₂·e^jψ2 be the relative complex phasor response at another point 2 elsewhere (could be the on the same scan line or on another scan line). Ideally, one would like the signal processing performed by the beamformation to yield identical complex beam signal responses everywhere that the uniform target is intersected by the scan lines so that, for example, linear interpolation between any two scan lines samples would yield the identical complex response. When such a condition is achieved, the scan line signal responses are said to be "amplitude coherent" (because A₁ = A₂ at the two points) and "phase coherent" (because Ψ₁ = Ψ₂ at the two points). However, geometric conditions of

acquisition and the finiteness of apertures produces systematic effects that lead to different complex

responses, as Table 1 clearly indicates. One must at least adjust the "relative" response at two points, using for example "complex multiplication" of the point 1 response by ρ₂/ ρ₁ or "complex multiplication" of the point 2 response by ρ₁/ρ₂, before there is amplitude coherence and phase coherence between at least these two spatial locations. Because ρ₃/ρ₁ = (A₂/A₁)e^{j(ψ2 - ψ1)} or ρ₁/ ρ₂ =

(A₁/A₂)e^{j(ψ1 - ψ2)}, amplitude coherence is achieved by

"amplitude adjustment" using the factor A₂/A₁ or A₁/A₂, and phase coherence is achieved by "phase adjustment" by the factor (ψ₂ - ψ₁) or (ψ₁ - ψ₂). The process of achieving phase coherence is also called "phase alignment" because the relative phase between two spatial sample points after phase adjustment is such that (ψ₂ - Ψ₁) = O, that is, the phase responses are "aligned" after adjustment to yield a zero phase difference. The term complex response adjustment will also be used here to mean that one or both phase adjustment and amplitude adjustment operations are to be used, unless indicated otherwise.

Because a complex number c may be represented in polar coordinate form as c=a·e^jθ or equivalently as the in-phase and quadrature (I/Q) format c=x_I+jx_Q, in which a= (x_I ² + x_Q ²)^1/2, θ=arctan (x_I/x_Q), x_I=a·cosθ, x_Q=a· sinθ, the amplitude and phase adjustments to yield ρ·c using some complex value ρ (or ratio of ρ values) may be performed using a number of means. A "complex multiplier" is the most generic means for applying a complex adjustment factor ρ to one or more complex beam signal samples. If the polar format representation for ρ is used, then separate steps using a single "real multiplication" by the amplitude factor "a" and using a "phase rotation" by a phase factor θ are the appropriate enabling means. If the in-phase and quadrature format representation for ρ is used, then separate "real multiplications" by the two factors a·cosθ and a·sinθ are the appropriate means, in which case the θ factor is represented as a "phase shift" operation. Describing a "complex multiplier" here includes those arrangements involving or using real multipliers, phase rotators, and phase shifters.

The three term product of the phasor ρ(r_t) has a geometric term G_ρ, an apodization term A_ρ, and an origin term O_ρ. G_ρ is dominated by dependencies on the

transducer array and scan geometry and includes broadband pulse effects. Observe that for targets at increasing ranges from the transmit/receive scan line axis, term (1 + σ²β²) increases so that the two-way Gaussian pulse bandwidth and center frequency decreases due to the lowpass filtering effect of temporal dispersion. The term A_ρ is dominated by dependencies on the apodization which is assumed to have Gaussian shape. The origin term O_ρ is simply the product of the transmit and receive origin terms O_x and O_r and contains the beam loss and an additional phase shift term imposed on the beam due to its transmit/receive apodization origins x_ax and x_ar being displaced from the transmit/receive scan line origins x_ax and x_ar respectively. O_ρ also accounts for the effective apodization shifting due to element factor and

attenuation effects.

Denoting the relative phasor response at range r_t along scan line 1 as ρ₁ (r_t) =A₁e^jψ1 and along scan line 2 as ρ_z(r_t) =A₂e^jψ2, where A₁ and A₂ are the relative amplitude responses and where ψ₁ and ψ₂ are the relative phase responses, the amplitude difference that requires

amplitude adjustment of one or both scan line responses at range r_t in order to match amplitude response involves the gain factor A₂/A₁ (or A₁/A₂), and the phase difference that requires phase adjustment of one or both scan line responses at range r_t is the factor ψ₁-ψ₂ (or Ψ₂-Ψ₁).

There is one scan format that basically requires no amplitude or phase adjustments in order to maintain amplitude and phase coherence for purposes, as an

example, of interpolating new scan line samples.

Consider a sector scan format of a linear array in which the scan vertex is located at the center of the array. Also assume that the same imaging center frequency is used to produce all scan lines and a symmetric

apodization is applied. In this case,

x_ox = x_or = x_t = x_ax = X_or and the various difference

expressions, such as x_tox - x_t - x_ox, are all zero. In this situation, the equation in Table 1 yields phase terms

(having the form e^jψ for some phase argument ψ) which are essentially zero, and amplitude terms for a given range that are identical for all scan lines. Thus, the sector scan format generally requires no amplitude or phase adjustments as there is already nearly full coherence among scan lines at a given range.

Although the phasor equation of Table 1 provides an analytical expression for a systematic understanding of amplitude and phase adjustments needed to maintain inter- scan-line coherence, evaluating such an expression as part of a real-time scanning operation would be

prohibitive. An alternative approach amenable to real- time implementation is to precalculate amplitude and phase correction values evaluated on a sparse grid of reference ranges indexed by scan line number.

Interpolation into the sparse grid of amplitude and phase correction values for arbitrary range and/or arbitrary scan lines between indexed scan lines can then produce approximate amplitude and phase corrections that are applied by the complex multiplier B-246, or comparable apparatus such as real multiplier for amplitude and phase rotator for phase, to each sample of each scan line.

Consider a specific case of generating the phase adjustment. The phase portion of the correction is simply the difference of the individual scan line

relative phase responses.

ψ₂(r_t)-ψ₁(r_t) =-ΔΦ(r_t)

Although the phase difference ΔΦ can be computed across scan lines, it does not determine the individual scan line phases ψ₁ and ψ₂ needed to phase correct or phase align the individual scan line samples. Consider Fig. 8 which shows the phase differences for a group of scan lines spaced at a nominal scan line spacing, where the dashed lines indicate target scan lines. The phase ψ for a given phase correction reference scan line, which for single beam scanning modes is set equal to the target scan line, is determined such that the phase differences formed from neighboring ψ result in either using -ΔΦ directly, or is the result of a linear interpolation of neighboring phase differences. Starting from the center scan line and scanning to the right in Fig. 8, the phases are accumulated in the following manner. The phase ψ₂ is set to zero since the absolute phase reference is

completely arbitrary; ψ₂=-1/2·(ΔΦ₂₁+ΔΦ₃₂),

ψ₃=ψ₂-1/2·(ΔΦ₃₂+ΔΦ₄₃), etc. This is simply numerical

rential phases across reference scan lines, which in this discussion employs the

trapezoidal rule. Trapezoidal integration approximates the integral of a function by accumulating the areas of trapezoids. This is shown in Fig. 9 where the dashed line indicate the trapezoidal approximation to the continuous differential phase correction values indicated by the solid line. The resulting trapezoidal

integration is given mathematically by

and so forth.

During real-time B-mode imaging, the baseband processor will apply the range-varying accumulated phase correction values to each range sample of each receive scan line. If a given range sample does not coincide with a phase correction value in the precalculated table at a reference scan line and a reference range, the baseband processor control linearly interpolates among the accumulated phase correction values in scan line and range to produce a correction phase. Practical

consideration of memory size will limit the number of reference ranges and reference scan lines, so it is desirable to distribute the reference points in the range-scan-line grid in a manner which minimizes the phase error between desired phase correction values and interpolated phase correction values over the image region. The differential phase varies most rapidly near the transmit focus. Since the phase accumulation occurs across scan lines, the phase correction values will also vary most rapidly near the transmit focus. Therefore, it is advantageous to place a higher density of reference ranges near the transmit focus to accurately track phase variations in range by the piecewise linear approximation approach which has been described, and a lower density of reference ranges in regions which exhibit phase

variations which vary slowly with range. The

distribution of reference scan lines will therefore use non-uniform spacing, with the highest concentration of reference ranges where the phase changes rapidly.

f_c(r_s)=effectiveimagingcenterfrequency f_c0=imagingcenterfrequency(zerodepth) fd(rs)=demodulationfrequency ψ_d(r_s)=demodulationphase γ=attenuation r_x=transmitfocusrange r_r=receivefocusrange r_s=targetrange θ_xI= transmitusl angle θ_rI=receiveuslangle θ_sI=targetuslangle x_ox=transmitusl origin x_oτ=receiveuslorigin x_s=targetuslorigin x_ax=transmitgaussianapodizationorigin x_ar=receivegaussianapodizationorigin W_x=transmitgaussian apodizationwidth W_r=receivegaussianapodizationwidth d=elementspacing W_e=elementwidth R_xdcr==transducerradius N=#ofelements

Note: Angles are with respect to the array surface normal and cis the acoustic propagation velocity.

Claims

WE CLAIM:

1. A baseband processor for processing signals from an ultrasound receive beamformer which generates samples representative of an ultrasound signal,

comprising:

a complex response adjuster, said complex response adjuster operating on said samples to effect coherence between samples.

2. The baseband processor of claim 1 wherein the complex response adjuster comprises a phase adjuster, said phase adjuster operating on said beamformed samples to effect phase coherence between said samples, thereby creating phase coherent samples.

3. The baseband processor of claim 2 further comprising an amplitude adjuster which operates on the beamformed samples to effect amplitude coherence between said samples, thereby creating amplitude coherent samples.

4. The baseband processor of claim 2 wherein: said phase adjuster operates on the phase of the beamformed samples to effect phase coherence between samples on different scan lines.

5. The baseband processor of claim 2 wherein: said phase adjuster operates on the beamformed samples to effect phase coherence between samples on the same scan line.

6. The baseband processor of claim 2 wherein: said phase adjuster remodulates the beamformed samples on a receive scan line.

7. The baseband processor of claim 6 wherein said phase adjuster remodulates the beamformed samples associated with distinct beams using different

frequencies to effect beam-to-beam coherence.

8. The baseband processor of claim 2 wherein the phase adjuster comprises a phase rotator.

9. The baseband processor of claim 2 wherein the phase adjuster comprises a complex value multiplier.

10. The baseband processor of claim 1 further comprising a baseband filter.

11. The baseband processor of claim 10 wherein the baseband filter is connected to receive as inputs said beamformed samples and wherein said complex response adjuster is connected to receive the output of the baseband filter.

12. The baseband processor of claim 10 wherein the complex response adjuster is connected to receive as inputs said beamformed samples and said baseband filter is connected to receive the output of said complex response adjuster.

13. The baseband processor of claim 10 wherein the baseband filter comprises a programmable filter.

14. The baseband processor of claim 10 or 13 wherein the baseband filter comprises a sample rate converter.

15. The baseband processor of claim 10 wherein: said baseband filter increases the signal to noise ratio of said signals from said ultrasound

beamformer by rejecting out of band noise frequencies.

16. The baseband processor of claim 10 wherein said baseband filter is adapted to shape said signals from said ultrasound beamformer.

17. The baseband processor of claim 16 wherein: said baseband filter is adapted to produce an approximately Gaussian-shaped waveform.

18. The baseband processor of claim 1 or 10 wherein the complex response adjuster comprises:

(a) a phase adjuster, said phase adjuster operating on said samples to create phase coherent samples; and

(b) an amplitude adjuster, said amplitude adjuster operating on said samples to create amplitude coherent samples.

19. The baseband processor of claim 1 or 10 wherein the complex response adjuster comprises a complex-value multiplier.

20. The baseband processor of claim 3 or 18 wherein the amplitude adjuster comprises a real-value multiplier.

21. The baseband processor of claim 1 further comprising a sample rate converter.