WO1996017432A1 - Removing low frequency interference in a digital fm receiver - Google Patents
Removing low frequency interference in a digital fm receiver Download PDFInfo
- Publication number
- WO1996017432A1 WO1996017432A1 PCT/US1995/015578 US9515578W WO9617432A1 WO 1996017432 A1 WO1996017432 A1 WO 1996017432A1 US 9515578 W US9515578 W US 9515578W WO 9617432 A1 WO9617432 A1 WO 9617432A1
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- WO
- WIPO (PCT)
- Prior art keywords
- sample
- frequency
- filtered
- phase sample
- signal
- Prior art date
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Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D3/00—Demodulation of angle-, frequency- or phase- modulated oscillations
- H03D3/006—Demodulation of angle-, frequency- or phase- modulated oscillations by sampling the oscillations and further processing the samples, e.g. by computing techniques
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01N—INVESTIGATING OR ANALYSING MATERIALS BY DETERMINING THEIR CHEMICAL OR PHYSICAL PROPERTIES
- G01N27/00—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means
- G01N27/02—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means by investigating impedance
- G01N27/04—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means by investigating impedance by investigating resistance
- G01N27/12—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means by investigating impedance by investigating resistance of a solid body in dependence upon absorption of a fluid; of a solid body in dependence upon reaction with a fluid, for detecting components in the fluid
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01N—INVESTIGATING OR ANALYSING MATERIALS BY DETERMINING THEIR CHEMICAL OR PHYSICAL PROPERTIES
- G01N30/00—Investigating or analysing materials by separation into components using adsorption, absorption or similar phenomena or using ion-exchange, e.g. chromatography or field flow fractionation
- G01N30/02—Column chromatography
- G01N30/62—Detectors specially adapted therefor
- G01N30/64—Electrical detectors
- G01N30/66—Thermal conductivity detectors
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D3/00—Demodulation of angle-, frequency- or phase- modulated oscillations
- H03D3/001—Details of arrangements applicable to more than one type of frequency demodulator
- H03D3/002—Modifications of demodulators to reduce interference by undesired signals
Definitions
- the present invention relates to techniques for removing low frequency interference from a received angle modulated signal carrying digital information, and more particularly to techniques for removing this interference from phase samples of the received angle modulated signal.
- Techniques for communicating digital information by using it to modulate a carrier signal are well known.
- a modulator-demodulator (modem) is a well-known device designed for this purpose.
- modems In response to the increasing demand for mobile capability, modems have been designed with interfaces for communicating the information over a wireless medium, such as cellular telephone technology.
- Manchester coding also known as split-phase coding
- An example of one such system that utilizes this technique is the Advanced Mobile Phone Service (AMPS) system in North America.
- AMPS Advanced Mobile Phone Service
- Manchester coding is first applied by representing each information bit as a two-bit codeword, or symbol: a "1" bit is represented by the symbol "10", and a "0" bit is represented by the symbol ,, 01 M .
- the encoded digital information is then impressed upon the radio carrier frequency by means of frequency modulation (FM) .
- FM frequency modulation
- Manchester-coded digital FM signal are known.
- a preferred technique is disclosed in U.S. Patent Application Serial No. 08/053,860 by Paul W. Dent, entitled “Demodulator For Manchester-Coded FM Signals", filed April 29, 1993, which is expressly incorporated herein by reference.
- One of the characteristics of this preferred technique is the fact that, instead of applying the received radio signal to a frequency discriminator, as in the earlier prior art, demodulation is instead based on the behavior of the phase or complex vector value of the received signal.
- the polarities of the information bits may be determined by measuring the phase excursions in the middles of the Manchester symbols.
- a phase reference is established from a plurality of candidate phase references as a basis for comparison of the mid-symbol phase. The phase can be measured at the start-points and end-points of the symbols and averaged, or measured a plurality of times during each symbol period to generate a reference phase.
- an apparatus in a digital FM receiver having a demodulator that receives a radio signal and generates therefrom a phase sample whose behavior determines an output of the demodulator, the apparatus being for removing a low-frequency interference signal from the phase sample.
- the apparatus converts the phase sample into a frequency sample which is then high-pass filtered to produce a filtered frequency sample.
- the filtered frequency sample is then converted into a filtered phase sample, which may be further demodulated in accordance with known techniques. Converting the phase sample into a frequency sample may be performed by a first order difference circuit. Converting the filtered frequency sample into the filtered phase sample may be performed by an integrator.
- the demodulator may produce degraded performance when the inventive apparatus is applied to a radio signal that does not have low- frequency interference
- the demodulator is further provided with a circuit for detecting the presence of the low- frequency interference signal in the radio signal, and alternatively selecting, in response thereto, either the filtered or original phase sample to be supplied to further demodulation circuitry.
- this detection is performed by low-pass filtering the frequency sample and measuring the energy in the band where the low-frequency impairment would appear, if present. This measured energy value is then compared to a threshold value. If the measured energy exceeds the threshold value, then the low-frequency interference signal is present.
- this detection may be performed by including, within the demodulator, a memory for storing a table of base station system identification values (IDs) of those base stations known to generate the low-frequency noise.
- IDs base station system identification values
- the detection of signal impairment is then performed by comparing a received system ID with those contained in the table. A match indicates the presence of the low-frequency interference.
- the apparatus for removing a low-frequency interference signal from the phase sample is provided in a demodulator to produce a corrected demodulated signal.
- the demodulator is further provided with means for demodulating the radio signal without the benefit of the inventive apparatus, thereby producing a non-corrected demodulated signal.
- Each of the corrected and non-corrected demodulated signals is checked for error content (for example by performing a cyclic redundancy check on each one) , and an output from the demodulator is selected from the corrected and non-corrected demodulated signals on the basis of their respective error contents.
- FIGS. 1(a) and 1(b) show illustrative embodiments of a low frequency interference remover in accordance with one aspect of the invention
- FIG. 2 is a block diagram of an alternative, preferred embodiment of an inventive low frequency interference remover in accordance with the present invention
- FIG. 3 is a graph of test results for a demodulator utilizing the preferred embodiment of the inventive low frequency interference remover operated in flat fading with a vehicle speed of 100 km/hr;
- FIG. 4 is a block diagram of a demodulator that utilizes the inventive low frequency interference remover, and which further includes circuitry that is capable of detecting the presence or absence of impairment, in accordance with another aspect of the invention
- FIG. 5 is a block diagram of one embodiment of the impairment detector in accordance with the invention.
- FIG. 6 is a block diagram of an alternative embodiment of the impairment detector in accordance with the present invention
- FIG. 7 is a block diagram of a demodulator that incorporates another method for reducing the performance degradation associated with high-pass filtering an unimpaired received signal, in accordance with another aspect of the present invention.
- the low frequency interference suppressor 100 has an input for receiving phase samples 101 that are preferably produced by a demodulator for Manchester-coded FM signals (not shown) in accordance with the techniques described in U.S. Patent Application Serial No. 08/053,860, which has been incorporated herein by reference. It is assumed that the phase samples 101 are impaired by the presence of the above-described unwanted low frequency interference.
- the phase samples 101 are converted into frequency samples 105 by a first order difference circuit 103, which determines the difference between successive samples. That is, each of the frequency samples 105 represents frequency change per sample.
- the frequency samples 105 are reduced modulo lit , and then supplied to a low-pass filter 107 which filters at a rate of 80 Ksamples/sec in order to allow for oversampling of the 10 Kbits/sec Manchester-coded data by a factor of eight.
- the low-pass filter 107 is preferably a second-order Butterworth filter mapped to the digital domain via a bilinear transformation.
- the corner frequency of this filter whose purpose is to pass only the impairment, is 150 Hz.
- the output of the low-pass filter 107 is supplied to an integrator 113, which generates modulo 2 ⁇ values of the estimated phase impairment 115.
- Integration is done at the rate of 80 KHz, allowing the estimated phase impairment 115 to be at the correct sample rate for subtraction from time aligned copies of the original phase samples 101 in subtractor 117. (Time alignment is necessary to compensate for the delays associated with generating the estimated phase impairment.)
- the output of the integrator 113 is supplied to a subtrahend input of the subtractor 117.
- the other (positive) input of the subtractor 117 receives the output from a delay line 153, which time aligns the original phase samples 101 with the estimated phase impairment 115.
- the output of the subtractor 117 after being reduced modulo 2 ⁇ , is the corrected phase signal 119, which is supplied back to the demodulator for Manchester-coded FM signals to complete the demodulation process.
- FIG. 1(a) The circuit depicted in FIG. 1(a) is useful for explaining the theory of operation of this aspect of the invention, but is not preferred because of the higher processing requirements imposed by the need to perform a second-order low-pass filtering operation at an 80 KHz sample rate.
- FIG. 1(b) A less processing- intensive circuit that operates in accordance with the theory described above with respect to FIG. 1(a) is illustrated in FIG. 1(b) .
- the low frequency interference suppressor 150 includes many of the same components described above with respect to FIG. 1(a).
- the low-pass filter 155 (equivalent to the low-pass filter 107 shown in FIG. 1(a)) operates on frequency samples that are being supplied at a 16 KHz rate rather than at 80 KHz. In practice, such samples may be readily available.
- such samples may readily be generated by supplying 80 KHz frequency samples 105 from the first order difference circuit 103 to an anti-aliasing finite impulse response (FIR) low-pass filter 151 which will allow sample rate reduction by the decimation filter 109.
- the FIR low-pass filter 151 is preferably a linear-phase 7- tap filter with a 3-dB cutoff frequency of 6 KHz.
- the low-pass filter 155 then operates on the 16 KHz output samples of the decimation filter 109.
- the low-pass filter 155 has the same frequency response as the low-pass filter 107 described above with respect to FIG. 1(a), but has to be scaled in the bilinear transformation for the lower sample rate of 16 KHz.
- the 16 KHz output of the low-pass filter 155 is then supplied to a sample and hold circuit 111 which repeats each of the 16 KHz samples five times to generate an 80 KHz supply of filtered samples that are supplied to the integrator 113.
- the integrator 113 generates modulo 2 ⁇ values of the estimated phase impairment 115 which are subtracted from time aligned copies of the original phase samples 101.
- This alternative embodiment is preferred because it avoids the problems of time delay and distortion of the phase of the noise by converting the phase samples 101 to frequency samples 105, directly removing the undesired residual FM component, and then converting the corrected frequency samples into corrected phase samples which are then demodulated in accordance with the preferred technique.
- the low frequency interference suppressor 200 has an input for receiving phase samples 101 that are preferably produced by a demodulator for Manchester-coded FM signals (not shown) in accordance with the techniques described in U.S. Patent Application Serial No. 08/053,860, which has been incorporated herein by reference. It is again assumed that the phase samples 101 are impaired by the presence of the above-described unwanted low frequency interference.
- the phase samples 101 are converted into frequency samples 105 by a first order difference circuit 103, which determines the difference between successive samples.
- the frequency samples 105 are reduced modulo 2 ⁇ , and then supplied to a high-pass filter 201 which filters at a rate of 80 Ksamples/sec.
- the high-pass filter 201 is preferably a first-order Butterworth filter in the high-pass configuration.
- a second- or third-order filter could also be used instead, but this would impose higher processing requirements on the low frequency interference suppressor 200.
- the Butterworth filter is preferred because it exhibits a flat response in the pass band. However, this type of filter is not a requirement; any type of high-pass filter could be used instead.
- a corner (i.e., 3-dB) frequency of approximately 750 Hz works best for a first-order filter in this application.
- a higher corner frequency harms the wide-band data too much, and a lower corner frequency inadequately attenuates the impairment.
- the filtered frequency samples 203 are then supplied to an integrator 113 whose output, after reduction by modulo 2 ⁇ t , is the sequence of corrected phase samples 205.
- this arrangement operates entirely at the 80 KHz sample rate of the incoming phase samples 101, and therefore places more demands on the processing hardware that is utilized to implement the various components.
- the performance of each of the above- described embodiments was assessed by means of simulation. First, without applying any correction whatsoever, the performance of a demodulator for Manchester-coded FM signals was simulated for a received signal with and without the impairment in a static gaussian channel.
- FIG. 1(b) was also simulated to determine how it would affect the performance of a demodulator for Manchester-coded FM signals. After optimizing the amount of delay produced by the delay line 153, it was found that the lost message rate above 10 dB E b /N 0 in a static gaussian channel could be brought back to zero, but at 10 dB the system suffered a lost frame rate of 99 per cent. A similar static gaussian simulation was performed for the preferred embodiment of the low frequency interference suppressor 200 shown in FIG. 2. It was found that this configuration brought the lost message rate at 10 dB I- b /N,, down to 8 per cent.
- test results for this embodiment in flat fading with a vehicle speed of 100 km/hr are summarized in the graphs depicted in FIG. 3. It can be seen that although the preferred technique is capable of restoring performance to an acceptable level of lost messages when the received signal is impaired by noise, a price is paid for high-pass filtering a portion of the wide-band data when no impairment is present. That is, when no impairment is present, some worsening in performance occurs relative to a system with no correction whatsoever.
- the drawback introduced by high-pass filtering of an unimpaired received signal is greatly reduced.
- this is accomplished by deploying the low frequency interference suppressor 200 in a system that is further capable of detecting the presence or absence of impairment.
- the impairment detector 401 determines that the received signal has been impaired by the introduction of noise, its output (impairment present 405) controls the multiplexor (MUX) 403 to select the output of the low frequency interference suppressor 200.
- the output of the MUX 403 is then supplied to further demodulation circuitry (not shown) which operates in accordance with known techniques. If no impairment is detected, however, then the impairment detector 401 causes the MUX 403 to select the original phase samples 101, thereby avoiding the degraded performance that would otherwise be introduced by use of the low frequency interference suppressor 200.
- the impairment detector 01' receives, as an input, the frequency samples 105 that are produced by the first order difference circuit 103 (see FIG. 2).
- a low-pass filter 501 filters the frequency samples 105 to allow only those frequency components to pass that are in the band where the low-frequency impairment would appear.
- the filtered frequency samples 503 are then supplied to an energy measuring circuit 505 which measures the strength of these signals, preferably by determining the sum of squared samples.
- This energy measurement 507 is supplied to one input of a comparator 509, so that it can be compared with a threshold value 511.
- the comparator When the energy measurement 507 exceeds the threshold value 511, the comparator generates the impairment present signal 405, which is supplied to the MUX 403 as shown in FIG. 4. To determine an appropriate threshold value, the energy in the impairment band may be measured with and without the impairment present for a large number of channel conditions. Then, histograms of the energy distributions may be plotted for the two cases. A threshold value that differentiates between the two cases may then be identified and selected.
- the impairment detector 401' ' receives, as an input, the system identification (ID) 601 of the base station (not shown) with which the receiver equipment is communicating. This information is typically provided by a base station to the mobile station that it is serving so that the mobile station will know what cell it is in.
- a processor 603 compares the system ID 601 with a table of IDs 607 that are stored in a memory 605, which is preferably a read-only memory (ROM) .
- the table of IDs 607 is generated by collecting the system IDs of all cells that are known to contain the particular type of base station equipment that generates the unwanted low-frequency noise.
- the processor 603 determines that the system ID 601 matches any one of the IDs stored in the table 607, then it asserts the impairment present signal 405. It is noted that, while this solution is good in theory, it may not always produce reliable results because cellular telephone system operators occasionally move base station equipment from one cell site to another. As a result, the table 607 may not always represent the most up-to-date allocation of base station hardware. However, in a geographical region where the type of base station equipment being used in any given cell is stable, this embodiment offers a practical solution.
- phase samples 101 are produced as described above. These are supplied to each of two parallel processing paths. In one path, the phase samples 101 are supplied directly to a first demodulator circuit 701-1. In the other path, the phase samples 101 are supplied to the low frequency interference suppressor 200, whose output is supplied to a second demodulator circuit 701-2 that is functionally identical to the first demodulator circuit 701-1. The outputs from each of the first and second demodulator circuits 701-1, 701-2 are then supplied to respective ones of vtwo cyclic redundancy check (CRC) circuits 703-1, 703-2, as well as to respective first and second inputs of a MUX 705.
- CRC vtwo cyclic redundancy check
- the outputs from the two CRC circuits 703-1, 703-2 are supplied to a processor 709 which determines the identity of the path that is producing the most error-free results.
- a select signal 707 is generated by the processor 709 which causes the MUX 705 to pass the most error-free signal as the demodulated output 709 from the demodulator 700.
Abstract
Description
Claims
Priority Applications (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
AU44125/96A AU690001B2 (en) | 1994-11-28 | 1995-11-28 | Removing low frequency interference in a digital FM receiver |
CA002206044A CA2206044C (en) | 1994-11-28 | 1995-11-28 | Removing low frequency interference in a digital fm receiver |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US08/345,848 | 1994-11-28 | ||
US08/345,848 US5680418A (en) | 1994-11-28 | 1994-11-28 | Removing low frequency interference in a digital FM receiver |
Publications (1)
Publication Number | Publication Date |
---|---|
WO1996017432A1 true WO1996017432A1 (en) | 1996-06-06 |
Family
ID=23356738
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
PCT/US1995/015578 WO1996017432A1 (en) | 1994-11-28 | 1995-11-28 | Removing low frequency interference in a digital fm receiver |
Country Status (5)
Country | Link |
---|---|
US (1) | US5680418A (en) |
CN (1) | CN1068158C (en) |
AU (1) | AU690001B2 (en) |
CA (1) | CA2206044C (en) |
WO (1) | WO1996017432A1 (en) |
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JP3392670B2 (en) * | 1996-11-28 | 2003-03-31 | 株式会社東芝 | Sampling device |
US6275680B1 (en) * | 1997-07-29 | 2001-08-14 | Philips Semiconductors, Inc. | Hardware PCH checking for personal handyphone system portable station |
US7515896B1 (en) | 1998-10-21 | 2009-04-07 | Parkervision, Inc. | Method and system for down-converting an electromagnetic signal, and transforms for same, and aperture relationships |
US6091940A (en) | 1998-10-21 | 2000-07-18 | Parkervision, Inc. | Method and system for frequency up-conversion |
US6694128B1 (en) | 1998-08-18 | 2004-02-17 | Parkervision, Inc. | Frequency synthesizer using universal frequency translation technology |
US6061551A (en) | 1998-10-21 | 2000-05-09 | Parkervision, Inc. | Method and system for down-converting electromagnetic signals |
US7039372B1 (en) | 1998-10-21 | 2006-05-02 | Parkervision, Inc. | Method and system for frequency up-conversion with modulation embodiments |
US6542722B1 (en) | 1998-10-21 | 2003-04-01 | Parkervision, Inc. | Method and system for frequency up-conversion with variety of transmitter configurations |
US6061555A (en) | 1998-10-21 | 2000-05-09 | Parkervision, Inc. | Method and system for ensuring reception of a communications signal |
US7236754B2 (en) | 1999-08-23 | 2007-06-26 | Parkervision, Inc. | Method and system for frequency up-conversion |
US6049706A (en) | 1998-10-21 | 2000-04-11 | Parkervision, Inc. | Integrated frequency translation and selectivity |
US6813485B2 (en) | 1998-10-21 | 2004-11-02 | Parkervision, Inc. | Method and system for down-converting and up-converting an electromagnetic signal, and transforms for same |
US6560301B1 (en) | 1998-10-21 | 2003-05-06 | Parkervision, Inc. | Integrated frequency translation and selectivity with a variety of filter embodiments |
US6370371B1 (en) | 1998-10-21 | 2002-04-09 | Parkervision, Inc. | Applications of universal frequency translation |
US6704558B1 (en) | 1999-01-22 | 2004-03-09 | Parkervision, Inc. | Image-reject down-converter and embodiments thereof, such as the family radio service |
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US6879817B1 (en) * | 1999-04-16 | 2005-04-12 | Parkervision, Inc. | DC offset, re-radiation, and I/Q solutions using universal frequency translation technology |
US6853690B1 (en) | 1999-04-16 | 2005-02-08 | Parkervision, Inc. | Method, system and apparatus for balanced frequency up-conversion of a baseband signal and 4-phase receiver and transceiver embodiments |
US6430401B1 (en) * | 1999-03-29 | 2002-08-06 | Lucent Technologies Inc. | Technique for effectively communicating multiple digital representations of a signal |
US7065162B1 (en) | 1999-04-16 | 2006-06-20 | Parkervision, Inc. | Method and system for down-converting an electromagnetic signal, and transforms for same |
US7110444B1 (en) | 1999-08-04 | 2006-09-19 | Parkervision, Inc. | Wireless local area network (WLAN) using universal frequency translation technology including multi-phase embodiments and circuit implementations |
US7693230B2 (en) | 1999-04-16 | 2010-04-06 | Parkervision, Inc. | Apparatus and method of differential IQ frequency up-conversion |
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US6774822B1 (en) * | 2003-01-09 | 2004-08-10 | Process Control Corporation | Method and systems for filtering unwanted noise in a material metering machine |
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US7689132B2 (en) * | 2005-06-07 | 2010-03-30 | Industrial Technology Research Institute | Interference-rejection coding method for an optical wireless communication system and the optical wireless communication system thereof |
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TWI426397B (en) * | 2009-06-29 | 2014-02-11 | Lee Ming Inst Technology | Can be used in a signal interval in the unequal spacing of the sample, the signal in this interval between a single and multiple numerical integration device. |
US8600332B2 (en) | 2010-03-31 | 2013-12-03 | Apple Inc. | Electronic devices having interferers aligned with receiver filters |
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-
1995
- 1995-11-28 CA CA002206044A patent/CA2206044C/en not_active Expired - Fee Related
- 1995-11-28 WO PCT/US1995/015578 patent/WO1996017432A1/en active Application Filing
- 1995-11-28 CN CN95196503A patent/CN1068158C/en not_active Expired - Fee Related
- 1995-11-28 AU AU44125/96A patent/AU690001B2/en not_active Ceased
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US3593150A (en) * | 1969-12-03 | 1971-07-13 | Kakusai Denshin Denwa Kk | Phase- and frequency-fluctuation included in a transmitted signal |
FR2122376A1 (en) * | 1971-01-21 | 1972-09-01 | Ibm France | |
WO1991014329A1 (en) * | 1990-03-08 | 1991-09-19 | Telefonaktiebolaget Lm Ericsson | Direct phase digitization |
US5283815A (en) * | 1991-05-21 | 1994-02-01 | General Electric Company | Tangental type differential detector for pulse shaped PI/4 shifted differentially encoded quadrature phase shift keying |
WO1994026024A1 (en) * | 1993-04-29 | 1994-11-10 | Ericsson Ge Mobile Communications Inc. | Demodulator for manchester-coded fm signals |
Also Published As
Publication number | Publication date |
---|---|
CN1068158C (en) | 2001-07-04 |
CA2206044A1 (en) | 1996-06-06 |
MX9703890A (en) | 1998-07-31 |
US5680418A (en) | 1997-10-21 |
AU4412596A (en) | 1996-06-19 |
CA2206044C (en) | 2005-03-29 |
CN1167549A (en) | 1997-12-10 |
AU690001B2 (en) | 1998-04-09 |
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